Low voltage sensing and control of battery referenced transistors in subscriber loop applications

This invention relates to the field of telecommunications. In particular, this invention is drawn to subscriber loop interface circuitry.

Subscriber line interface circuits are typically found in the central office exchange of a telecommunications network. A subscriber line interface circuit (SLIC) provides a communications interface between the digital switching network of a central office and an analog subscriber line. The analog subscriber line connects to a subscriber station or telephone instrument at a location remote from the central office exchange.

The analog subscriber line and subscriber equipment form a subscriber loop. The interface requirements of an SLIC result in the need to provide relatively high voltages and currents for control signaling with respect to the subscriber equipment on the subscriber loop. Voiceband communications are low voltage analog signals on the subscriber loop. Thus the SLIC must detect and transform low voltage analog signals into digital data for transmitting communications received from the subscriber equipment to the digital network. For bi-directional communication, the SLIC must also transform digital data received from the digital network into low voltage analog signals for transmission on the subscriber loop to the subscriber equipment. Strict gain and longitudinal balance control are required for subscriber loop applications.

Multiple high voltage operational amplifiers are frequently used to drive the subscriber loop. The battery feed potential is often used as the negative supply voltage for the operational amplifiers. The operational amplifiers drive the tip and ring voltages of the subscriber loop at levels ranging from ground potential to the battery feed potential, during battery feed and subscriber loop ringing.

One disadvantage of the use of high voltage operational amplifiers is that the operational amplifier bias currents result in non-negligible power losses due to the large potential difference between the positive power supply potential and the battery feed potential. The operational amplifiers are also custom integrated circuits manufactured in a “non-mainstream” high voltage bipolar technology. Yet another disadvantage of this approach is that the operational amplifiers must be closely matched in performance characteristics over a large operating range to ensure proper performance.

In view of limitations of known systems and methods, a subscriber line interface circuit is described. Generally the tip and ring currents or the departure from the desired tip and ring currents is calculated without direct sensing of either the tip or ring lines of the subscriber loop. These calculated signals are then used to control the tip and ring currents.

A method includes the step of providing subscriber loop pull-down circuitry operating in a first voltage domain. The subscriber loop pull-down circuitry decreases at least one of a tip and a ring line current in response to a corresponding pull-down control signal. The method further includes the step of providing control circuitry operating in a second voltage domain, wherein the first and second voltage domains are substantially distinct. The control circuitry varies the pull-down control signal to vary a selected one of the tip and ring line currents in response to a sensed current corresponding to an associated one of a tip pull-down current and a ring pull-down current.

A subscriber line interface circuit apparatus includes pull-down circuitry to vary line currents of the tip and ring lines in response to a pull-down control signal. The pull-down circuitry operates in a first voltage domain. The apparatus includes control circuitry generating the pull-down control signal. The control circuitry operates in a second voltage domain substantially distinct from the first voltage domain. A control isolation stage is coupled to provide the pull-down control signals from the control circuitry to the pull-down circuitry. A feedback isolation stage provides feedback signals from the pull-down circuitry to the control circuitry. The feedback signals represent sensed pull-down currents associated with the tip and ring lines. The control circuitry provides the pull-down control signals for the tip and ring lines in response to the sensed pull-down currents.

Other features and advantages of the present invention will be apparent from the accompanying drawings and from the detailed description that follows.

The present invention is illustrated by way of example and not limitation in the figures of the accompanying drawings, in which like references indicate similar elements and in which:

FIG. 1 illustrates one embodiment of a central office exchange including a subscriber line interface circuit (SLIC) coupling subscriber equipment to a digital switching system.

FIG. 2 illustrates a block diagram of an SLIC including a signal processor and a linefeed driver.

FIG. 3 illustrates high voltage precision matched operational amplifiers for driving tip and ring line voltages.

FIG. 4 illustrates one embodiment of a linefeed driver circuit.

FIG. 5 illustrates current differencing and mirroring circuitry.

FIG. 6 illustrates the current differencing and mirroring circuitry used to generate feedback error signals for the tip and ring control circuitry within the signal processor.

FIG. 1 illustrates functional elements of one embodiment of a subscriber line interface circuit (SLIC) 110 typically associated with plain old telephone services (POTS) telephone lines. The subscriber line interface circuit (SLIC) provides an interface between a digital switching network 120 of a local telephone company central exchange and a subscriber loop 132 including subscriber equipment 130 .

The subscriber loop 132 is typically used for communicating analog data signals (e.g., voiceband communications) as well as subscriber loop “handshaking” or control signals. The analog data signals are typically on the order of 1 volt peak-to-peak (i.e., “small signal”). The subscriber loop control signals typically consist of a 48 VDC offset and an AC signal of 40-140 Vrms (i.e., “large signal”). The subscriber loop state is often specified in terms of the tip 180 and ring 190 portions of the subscriber loop.

The SLIC is expected to perform a number of functions often collectively referred to as the BORSCHT requirements. BORSCHT is an acronym for “battery feed,” “overvoltage protection,” “ring,” “supervision,” “codec,” “hybrid,” and “test.”

Recent transformerless SLIC designs tend to distribute the functional requirements between two integrated circuits based on whether the functions are traditionally associated with the high voltage subscriber loop controls or the low voltage data processing. For example, in one embodiment, the codec is implemented in a low voltage integrated circuit and the remaining functions (e.g., supervision) are implemented primarily in a high voltage integrated circuit such as a bipolar integrated circuit. Although this design tends to offer considerable space, weight, and power efficiencies over designs requiring passive inductive components, this distribution of the functional requirements tends to result in a relatively expensive high voltage integrated circuit.

FIG. 2 illustrates one embodiment of a SLIC wherein the BORSCHT functions are distributed between a signal processor 210 and a linefeed driver 220 . Signal processor 210 is responsible for at least the battery feed control, ringing control, supervision, codec, and hybrid functions. Signal processor 210 controls and interprets the large signal subscriber loop control signals as well as handling the small signal analog voiceband signals and the digital voiceband data. In one embodiment, the signal processor 210 is an integrated circuit.

In one embodiment, the signal processor includes a processor interface 214 to enable programmatic control of the signal processor 210 . The processor interface effectively enables programmatic or dynamic control of battery control, battery feed state control, voiceband signal amplification and level shifting, longitudinal balance, ringing currents, and other subscriber loop control parameters as well as setting thresholds such as a ring trip detection thresholds and an off-hook detection threshold.

Signal processor 210 includes a codec for bi-directional transformation of the voiceband communications between the digital and analog domains as is well known in the art. The digital voiceband data is received from the digital switching network on interface 216 . Within the signal processor, the digital voiceband data is coupled to a digital codec interface. An analog codec interface provides outgoing analog voiceband signals to the linefeed driver. The analog codec interface also receives incoming analog voiceband signals from the linefeed driver. The terms “incoming” and “outgoing” used in reference to the voiceband (i.e., audio) signal refer to the intended data flow from the perspective of the digital switching network. Thus, incoming voiceband signals received from the subscriber line are transformed from analog to digital form and provided to the digital switching network. Outgoing voiceband signals from the digital switching network are transformed from digital to analog form and provided to the subscriber line for use by the subscriber equipment.

Signal processor 210 receives subscriber line state information from linefeed driver 220 as indicated by tip/ring sense 222 . This information is used to generate control signals for linefeed driver 220 as indicated by linefeed driver control 212 . In one embodiment, the linefeed driver control and outgoing analog voiceband signals are communicated on the same signal lines 212 . Incoming analog voiceband signals are received by the signal processor on line 230 .

Linefeed driver 220 maintains responsibility for battery feed to tip 280 and ring 290 . Overvoltage protection is not explicitly illustrated, however, overvoltage protection can be provided by fuses and/or a network of clamping devices incorporated into linefeed driver 220 , if desired. Linefeed driver 220 includes sense circuitry to provide signal processor 210 with pre-determined sensed subscriber loop operating parameters as indicated by tip/ring sense 222 . Signal processor 210 performs any necessary processing on the sensed parameters in order to determine the operational state of the subscriber loop. For example, differences or sums of sensed voltages and currents are performed as necessary by signal processor 210 rather than linefeed driver 220 . Thus common mode and differential mode components (e.g., voltage and current) of the subscriber loop are calculated by the signal processor rather than the linefeed driver.

Linefeed driver 220 modifies the large signal tip and ring operating conditions in response to linefeed driver control 212 provided by signal processor 210 . This arrangement enables the signal processor to perform processing as needed to handle the majority of the BORSCHT functions. For example, the supervisory functions of ring trip, ground key, and off-hook detection can be determined by signal processor 210 based on operating parameters provided by tip/ring sense 222 .

FIG. 3 illustrates one embodiment of the large signal linefeed driver circuitry for controlling tip and ring line voltages. Operational amplifier 320 controls the tip 380 portion of the subscriber line. Operational amplifier 330 controls the ring 390 portion of the subscriber line. Operational amplifiers 320 and 330 drive the tip 380 and ring 390 lines in accordance with linefeed control signals 310 .

As illustrated in FIG. 3 , operational amplifiers typically use V DD (approximately 5 VDC) or ground for the positive rail supply voltage and V BAT (−24 to −75 VDC) for the negative rail supply voltage. Thus the amplifiers are relatively high voltage amplifiers. Moreover, due to the potential difference between V DD and V BAT , the operational amplifier bias currents result in non-negligible power consumption. Due to subscriber loop operational specifications, operational amplifiers 320 and 330 must be matched over a relatively large operating voltage range. The operational amplifiers are frequently constructed on a shared substrate within a high voltage integrated circuit in part to facilitate matching. The dual high voltage precision operational amplifier approach tends to result in increased costs for the SLIC.

FIG. 4 illustrates an alternative SLIC linefeed driver 410 . In one embodiment, the linefeed driver 410 is implemented as a number of discrete components. Linefeed driver 410 includes voiceband sensing circuitry 420 and power circuitry 440 .

Voiceband circuitry 420 enables retrieval of voiceband communications from the subscriber loop. Nodes 424 and 428 serve to communicate voiceband signals from the subscriber loop to signal processor 210 (i.e., “incoming audio”). Capacitors CR and CT effectively provide AC coupling for the incoming audio signal from the subscriber loop to the signal processor while decoupling signal processor 210 from the DC offsets of the tip 480 and ring 490 nodes. Thus capacitors CR and CT effectively provide DC isolation of the incoming analog audio interface formed by nodes 424 and 428 from the subscriber loop. In the embodiment illustrated, voiceband circuitry 420 provides AC coupling of the incoming analog audio signal between the subscriber loop and the signal processor using only passive components.

Power circuitry 440 provides the battery feed and other relatively high voltage functions to the subscriber loop in accordance with analog linefeed control signals provided by the signal processor 210 at nodes 442 , 444 , 446 , and 448 . These control signals act as pull-up/pull-down controls to manipulate the tip and ring currents. Processing of the sensed parameters of the tip and ring lines for generating the linefeed control signals is handled exclusively by signal processor 210 .

The subscriber loop current and the tip and ring voltages are controlled by transistors Q 1 -Q 6 . Transistors Q 1 -Q 4 are coupled in a common base configuration. Transistors Q 5 -Q 6 are coupled in a common emitter configuration. The common base/common emitter combination isolates the low voltage circuitry of the signal processor from the high voltage of the subscriber loop while providing pull-down current capabilities into the battery feed.

Transistors Q 5 -Q 6 function as pull-down circuitry to enable decreasing the tip and ring currents. Transistors Q 1 -Q 4 serve as a control isolation stage to provide the control signals from the low voltage domain of the signal processor to the high voltage domain of the pull-down circuitry and the subscriber line. The voltage domain of the signal processor is approximately 0-5.0 volts. The voltage domain of the pull-down circuitry is approximately -V BAT to 0.0 volts. Due to the transistor junctions between the signal ground and the remainder of the pull-down circuitry, the voltage domains of the signal processor and the pull-down circuitry will not intersect during normal operation.

In one embodiment, Q 1 -Q 4 are PNP bipolar junction transistors and Q 5 -Q 6 are NPN bipolar junction transistors. Given that the base terminals of Q 1 -Q 4 are coupled to ground, nodes 442 - 448 need only be approximately 0.7 volts to turn on transistors Q 1 -Q 4 . Due to the small voltage drop between the base and emitters of Q 1 -Q 4 , control of the linefeed circuitry requires relatively low power and thus linefeed driver control currents I 1 -I 4 may be provided by a signal processor 210 implemented as a low voltage complementary metal oxide semiconductor (CMOS) integrated circuit.

Transistors Q 1 , Q 4 , and Q 6 (and resistor R 2 ) control the tip voltage 480 . The tip voltage is increased by the application of control current I 1 to Q 1 . The tip voltage (node 480 ) is decreased by the application of control current I 4 to Q 4 . Control currents I 1 and I 4 provide pull-up and pull-down tip control signals for manipulating the tip voltage at node 480 .

Similarly, transistors Q 2 , Q 3 , and Q 5 (and resistor R 1 ) control the ring voltage 490 . The application of control current I 3 to Q 3 increases the ring voltage. Thus I 3 represents the pull-up control signal for the ring voltage. The ring voltage is decreased by the application of control current I 2 to Q 2 . Control current I 2 is the ring voltage pull-down control signal. Control currents I 2 and I 3 provide pull-down and pull-up ring control signals for manipulating the ring voltage at node 490 .

Control currents I 1 -I 4 thus control the large signal subscriber loop current and tip and ring voltages. For example, the ringing signal can be generated by using the control signals at nodes 442 - 448 to periodically reverse the polarity of tip 480 with respect to ring 490 (i.e., battery polarity reversal) at the nominal ringing frequency.

Transistors Q 1 -Q 6 are selected to have sufficiently high betas so that base currents are negligible. Thus the tip current (I TIP ) can be approximated as I TIP =I 1 −I EQ6 . The pull-up and pull-down controls are operated in a substantially mutually exclusive manner such that only one of I 1 or I 4 is nonzero at given point in time. The tip current is thus either I 1 or −I EQ6 . As a result, the tip current can be determined indirectly by sensing the emitter current of transistor Q 6 . The ring current is either I 3 or −I EQ5 and can be determined indirectly by sensing the emitter current of Q 5 . The tip and ring currents can thus be determined without direct sensing of the tip and ring lines.

Power circuitry 440 includes line sensing circuitry to enable determination of currents I EQ5 and I EQ6 . The line sensing circuitry includes sense resistor RR located in the emitter path of Q 5 between the emitter of Q 5 and V BAT and sense resistor RT located in the emitter path of Q 6 between the emitter of Q 6 and V BAT . Resistors RT and RR are used as sense impedances to generate a voltage drop (e.g., V EQ6 -V BAT and V EQ5 -V BAT ) for determining I EQ6 and I EQ5 . The voltage drop across RT is sensed using resistors RS 1 and RS 3 . The voltage drop across RR is sensed using RS 2 and RS 3 . Resistors RS 1 , RS 2 , and RS 3 convert the voltages at nodes V EQ5 , V EQ6 , and V BAT into sense currents IS 1 , IS 2 , and IS 3 , respectively, for processing by signal processor 210 . In one embodiment, the line sensing circuitry consists only of passive discrete components.

Referring to FIGS. 2 and 4 , tip/ring sense 222 includes the sensed currents IS 1 , IS 2 , and IS 3 for determination of I EQ6 . Currents IS 1 , IS 2 , and IS 3 are provided to nodes 432 , 434 , and 436 so that the signal processor can perform the appropriate calculations to control the tip and ring currents from the sensed currents. The sensed parameters (IS 1 , IS 2 , and IS 3 ) enable the signal processor 210 to determine the subscriber loop common mode and differential mode currents. Generally, the tip and ring currents or the departure from the desired tip and ring currents is determined indirectly in the low voltage domain of the signal processor without directly sensing the high voltage, high current tip and ring lines. Thus resistors RS 1 , RS 2 , and RS 3 form a sense or feedback isolation stage to enable providing sensed parameters from the high voltage, high current domain of the power circuitry to the low voltage domain of the signal processor.

The voltage across the emitter resistor RT is proportional to the current flowing in the emitter of Q 6 . In particular,

V EQ6 −V BAT =I EQ6 ·RT

Assuming RS 1 and RS 2 are significantly larger than emitter resistors RR and RT (and RS 1 ≈RS 3 and the nodal voltages V 432 , V 434 , and V 436 are substantially the same), the difference in sense currents IS 1 and IS 3 represents a measure of I EQ6 in accordance with the equation:

I EQ ⁢ ⁢ 6 = ( IS ⁢ ⁢ 3 - IS ⁢ ⁢ 1 ) · RS ⁢ ⁢ 1 RT

Similarly,

=

FIG. 5 illustrates low voltage tip/ring current differencing and mirroring circuitry 550 that may be incorporated into signal processor 210 for determining currents I EQ6 and I EQ5 . Operational amplifier 510 and transistors M 1 , M 2 , and M 3 are used to invert and mirror IS 3 to enable calculating the difference between currents IS 1 and IS 3 as well as the difference between IS 2 and IS 3 . Operational amplifier 520 provides a tip sense signal 580 indicative of the current I EQ6 flowing through the emitter of Q 6 . Similarly operational amplifier 530 provides a ring sense signal 590 indicative of the current I EQ5 flowing through the emitter of Q 5 . Assuming a high beta (β) for Q 5 and Q 6 , these emitter currents represent the subscriber line tip and ring pull-down currents. If the pull-up and pull-down control signals are operated mutually exclusively in a push-pull fashion, the pull-down currents (when non-zero) represent the tip and ring currents (allowing for a change of sign).

Operational amplifier 510 provides a virtual short circuit terminating one end of resistor RS 3 at a potential voltage equivalent to V REF due to the presence of V REF at the inverting input of operational amplifier 510 . If V REF is signal ground, then RS 3 is effectively terminated at a virtual ground.

Transistors M 1 and M 2 are coupled in a current mirror configuration such that current IS 3 is mirrored through the drain of transistor M 2 . Similarly transistor M 3 is coupled in a current mirror configuration with transistor M 1 to provide current IS 3 through the drain of transistor M 3 .

Operational amplifier 520 is a transimpedance amplifier that forces the difference between IS 3 and IS 1 to flow across feedback resistor RFT. As long as V REF at the inverting input of amplifier 510 is the same as V REF at the noninverting input of operational amplifier 520 , operational amplifier 520 effectively generates a signal corresponding to the difference between IS 1 and IS 3 . The difference between these currents is proportional to the emitter current through transistor Q 6 (i.e., I EQ6 ∝IS 3 −IS 1 ). Thus the voltage produced as the tip sense signal 580 is proportional to I EQ6 (i.e., V 580 ∝RFT·I EQ6 ).

Transimpedance amplifier 530 similarly generates a ring sense voltage 590 proportional to the difference between IS 2 and IS 3 . The difference between these currents is proportional to the emitter current through transistor Q 5 (i.e., I EQ5 ∝IS 3 −IS 2 ). Thus the ring sense voltage V 590 is proportional to I EQ5 (i.e., V 590 −RFR·I EQ5 ).

FIG. 6 illustrates the current differencing and mirroring circuitry 550 in block form as tip/ring sense circuitry 650 within signal processor 610 . In one embodiment, amplifiers 624 and 634 are transconductance amplifiers for converting voltage levels corresponding to desired tip and ring currents into tip and ring pull-up currents I 1 and I 3 . When non-zero, the tip and ring pull-up currents also correspond to the desired tip and ring currents.

The tip sense signal 680 is compared with a signal 622 corresponding to the desired tip current. Differential amplifier 620 generates a tip error signal ε TIP as feedback for the tip pull-down control 660 . Tip pull-down control 660 varies 14 in response to ε TIP . Generally, when the actual tip current falls below the desired tip current, tip pull-up current I 1 is applied. When the actual tip current exceeds the desired tip current, I 4 is applied to increase the pull-down current I EQ5 . In one embodiment, tip and ring pull-down controls 660 and 670 are transconductance amplifiers.

The ring sense signal 690 is similarly compared with a signal 632 corresponding to the desired ring current. Differential amplifier 630 generates a ring error signal ε RING as feedback for the ring pull-down control 670 . Ring pull-down control 670 varies control currents I 2 in response to ε RING . When the actual ring current falls below the desired ring current, ring pull-up current I 3 is applied. When the actual ring current exceeds the desired ring current, I 2 is applied to increase the pull-down current I EQ6 .

Referring to FIG. 4 , the values of the impedances embodied by R 1 and R 2 may be selected to achieve the desired frequency response and current transfer characteristics between the control currents I 1 -I 4 and the emitter currents I EQ5 and I EQ6 . In alternative embodiments the impedances may comprise, for example, passive networks of resistors and capacitors, or active components rather than single resistors R 1 and R 2 as illustrated.

The line sensing circuitry enables signal processor 210 to determine the large signal state of the subscriber loop without the need for intervening active circuitry or level shifters. The line sensing circuitry allows sensing of the high voltage circuitry by the low voltage signal processor. The tip and ring error signals are generated in the low voltage domain of the signal processor. The common base/common emitter configuration isolates the low voltage signal processor from the high voltages of the subscriber line while providing pull-down current capabilities into the battery feed. In one embodiment, the signal processor resides in a low voltage integrated circuit package and the linefeed control circuitry is external to that package such that the signal processor and linefeed control circuitry do not reside on a same semiconductor substrate.

In one embodiment, the line sensing circuitry comprises only passive discrete components. The linefeed control inputs 442 - 448 enable signal processor 210 to actively manage the large signal state of the subscriber loop. The large signal AC and DC control loops are effectively terminated at the signal processor 210 . In particular, the large signal AC and DC components of the subscriber loop control protocol can now be controlled directly by a low voltage integrated circuit. Signal processing and state determination such as off-hook, ring trip, and ring control formerly associated with high power analog circuitry can be handled predominately by the low voltage integrated circuit. In addition, the integrated circuit signal processor can handle processing of the small signal analog voiceband signals from the subscriber loop without the need for intervening active elements or level shifting circuitry.

In one embodiment, the outgoing analog audio signal is superimposed on the control currents I 1 and I 3 for power circuitry 440 . Thus the outgoing audio signal and the linefeed control signals are provided on the same signal lines to the linefeed driver circuitry. The outgoing audio signal is communicated using nodes 442 and 446 . One advantage of this configuration is that the termination impedance can be set by controlling currents I 1 and I 3 . The use of a programmable signal processor effectively places the value of the termination impedance under programmatic control.

Transistors Q 1 and Q 3 are coupled in a common base configuration. Transistors Q 1 and Q 3 couple the outgoing audio signal received from the signal processor. In one embodiment, an audio current source manipulates I 1 and I 3 to put the outgoing audio signal onto the tip 480 and ring 490 nodes. This can be accomplished, for example, by superimposing the audio signal current source on the large signal control currents provided by tip control 660 and ring control 670 . The common base isolation stage effectively isolates the signal processor from the DC offset of the tip 480 and ring 490 nodes.

A DC bias current is established in Q 1 and Q 3 with non-precision low voltage and high voltage circuitry. The DC bias does not directly affect the audio gain or balance and thus high precision is not required. Subscriber line impedance synthesis can be accomplished by providing sensed tip and ring voltages as feedback for the audio current source. Greater gain and balance control can be achieved through the use of transistors with higher or better matched betas. Alternatively, other configurations such as Darlington pairs can be used to achieve a greater beta. Different types of transistors such as metal oxide semiconductor or junction field effect transistors (i.e., MOSFET or JFET) can be used for either the common base isolation stage (Q 1 -Q 4 ) or the drive transistors (Q 5 -Q 6 ) in alternative embodiments. The term “common base” includes “common gate” equivalents for MOSFET and JFET transistors. Thus a “common base isolation stage” is intended to include field effect transistors coupled in a common gate configuration.

In the preceding detailed description, the invention is described with reference to specific exemplary embodiments thereof. Various modifications and changes may be made thereto without departing from the broader spirit and scope of the invention as set forth in the claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.