Converter circuit including switch element unit

The present invention relates to a converter circuit (DC/DC converter) for a hybrid power source including, e.g., a solar cell, a secondary battery as a backup power source of the solar cell and the like.

In a photovoltaic system, an electric power generated by a solar cell is influenced by weather conditions and the like, and a voltage is changed by a temperature variation in the solar cell itself and the like. Therefore, a secondary battery is used as a backup power source, and when the amount of the electric power generated by the solar cell is small, the electric power is discharged from the secondary battery to stabilize the electric power supplied to a load. The voltage of the secondary battery is set to be slightly lower than the voltage at which the electric power generation of the solar cell is stable and is charged from the solar cell when the electric power consumed by the load is small.

In a hybrid power source including a solar cell, a secondary battery for backup of the solar cell and the like, since the voltage of the solar cell is different from the rated voltage of the load and the secondary battery, the voltage is stepped up/down by using a DC/DC converter to supply an electric power to the load. In the general circuit configuration, a DC/DC converter is provided for each of the solar cell and the secondary battery.

Therefore, in designing the DC/DC converter, it is important to achieve both the miniaturization and high efficiency of the DC/DC converter, and it has been proposed that a plurality of DC power sources share a transformer and a rectifier circuit (see, e.g., JP 2005-2297, herein after Patent Document 1).

Patent Document 1 does not disclose a specific circuit configuration of the DC/DC converter, but FIG. 12 shows a circuit configuration of a DC/DC converter 50 using a general MOSFET as a switch element. A first and a second primary winding N 51 and N 52 corresponding to a first and a second DC power source 51 and 52 , respectively, are provided on a primary side of a transformer 53 , and one secondary winding N 53 corresponding to the load 57 is provided on the secondary side of the transformer 53 . A first oscillation circuit 54 having a full-bridge structure, which is formed of four switch elements Q 51 to Q 54 , is connected to the first primary winding N 51 . Further, a second oscillation circuit 55 having a full-bridge structure, which is formed of four switch elements Q 55 to Q 58 , is connected to the second primary winding N 52 . A rectifier circuit 56 is connected to the secondary winding N 53 .

The first DC power source 51 is a solar cell and the second DC power source 52 is a secondary battery. The voltage of the solar cell is VDC 1 , and a reference voltage of the solar cell is Vref 1 . The voltage of the secondary battery is VDC 2 , and a reference voltage of the secondary battery is Vref 2 . The number of turns of the first primary winding N 51 and the number of turns of the second primary winding N 52 are n 1 and n 2 , respectively.

In order that the output voltage of the load by the discharge operation from the secondary battery and the power generation of the solar cell is kept constant, it is preferable to set a turns ratio n 2 /n 1 of the primary windings N 51 and N 52 to satisfy Vref 1 ×n 2 /n 1 =Vref 2 .

However, considering that the secondary battery is charged from the solar cell, it is preferable to set a turns ratio n 2 /n 1 of the primary windings N 51 and N 52 to satisfy Vref 1 ×n 2 /n 1 >Vref 2 .

In practice, since the voltage applied to the load 57 is not constant and has a tolerance value, it is set to satisfy Vref 1 ×n 2 /n 1 >Vref 2 . However, in order to more easily describe the nature of the problem in the present invention, the problem will be described below on the assumption that a turns ratio satisfies Vref 1 ×n 2 /n 1 =Vref 2 .

FIG. 13 shows a state where in the case of VDC 1 ×n 2 /n 1 >VDC 2 , for example, under the condition that the voltage of the solar cell is varied to be larger than the reference voltage Vref 1 and the voltage of the secondary battery is the reference voltage Vref 2 (VDC 1 >Vref 1 , VDC 2 =Vref 2 ), the switch elements Q 55 to Q 58 are turned off while the switch elements Q 51 and Q 54 and the switch elements Q 52 and Q 53 are alternately turned on and off, so that the electric power is supplied to the load 57 from the first DC power source 51 . In FIG. 13 , the switch elements Q 51 and Q 54 are being turned on. When supplying the electric power to the load 57 from both the first and the second DC power source 51 and 52 , the switch elements Q 51 and Q 54 , the switch elements Q 52 and Q 53 , the switch elements Q 55 and Q 58 , and the switch elements Q 56 and Q 57 may be sequentially turned on by time division.

Under the conditions of VDC 1 >Vref 1 and VDC 2 =Vref 2 , the voltage VN 52 of the primary winding N 52 satisfies VN 52 =VDC 1 ×n 2 /n 1 =VDC 1 ×Vref 2 /Vref 1 >VDC 2 by the induced electromotive force generated in the primary winding N 52 from the primary winding N 51 . Thus, the voltage of the primary winding N 52 becomes larger than VDC 2 . Since MOSFET has a body diode (parasitic diode), a reverse current flows through the second DC power source 52 by the electromotive force generated in the second primary winding N 52 via the body diodes of the switch elements Q 55 and Q 58 . The same is true when the switch elements Q 52 and Q 53 are turned on. Since such a reverse current becomes a charging current to the secondary battery, substantially, the first DC power source 51 charges the secondary battery while supplying the electric power to the load 57 , thereby resulting in an increase in the current flowing through the first oscillation circuit 54 on the side of the first DC power source 51 .

Accordingly, the loss due to the switch elements Q 51 to Q 54 included in the first oscillation circuit 54 is increased, and the power supply efficiency of the first DC power source 51 is decreased. Further, the secondary battery is charged through the body diodes of the switch elements Q 55 to Q 58 , and there occurs a problem such that it cannot be charged at certain timings (even if charging is not desired, it is charged arbitrarily).

Similarly, under the condition of VDC 1 ×n 2 /n 1 <VDC 2 , when the electric power is outputted from the second DC power source 52 , a reverse current flows through the first DC power source 51 . That is, in the configuration of FIG. 13 , the voltage of the first and the second DC power source 51 and 52 varies, and there occurs a problem such that the efficiency is deteriorated in the case of VDC 1 ×n 2 /n 1 ≠VDC 2 .

In another conventional example shown in FIG. 14 , in order to prevent the reverse current from flowing through the first DC power source 51 or the second DC power source 52 , backflow prevention diodes D 51 to D 58 are connected in series to the switch elements Q 51 to Q 58 , respectively, in opposite directions to the body diodes (see Chen et al. “Multi-Input DC/DC Converter Based on the Flux Additivity,” herein after Non-patent Document 1).

However, when a current flows in the forward direction through the backflow prevention diodes, the loss due to the diodes is increased and the power supply efficiency from the first DC power source 51 or the second DC power source 52 is decreased. Further, it is necessary to add the backflow prevention diodes D 51 to D 58 or choke coils C 51 and C 52 to the oscillation circuits 54 and 55 , which results in reducing an advantage of the miniaturization of the DC/DC converter 50 obtained by sharing the transformer 53 and the rectifier circuit 56 . In addition, since the reverse current does not flow through the second DC power source 52 by the backflow prevention diodes, the secondary battery cannot be charged by using this DC/DC converter.

The present invention provides a converter circuit for a hybrid power source in which a transformer and a rectifier circuit are commonly used, capable of preventing a reverse current from flowing through one DC power source when supplying an electric power to a load from the other DC power source and reducing a loss without reducing a power supply efficiency. Further, if necessary, it is possible to charge the secondary battery via a DC/DC converter.

In accordance with an embodiment of the present invention, there is provided a converter circuit including: a transformer having primary windings and at least one secondary winding; a rectifier circuit connected to the secondary winding; and oscillation circuits connected to the primary windings. Each of the oscillation circuits includes a switch element unit having no body diode.

In accordance with another embodiment of the present invention, there is provided a converter circuit including: a transformer having a first primary winding, a second primary winding and a secondary winding; a rectifier circuit connected between the secondary winding and a load; a first oscillation circuit connected to a first DC power source and the first primary winding; a second oscillation circuit connected to a second DC power source and the second primary winding; and a control circuit configured to control the first and the second oscillation circuit. Each of the first and the second oscillation circuit includes a switch element unit having no body diode.

Further, the switch element unit having no body diode may have a lateral transistor structure using a GaN/AlGaN structure.

Further, the switch element unit having no body diode may be a bidirectional switch element.

Further, the switch element unit of each of the first and the second oscillation circuit may include two pairs of switch elements forming a full-bridge circuit.

Further, the switch element unit of each of the first and the second oscillation circuit may include a pair of switch elements forming a half-bridge circuit.

Further, the switch element unit of each of the first and the second oscillation circuit may include one switch element.

Further, the transformer may be a flyback transformer.

Further, the number of turns may be variable in at least one of the first and the second primary winding.

In accordance with the present invention, since the switch element unit having no body diode is used as a switch element constituting the oscillation circuit, a reverse current does not flow through the oscillation circuit when the switch element is not turned on. Therefore, when supplying the electric power to the load from one DC power source, if the switch element of the oscillation circuit connected to the other DC power source is turned off, a reverse current does not flow through the other DC power source, and the power supply efficiency is not reduced. Further, there occurs no loss due to the body diode. Further, if necessary, when one DC power source is a secondary battery and the voltage of the other DC power source is higher than the voltage of the secondary battery, it is possible to charge the secondary battery through the DC/DC converter.

The objects and features of the present invention will become apparent from the following description of embodiments, given in conjunction with the accompanying drawings, in which:

FIG. 1 is a circuit diagram showing a circuit configuration of a DC/DC converter in accordance with an embodiment of the present invention;

FIG. 2 shows a modification example of the DC/DC converter in accordance with the embodiment of the present invention;

FIGS. 3A to 3C are time charts each showing waveforms of the gate signals of the bidirectional switches elements for driving the DC/DC converter;

FIG. 4 is a plan view showing a configuration of the bidirectional switch element (single gate);

FIG. 5 is an enlarged view of area A shown in FIG. 4 ;

FIG. 6 is a cross-sectional view taken along line VI-VI shown in FIG. 4 ;

FIG. 7 is a plan view showing a configuration of the bidirectional switch element (dual gate);

FIG. 8 is a cross-sectional view taken along line VIII-VIII shown in FIG. 7 ;

FIG. 9 shows another modification example of the DC/DC converter in accordance with the embodiment of the present invention;

FIG. 10 shows still another modification example of the DC/DC converter in accordance with the embodiment of the present invention;

FIG. 11 shows still another modification example of the DC/DC converter in accordance with the embodiment of the present invention;

FIG. 12 shows a circuit configuration of a conventional DC/DC converter using a MOSFET as a switch element;

FIG. 13 shows a state where the electric power is supplied to the load from the first DC power source in the conventional DC/DC converter; and

FIG. 14 shows a circuit configuration of the conventional DC/DC converter in which a backflow prevention diode is connected to the MOSFET.

Hereinafter, an embodiment of the present invention will be described in detail with reference to the accompanying drawings which form a part hereof. Throughout the specification and drawings, like reference numerals will be given to like parts having substantially the same function and configuration, and a redundant description thereof will be omitted.

A converter circuit (DC/DC converter) in accordance with an embodiment of the present invention will be described. FIG. 1 is a circuit diagram showing a circuit configuration of a DC/DC converter 1 . In order to simplify the explanation, a case with two power sources of a first DC power source 11 and a second DC power source 12 will be described, but the number of power sources is not limited to two.

The DC/DC converter 1 is a DC/DC converter for a hybrid power source for supplying an electric power supplied from a plurality of DC power sources including the first and the second DC power source 11 and 12 to a load 57 . The DC/DC converter 1 functions as a DC/DC converter for supplying an electric power to the load 57 from the first DC power source 11 , and also functions as a DC/DC converter for supplying an electric power to the load 57 from the second DC power source 12 . The DC/DC converter 1 may be regarded as two DC/DC converters which share a transformer 13 , a rectifier circuit 16 and a control circuit 17 .

A first primary winding N 1 and a second primary winding N 2 are provided at the primary side of the transformer 13 , and one secondary winding N 3 is provided at the secondary side of the transformer 13 . A first oscillation circuit 14 is connected to the first primary winding N 1 , and the first DC power source 11 is connected to the first oscillation circuit 14 . Similarly, a second oscillation circuit 15 is connected to the second primary winding N 2 , and the second DC power source 12 is connected to the second oscillation circuit 15 . A rectifier circuit 16 is connected to the secondary winding N 3 , and the load 57 is connected to the rectifier circuit 16 . The first oscillation circuit 14 and the second oscillation circuit 15 have basically the same configuration. The first oscillation circuit 14 includes a full bridge circuit formed of four bidirectional switch elements Q 1 to Q 4 . Similarly, the second oscillation circuit 15 includes a full bridge circuit formed of four bidirectional switch elements Q 5 to Q 8 .

In FIG. 1 , a wiring connecting the gate of each of the bidirectional switch elements Q 1 to Q 8 to the control circuit 17 is omitted. Further, in this embodiment, one secondary winding N 3 is merely provided at the secondary side of the transformer 13 , but two or more secondary windings may be provided without being limited thereto.

For example, it is assumed that the first DC power source 11 is a solar cell and the second DC power source 12 is a secondary battery. When supplying an electric power to the load 57 from the first DC power source 11 alone, as shown in FIG. 3A , the control circuit 17 alternately turns on and off a pair of the bidirectional switch elements Q 1 and Q 4 and a pair of the bidirectional switch elements Q 2 and Q 3 of the first oscillation circuit 14 . In the meantime, the control circuit 17 turns off all of the bidirectional switch elements Q 5 to Q 8 of the second oscillation circuit 15 .

On the other hand, when supplying an electric power to the load 57 from the second DC power source 12 alone, as shown in FIG. 3B , the control circuit 17 alternately turns on and off a pair of the bidirectional switch elements Q 5 and Q 8 and a pair of the bidirectional switch elements Q 6 and Q 7 of the second oscillation circuit 15 . In the meantime, the control circuit 17 turns off all of the bidirectional switch elements Q 1 to Q 4 of the first oscillation circuit 14 .

Further, when an electric power is alternately supplied from the first and the second DC power source 11 and 12 to the load 57 , as shown in FIG. 3C , a pair of the bidirectional switch elements Q 1 and Q 4 , a pair of the bidirectional switch elements Q 2 and Q 3 , a pair of the bidirectional switch elements Q 5 and Q 8 , and a pair of the bidirectional switch elements Q 6 and Q 7 are sequentially turned on and off.

In this case, assuming VDC 1 ×n 2 /n 1 >VDC 2 , where a voltage of the solar cell is VDC 1 , a voltage of the secondary battery is VDC 2 , and the number of turns of the first primary winding N 1 and the number of turns of the second primary winding N 2 are n 1 and n 2 , respectively.

In a conventional example shown in FIGS. 12 and 13 , even if the bidirectional switch elements Q 55 to Q 58 of a second oscillation circuit 55 are turned off, an electromotive force is generated in the second primary winding N 52 , and a current flows in a body diode (parasitic diode) of each of the switch elements. Accordingly, a reverse current flows in the second DC power source 52 via the switch elements Q 55 to Q 58 .

In contrast, as will be described below, since each of the bidirectional switch elements Q 1 to Q 8 has no body diode, a reverse current does not flow therethrough. That is, in accordance with the configuration of the DC/DC converter 1 of this embodiment shown in FIG. 1 , even in the case of VDC 1 ×n 2 /n 1 >VDC 2 , a reverse current does not flow in the second DC power source 12 . The first DC power source 11 does not charge the secondary battery while supplying an electric power to the load 57 , and the current, which flows through the first oscillation circuit 14 on the side of the first DC power source 11 , is not increased. In addition, the bidirectional switch elements Q 1 to Q 8 have a much lower loss compared with the MOSFET since they have no body diode. Accordingly, there is less loss due to the bidirectional switch elements Q 1 to Q 4 , and the power supply efficiency of the first DC power source 11 is improved as compared with the conventional example described above.

When charging the secondary battery, the control circuit 17 turns on/off the bidirectional switch elements Q 5 to Q 8 of the second oscillation circuit 15 in synchronization with the on/off of the bidirectional switch elements Q 1 to Q 4 of the first oscillation circuit 14 . Since the loss due to the bidirectional switch elements Q 5 to Q 8 is small, it is possible to flow a larger current as a charging current. Thus, the charging efficiency becomes higher than that in the conventional example.

Further, when an electric power is outputted from the second DC power source 12 , the reverse current does not flow in the first DC power source 11 in the similar manner described above. Therefore, the power supply efficiency when supplying an electric power from the second DC power source 12 is also improved as compared with the conventional example described above.

FIG. 2 shows a modification example of the DC/DC converter 1 . In this modification example, the number of turns of the second primary winding N 2 of the transformer 13 is varied. In FIG. 2 , although a changeover switch for changing the number of turns is not illustrated in detail, it is possible to configure a non-contact switch with low loss by using the bidirectional switch element as described above.

As shown in FIG. 3C , if the electric power is outputted alternately to the load 57 from the first and the second DC power source 11 and 12 , it is preferable to output the voltage of the second DC power source 12 after the voltage of the second DC power source 12 is stepped up to the same level as that of the first DC power source 11 .

On the other hand, when charging the second DC power source 12 serving as the secondary battery, the voltage of the electromotive force generated in the second primary winding N 2 needs to be higher than the voltage of the second DC power source 12 . Therefore, the second primary winding N 2 has at least two types of the number of turns corresponding to a turns ratio n 2 /n 1 satisfying Vref 1 ×n 2 /n 1 =Vref 2 and a turns ratio n 2 /n 1 satisfying Vref 1 ×n 2 /n 1 >Vref 2 (n 1 is constant). Further, the second primary winding N 2 may have another type of the number of turns depending on the voltage variation of the first DC power source 11 . In this case, in order to satisfy the setting condition of the turns ratio, n 2 may be constant while n 1 may be varied. Alternatively, both of n 1 and n 2 may be varied.

As a specific example of the bidirectional switch elements Q 1 to Q 8 , a bidirectional switch element 100 having a lateral transistor structure using a GaN/AlGaN structure will be described in detail. FIG. 4 is a plan view showing a configuration of the bidirectional switch element 100 . FIG. 5 is an enlarged view of area A shown in FIG. 4 , and FIG. 6 is a cross-sectional view taken along line VI-VI shown in FIG. 4 . Further, the bidirectional switch element 100 in which only one gate G is provided between two electrodes D 1 and D 2 is referred to as a single gate type.

As shown in FIG. 6 , a substrate 101 of the bidirectional switch element 100 includes a conductive layer 101 a and a GaN layer 101 b and an AlGaN layer 101 c which are formed on the conductive layer 101 a . In this embodiment, a two-dimensional electron gas layer, which is generated at a hetero interface between AlGaN and GaN, is used as a channel layer. As shown in FIG. 4 , a first electrode D 1 and a second electrode D 2 respectively connected in series with respect to the DC power source 11 or 12 and the winding N 1 or N 2 , and an intermediate potential portion S having an intermediate potential relative to the potential of the first electrode D 1 and the potential of the second electrode D 2 are formed on a surface 101 d of the substrate 101 . Further, a control electrode (gate) G is formed on the intermediate potential portion S. For example, a Schottky electrode is used as the control electrode G.

The first electrode D 1 has a comb shape having electrode portions 111 , 112 , 113 . . . arranged in parallel to one another, and the second electrode D 2 has a comb shape having electrode portions 121 , 122 , 123 . . . arranged parallel to one another. The comb-shaped electrode portions of the first electrode D 1 and the comb-shaped electrode portions of the second electrode D 2 are arranged opposite to each other. Since the intermediate potential portion S and the control electrode G are respectively disposed between the comb-shaped electrode portions 111 , 112 , 113 . . . and 121 , 122 , 123 . . . , they have a shape (substantially fish spine shape) similar to the planar shape of the space defined between the electrode portions.

Next, a lateral transistor structure of the bidirectional switch element 100 will be described. As shown in FIG. 5 , the electrode portion 111 of the first electrode D 1 and the electrode portion 121 of the second electrode D 2 are arranged such that center lines in the width direction thereof are aligned. In addition, the intermediate potential portion S and the control electrode G are positioned in parallel to the electrode portion 111 of the first electrode D 1 and the electrode portion 121 of the second electrode D 2 . Distances in the width direction from the electrode portion 111 of the first electrode D 1 and the electrode portion 121 of the second electrode D 2 to the intermediate potential portion S and the control electrode G are set such that a predetermined withstand voltage can be maintained. Distances in the longitudinal direction of the electrode portion 111 of the first electrode D 1 and the electrode portion 121 of the second electrode D 2 , i.e., perpendicular to the width direction are also set in the same manner.

In addition, such relationships are the same as those of the other electrode portions 112 and 122 , and 113 and 123 . That is, the intermediate potential portion S and the control electrode G are disposed at positions at which a predetermined withstand voltage can be maintained with respect to the first electrode D 1 and the second electrode D 2 .

Therefore, assuming that the first electrode D 1 is in a high potential side and the second electrode D 2 is in a low potential side, when the bidirectional switch element 100 is turned off, the current is completely interrupted between at least the first electrode D 1 , and the control electrode G and the intermediate potential portion S (the current is blocked directly under the control electrode (gate) G).

On the other hand, when the bidirectional switch element 100 is turned on, i.e., when a signal having a voltage equal to or higher than a predetermined threshold is applied to the control electrode G, a current flows through a path of the first electrode D 1 (electrode portion 111 . . . ), the intermediate potential portion S, and the second electrode D 2 (electrode portion 121 . . . ) as indicated by the arrow in the figure, and vice versa.

As a result, even though a threshold voltage of the signal applied to the control electrode G is lowered to the required minimum level, it is possible to securely turn on/off the bidirectional switch element 100 , thereby enabling a low on-resistance. Further, since the electrode portions 111 112 , 113 . . . of the first electrode D 1 and the electrode portions 121 , 122 , 123 . . . of the second electrode D 2 can be arranged in a comb shape, a high current can be obtained without increasing a chip size of the bidirectional switch element 100 .

FIGS. 7 and 8 show a configuration of another bidirectional switch element 300 having a lateral transistor structure using a GaN/AlGaN structure. FIG. 7 is a plan view showing the configuration of the bidirectional switch element 300 . FIG. 8 is a cross-sectional view taken along line VIII-VIII shown in FIG. 7 . Further, the bidirectional switch element 300 is referred to as a dual gate type because two gates G 1 and G 2 are provided between two electrodes D 1 and D 2 .

As shown in FIGS. 7 and 8 , the bidirectional switch element 300 of the lateral dual transistor structure is configured to have a single portion for maintaining a withstand voltage, so that it is possible to implement a bidirectional switch element with a small loss. In other words, the drain electrodes D 1 and D 2 are formed on the GaN layer, and the gate electrodes G 1 and G 2 are formed on the AlGaN layer. In a state where no voltage is applied to the gate electrodes G 1 and G 2 , an electron depletion region occurs in the two-dimensional electron gas layer generated at the AlGaN/GaN heterogeneous interface directly below the gate electrodes G 1 and G 2 , and no current flows. On the other hand, when a voltage is applied to the gate electrodes G 1 and G 2 , a current flows in the AlGaN/GaN heterogeneous interface from the drain electrode D 1 toward the drain electrode D 2 (or reversely).

To obtain a withstand voltage, a predetermined distance is required between the gate electrodes G 1 and G 2 . However, no withstand voltage is required between the drain electrode D 1 and the gate electrode G 1 and between the drain electrode D 2 and the gate electrode G 2 . Therefore, the first electrode D 1 and the gate electrode G 1 , or the drain electrode D 2 and the gate electrode G 2 may be overlapped with each other via an insulating layer In interposed therebetween. Further, the element with such a configuration needs to be controlled based on the voltages of the drain electrodes D 1 and D 2 , and therefore it is necessary to input a drive signal to the respective gate electrodes G 1 and G 2 (thus, referred to as a dual gate transistor structure).

FIG. 9 shows another modification example of the DC/DC converter 1 . In this modification example, each of the first and the second oscillation circuit 14 and 15 has a half-bridge circuit. The first oscillation circuit 14 includes a pair of bidirectional switch elements Q 11 and Q 12 connected in series, and a series circuit of capacitors C 11 and C 12 connected in parallel to the series circuit of the bidirectional switch elements Q 11 and Q 12 . The first primary winding N 1 is connected between a midpoint of the series circuit of the bidirectional switch elements Q 11 and Q 12 and a midpoint of the series circuit of the capacitors C 11 and C 12 .

Similarly, the second oscillation circuit 15 includes a pair of bidirectional switch elements Q 13 and Q 14 connected in series, and a series circuit of capacitors C 13 and C 14 connected in parallel to the series circuit of the bidirectional switch elements Q 13 and Q 14 . The second primary winding N 2 is connected between a midpoint of the series circuit of the bidirectional switch elements Q 13 and Q 14 and a midpoint of the series circuit of the capacitors C 13 and C 14 .

When the bidirectional switch element Q 11 is turned on and the bidirectional switch element Q 12 is turned off, the current flows in the bidirectional switch element Q 11 and the first primary winding N 1 of the transformer 13 by the electric charges charged in the capacitor C 11 . Further, when the bidirectional switch element Q 12 is turned on and the bidirectional switch element Q 11 is turned off, the current flows in the opposite direction in the bidirectional switch element Q 12 and the first primary winding N 1 of the transformer 13 by the electric charges charged in the capacitor C 12 . This operation is repeated, and the current flows alternately in different directions in the first primary winding N 1 of the transformer 13 . The same applies to the second oscillation circuit 15 .

In the case of this modification example, as compared with the case where each of the first and the second oscillation circuit 14 and 15 shown in FIG. 1 has a full-bridge circuit, the voltage applied to the first and the second primary winding N 1 and N 2 of the transformer 13 is ½ of the voltage of the first and the second DC power source 11 and 12 . However, it has an advantage of simplifying the structure and facilitating the control of the first and the second oscillation circuit 14 and 15

FIG. 10 shows still another modification example of the DC/DC converter 1 . In this modification example, each of the first and the second oscillation circuit 14 and 15 is configured to include a forward converter having only one of bidirectional switch elements Q 21 and Q 22 . Further, in the bidirectional switch elements Q 21 and Q 22 , there are provided reset circuits R 21 and R 22 for preventing the magnetization of the core. The control circuit 17 controls the bidirectional switch element Q 21 or Q 22 so as to repeatedly turn on and off at a predetermined frequency. When the bidirectional switch element Q 21 or Q 22 is turned on, a current flows in the first primary winding N 1 or the second primary winding N 2 of the transformer 13 , and an electromotive force is generated and a current flows in the secondary winding N 3 of the transformer 13 . Thus, by configuring the forward converter, it becomes possible to simplify the configuration of the first oscillation circuit 14 , the second oscillation circuit 15 and the rectifier circuit 16 .

FIG. 11 shows still another modification example of the DC/DC converter 1 . In this modification example, a flyback converter is configured by using a flyback transformer serving as the transformer 13 .

Each of the first and the second oscillation circuit 14 and 15 uses only one of bidirectional switch elements Q 31 and Q 32 . The control circuit 17 controls the bidirectional switch element Q 31 or Q 32 so as to repeatedly turn on and off at a predetermined frequency.

When the bidirectional switch element Q 31 or Q 32 is turned on, a current flows in the first primary winding N 1 or the second primary winding N 2 of the transformer 13 to magnetize the core of the transformer 13 . Further, when the bidirectional switch element Q 31 or Q 32 is turned on, no current flows in the secondary winding N 3 of the transformer 13 .

On the other hand, when the bidirectional switch element Q 31 or Q 32 is turned off, a current flows in the secondary winding N 3 of the transformer 13 so as to release the magnetic energy from the core. Thus, by configuring a flyback converter using a flyback transformer, it is possible to simplify the configuration of the first oscillation circuit 14 , the second oscillation circuit 15 and the rectifier circuit 16 . In the case of the flyback converter, as compared with the forward converter, the direction of the current flowing through the secondary winding N 3 of the transformer 13 is reversed.

As described above, in accordance with the embodiments of the present invention, for a hybrid power source including a plurality of DC power sources such as a solar cell, a secondary battery as a backup power source of the solar cell and the like, it is possible to provide a DC/DC converter which shares a transformer and a rectifier circuit and includes a plurality of oscillation circuits with low loss.

Further, since bidirectional switch elements, each having no body diode, are used as switch elements included in the oscillation circuit, when supplying an electric power to the load from one DC power source while the bidirectional switches elements of the oscillation circuit connected to the other DC power source are turned off, no reverse current flows through the other DC power source. Therefore, the power supply efficiency of the DC power source (the one that supplies the electric power) is not reduced, and there occurs no loss due to the body diode.

Further, if necessary, when one DC power source is used as a secondary battery and the voltage of the other DC power source is set to be higher than the voltage of the secondary battery, it is possible to charge the secondary battery through the DC/DC converter.

Further, in addition to the solar cell and the secondary battery, other DC power sources such as a fuel cell and the like may be used as a DC power source. Further, in the description of the above embodiments, a bidirectional switch element having no body diode has been illustrated as an example of a switch element included in the DC/DC converter, but any element can be used as long as it has no body diode. For example, two unidirectional switches may be used in combination.

While the invention has been shown and described with respect to the embodiments, it will be understood by those skilled in the art that various changes and modification may be made without departing from the scope of the invention as defined in the following claims.