Current-parking switching regulator downstream controller

The present invention relates to regulator circuits.

Conventional devices such as microprocessors and graphics processors that are used in high-performance digital systems may have varying current demands based on the processing workload. For example, current demands may increase dramatically when a block of logic is restarted after a stall or when a new request initiates a large computation such as the generation of a new image. Conversely, current demands may decrease dramatically when a block of logic becomes idle. When the current demand increases and sufficient power is not available, the supply voltage that is provided to the device may drop below a critical voltage level, potentially causing the device to fail to function properly. When the current demand decreases and the supply voltage that is provided to the device rises above a critical voltage level, circuits within the device may fail to function properly and may even be destroyed.

A conventional multi-phase switching regulator is an electric power conversion device that interfaces between a power supply and a device, providing current to the device and responding to changes in current demands to maintain a supply voltage level. However, a conventional multi-phase switching regulator relies on a large inductor for voltage conversion and the large inductor limits the ability of the conventional multi-phase switching regulator to quickly respond to dramatic changes in current demands (i.e., current transients). A typical 30 A phase of the conventional multi-phase switching regulator may use a 0.5 μH inductor for voltage conversion. The current response is limited to di/dt=V/L which for V=11V (dropping a 12V input to a 1V supply voltage level) and L=0.5 μH gives 22 A/μs. Increasing the current provided to a device by 10 A in would require at least 500 ns. Additionally, synchronization of the pulse width modulation switching operation may increase the current response time of the conventional multi-phase switching regulator by several microseconds. When a clock period of the device is less than the current response time, the device may fail to function properly. A 500 MHz dock has a period of 2 ns, so hundreds of dock periods may occur during a 500 ns current response time.

Thus, there is a need for improving regulation of voltage levels and/or other issues associated with the prior art.

A system and method are provided for regulating a voltage level at a load. The method configures a current source to generate a current and a voltage control mechanism to provide a portion of the current to regulate the voltage level at the load. When the voltage level at the load is greater than a maximum voltage level the current source is decoupled from the load and the current source is coupled to a current sink to reduce the voltage level at the load. An electric power conversion device for regulating the voltage level at the load comprises a current source that is coupled to an electric power source and configured to generate a current and a voltage control mechanism that is coupled between the current source and the load. A downstream controller is coupled to the voltage control mechanism and is configured to control the voltage control mechanism to provide a portion of the current to regulate the voltage level at the load and decouple the current source from the load and couple the current source to a current sink to reduce the voltage level at the load when the voltage level at the load is greater than a maximum voltage level.

FIG. 1A illustrates an electric power conversion system including an electric power conversion device that is implemented as a current-parking switching regulator with a single inductor, in accordance with one embodiment;

FIG. 1B illustrates a multi-phase switching regulator that includes multiple electric power conversion devices, in accordance with one embodiment;

FIG. 1C illustrates a current-parking switching regulator with a split inductor, in accordance with one embodiment;

FIG. 2 illustrates a flowchart of a method for regulating the voltage level provided to a load, in accordance with one embodiment;

FIG. 3A illustrates a downstream controller portion of a current-parking switching regulator, in accordance with one embodiment;

FIG. 3B illustrates waveforms corresponding to the current-parking switching regulator using the downstream controller shown in FIG. 3A , in accordance with one embodiment;

FIG. 3C illustrates another downstream controller portion of a current-parking switching regulator, in accordance with one embodiment;

FIG. 4A illustrates yet another downstream controller portion of a current-parking switching regulator, in accordance with one embodiment;

FIG. 4B illustrates waveforms corresponding to the current-parking switching regulator using the downstream controller shown in FIG. 4A , in accordance with one embodiment;

FIG. 5A illustrates another flowchart of a method for regulating the voltage level provided to a load, in accordance with one embodiment;

FIG. 5B illustrates a system including a current-parking switching regulator driving two loads using a shared current source, according to one embodiment;

FIG. 6C illustrates a diagram of the current-parking switching regulator within a system, according to one embodiment; and

FIG. 6 illustrates an exemplary system in which the various architecture and/or functionality of the various previous embodiments may be implemented.

An electric power conversion device provides a desired output voltage level to a load, such as a device. The electric power conversion device converts power received from a power source (e.g., battery or main power supply) to a supply voltage level that is provided to the load. An inductor is used to deliver additional current to the load and regulate the output voltage level with switching mechanisms modulating the average current that flows through the inductor. A capacitor is coupled between the load and ground to store any excess current (difference between the current provided through the inductor and the current delivered to the load).

FIG. 1A illustrates an electric power conversion system 100 including an electric power conversion device 120 that is implemented as a current-parking switching regulator with a single inductor L 1 , in accordance with one embodiment. The electric power conversion device 120 may be one phase of a multi-phase switching regulator, as shown in FIG. 1B . The electric power conversion device 120 is configured to provide a desired output voltage level (V L ) at the load 110 by converting power received from an electric power source 108 . The electric power conversion device 120 includes a current control mechanism and a voltage control mechanism. The current control mechanism is coupled to the electric power source 108 and the controller 105 and is operable to control the average of the current I L1 flowing through the inductor L 1 and ensure a minimum current is provided across the multiple phases of a multi-phase switching regulator. For example, as illustrated, the current control mechanism may include one or more first switching mechanisms M 1 and one or more second switching mechanisms M 2 . The switching mechanisms M 1 and M 2 may each include, for example, N-type power MOSFETs (metal oxide semiconductor field-effect transistor), and/or other switching mechanisms. Although single switching mechanisms M 1 and M 2 is illustrated for the ease of understanding, it will be appreciated that a plurality of switching mechanisms M 1 and M 2 may be connected in parallel to increase current capacity, decrease conduction losses, and the like.

The controller 105 is configured to apply one or more control signals to the switching mechanisms M 1 and M 2 . For example, the controller 105 may be configured to generate pulse width modulation (PWM) signals or pulse frequency modulation (PFM) signals, a combination of PWM and PFM, and/or different control signals to selectively enable the switching mechanisms M 1 and M 2 according to a duty factor. Regardless of the specific configuration, the controller 105 is configured to provide control signals such that the switching mechanisms M 1 and M 2 are not concurrently enabled (i.e., turned on). In other words, only one of switching mechanism M 1 and M 2 is enabled at a time. Enabling switching mechanisms M 1 and M 2 concurrently provides a direct path between the supply of electric power source 108 and ground, thereby potentially damaging the electric power conversion device 120 and/or the load 110 and/or resulting in undesirable high power usage.

In contrast with conventional electric power conversion devices, the electric power conversion device 120 includes the voltage control mechanism in addition to the current control mechanism. The voltage control mechanism is coupled between the current control mechanism (at the downstream end of the inductor L 1 ) and the load 110 and is operable to control the V L . The current control mechanism is configured to generate current I L1 that is “parked” in the inductor L 1 . The voltage control mechanism is operable to control the amount of the inductor current I L1 that is delivered to a capacitor C 1 . As such, the voltage control mechanism comprises one or more switching mechanisms M 3 and one or more switching mechanisms M 4 . The switching mechanisms M 3 and M 4 may each include, for example, N-type planar MOSFETs (metal oxide semiconductor field-effect transistor), and/or other switching mechanisms. Although single switching mechanisms M 3 and M 4 are illustrated for the ease of understanding, it will be appreciated that a plurality of switching mechanisms M 3 and M 4 may be connected in parallel to increase current capacity, decrease conduction losses, and the like.

A conventional electric power conversion device does not include the switching mechanisms M 3 and M 4 , so the inductor L 1 would instead be coupled directly to the capacitor C 1 and the load 110 . Any excess current that flows through the inductor L 1 and is not consumed by the load 110 is accumulated on the capacitor C 1 and any current drawn by the load 110 that exceeds the current provided by the inductor L 1 is sourced by the capacitor C 1 . The inductor L 1 resists changes in current, thereby preventing the stored energy in the inductor L 1 from being released all at once to the load 110 when the current demands of the load 110 increase. This property of inductors, along with the storage capacity of the capacitor C 1 enable V L to be sufficiently stable during steady-state operation (i.e., when the current demand of the load 110 is relatively constant). Nonetheless, there is some “ripple” in V L that depends on the size of the inductor L 1 , the size of the capacitor C 1 , and/or the switching frequency of the controller 105 , among other factors. Generally speaking, as the size of the inductor L 1 increases, the output ripple during steady state operation (i.e., approximately constant current demand at the load 110 ) proportionally decreases. Accordingly, the inductor L 1 may be sized large enough in order to provide a V L that does not fluctuate outside a desired supply voltage range for the load 110 . However, as previously explained, a conventional electric power conversion device is typically unable to respond to changes in the current needs of the load 110 quickly enough. The large inductance of L 1 that is needed to reduce the ripple at V L increases the response time, producing larger voltage deviations when the current demand of the load 110 varies. The voltage control mechanism that is included in the electric power conversion device 120 enables faster response time to changes in current demand of the load 110 without necessitating decreasing the size of the inductor L 1 which may cause the voltage ripple at V L to increase.

In contrast to the switching mechanisms M 1 and M 2 , the voltage across the switching mechanisms M 3 and M 4 may be substantially less than the voltage drop across the inductor L 1 . For example, the voltage supplied at the downstream of the inductor L 1 may be substantially equivalent to the output voltage at the load 110 . Because the switching mechanisms M 3 and M 4 are switching a lower voltage, the switching mechanisms M 3 and M 4 may be constructed from lower-voltage devices, such as “planar” MOS transistors, as compared to the switching mechanisms M 1 and M 2 . Lower-voltage devices can typically be switched at higher frequencies compared with higher-voltage devices, such as power MOSFETs. Therefore, power loss due to switching is reduced for the switching mechanisms M 3 and M 4 compared with the switching mechanisms M 1 and M 2 . Hence switching mechanisms M 3 and M 4 can be switched at a substantially higher frequency than switching mechanisms M 1 and M 2 .

Switching mechanisms M 3 and M 4 may be incorporated into an integrated circuit, thereby potentially reducing space used and/or reducing cost compared with using discrete components. For example, the switching mechanisms M 3 and M 4 may be realized on the same integrated circuit as the load 110 , may be integrated on a separate die on the same package as the load 110 , or may be integrated on a separate package. The switching mechanisms M 3 and M 4 may be realized as standard-voltage “core” transistors in a typical digital integrated-circuit process, or the switching mechanisms M 3 and M 4 may be realized as higher-voltage thick-oxide input-output transistors in a typical integrated-circuit process. In a preferred embodiment, the switching mechanism M 4 is a P-type planar MOSFET and the switching mechanism M 3 is an N-type planar MOSFET. However, one of ordinary skill in the art will understand that either type of MOSFET may be used for either switching mechanism with appropriate gate-drive circuitry without departing from the scope of the present disclosure.

The controller 105 may be further configured to apply one or more control signals to the voltage control mechanism. For example, the controller 105 may be configured to provide control signals to the switching mechanisms M 3 and M 4 . As with the control signals provided to the switching mechanisms M 1 and M 2 , the control signals that are provided to the switching mechanisms M 3 and M 4 may utilize PWM, PFM, bang-bang control, and/or any other suitable control schema in order to selectively enable the switching mechanism M 3 or the switching mechanism M 4 . In some embodiments the control signals coupled to the switching mechanisms M 3 and M 4 may be at least partially synchronous with the control signals coupled to the switching mechanisms M 1 and M 2 . In other embodiments, the control signals coupled to the switching mechanisms M 3 and M 4 may be asynchronous with the control signals coupled to the switching mechanisms M 1 and M 2 . Furthermore, the control signals coupled to the switching mechanisms M 3 and M 4 may be provided at a different frequency than the control signals that are coupled to the switching mechanisms M 1 and M 2 .

Regardless of the specific configuration of the control signals that are coupled to the switching mechanisms M 3 and M 4 , the controller 105 may be configured to selectively enable the switching mechanism M 3 and disable the switching mechanism M 4 to disable the flow of current I L1 to the load 110 . Specifically, by enabling the switching mechanism M 3 and disabling the switching mechanism M 4 , the instantaneous inductor current I L1 flowing through the inductor L 1 is diverted through the switching mechanism M 3 to ground instead of being delivered to the capacitor C 1 . Conversely, by enabling the switching mechanism M 4 and disabling the switching mechanism M 3 , substantially all of the instantaneous inductor current I L1 flowing through the inductor L 1 (less transistor conduction losses, inductor winding resistance, and the like) is provided to the capacitor C 1 .

The controller 105 may use PWM or PFM to switch the voltage control mechanism or may use a bang-bang technique. In either case, the duty factor (DE) determines the portion of the inductor current I L1 that on average is supplied to the capacitor C 1 . The duty factor may range from 0-100%, where 0% corresponds to the state where the switching mechanism M 4 is disabled (i.e., turned off) and the switching mechanism M 3 is enabled and a 100% corresponds to the state where the switching mechanism M 4 is enabled and the switching mechanism M 3 is disabled. Changing the duty factor thereby changes the charge/discharge timing of the capacitor C 1 —a higher duty factor increases the current flow to the capacitor C 1 and the load 110 .

The capacitor C 1 smoothes the square wave supply current provided through the switching mechanism M 4 to generate I Load that is provided to the load 110 . The I Load is provided to the load 110 according to the duty factor and the inductor current I L1 , as follows: I Load =DF×I L1 . As with the switching mechanisms M 1 and M 2 , control signals are provided to the switching mechanisms M 3 and M 4 such that the switching mechanisms M 3 and M 4 are not concurrently enabled to avoid providing a direct path between the load 110 and ground (i.e., short circuit across the capacitor C 1 ).

During steady-state operation, the switching mechanism M 3 is disabled and the switching mechanism M 4 is enabled, such that substantially all of the inductor current I L1 is provided to the load 110 as I Load . The switching mechanisms M 1 and M 2 are selectively enabled (“switched”) in order to control the inductor current I L1 , thereby controlling V L . In this manner, if the voltage provided to the load 110 (V L ) is constant, the current provided through the switching mechanism M 4 is substantially equivalent to the inductor current I L1 .

In sum, the current control mechanism is configured to generate current I L1 that is parked in the inductor L 1 and metered out to the load 110 by the voltage control mechanism. Because the voltage levels applied to the switching mechanisms M 3 and M 4 are low (i.e., the supply voltage of the load 110 ), the switching mechanisms M 3 and M 4 may be implemented as fast, inexpensive planar transistors and can be operated at very high frequency (e.g., 300 MHz) allowing very rapid response to current transients at the load 110 . When the current demand at the load 110 changes (i.e., not steady-state operation), the switching mechanisms M 3 and M 4 of the voltage control mechanism may be controlled to quickly respond to the change in current demand by increasing or decreasing the amount of the current I L1 this is metered out to the load 110 . In general, the switching frequency of the current control mechanism is slower than the switching frequency of the voltage control mechanism due to the different types of switching mechanisms that are used.

A lumped element CP in the electric power conversion device 120 represents the parasitic capacitance on the downstream side of the inductor L 1 . Each time the switching mechanisms M 3 and M 4 are switched, the parasitic capacitance CP is charged to the load voltage V L (when the switching mechanism M 4 is enabled) and then discharged to ground (when the switching mechanism M 3 is enabled). Thus, each switching cycle of the switching mechanisms M 3 and M 4 an energy E P of

E P =( CP ) V L 2

is dissipated by charging and discharging the parasitic capacitance CP.

In a typical embodiment of the electric power conversion device 120 , the inductor L 1 is a surface mount 0.5 uH 30 A inductor, the switching mechanisms M 3 and M 4 are located on the package, and the capacitor C 1 is on-chip and on-package bypass capacitance. Capacitor CP includes the capacitance of the vias, board traces, and package traces between the inductor L 1 and the switching mechanisms M 3 and M 4 . In a typical application the capacitance CP may total as much as 500 pF. CP=500 pF and V L =1V, then E P is 500 pJ. At a switching frequency of 300 MHz, 150 mW is dissipated charging and discharging CP. When the current control mechanism and the voltage control mechanism of the electric power conversion device 120 are configured as one of a plurality of phases of a regulator, E P is scaled by the number of phases for the total energy that is dissipated due to the cumulative parasitic capacitance.

This switching power increases as the switching frequency (f s ) of the switching mechanisms M 3 and M 4 is increased. One would like to switch the switching mechanisms M 3 and M 4 at a high frequency to minimize the required size of C 1 that is given by

C 1 = I L ⁢ ⁢ 1 ⁡ ( 1 - DF ) f S ⁢ V R

where DF is the duty factor of the switching mechanism M 4 and V R is a ripple voltage of V L .

For example with a phase current of 30 A, a frequency of 300 MHz, and a ripple voltage of 20 mV the required capacitance C 1 is 5 uF per phase. C 1 is typically distributed across many smaller capacitors on the package to give low series inductance and to provide a flat impedance as a function of the switching frequency. Increasing the switching frequency reduces the required size of C 1 but at the expense of increased switching power E P .

An advantage of a current-parking switching regulator, such as the electric power conversion device 120 , is that C 1 is the only filter capacitance needed. In comparison, a conventional electric power conversion device that does not include the switching mechanisms M 3 and M 4 relies on a large (hundreds of μF) filter capacitance to filter the low frequency (typically 300 kHz) ripple.

The configuration of the electric power source 108 , the controller 105 , the switching devices M 1 and M 2 , and the inductor L 1 shown in FIG. 1A is typically referred to as a “buck” converter. While the electric power conversion device 120 is described in the context of this buck converter, one of ordinary skill in the art will understand that the techniques described to regulate a voltage provided to the load 110 can be applied to other “switch-mode” power conversion circuits including, but not limited to, a forward converter, a half-bridge converter, a full-bridge converter, a flyback converter, and/or variants thereof.

FIG. 1B illustrates a multi-phase switching regulator 150 that includes electric power conversion devices 120 , in accordance with one embodiment. Each of the electric power conversion devices 120 is one phase of a six-phase switching regulator. Each electric power conversion device 120 is configured to provide a desired output voltage level (V L ) at the load 110 by converting power received from an electric power source 108 for one phase of the six phases. A single controller may be used to control each of the electric power conversion devices 120 or each electric power conversion device 120 may include a dedicated controller 105 (as shown in FIG. 1B ). A single filter capacitor C 1 may be shared by the different electric power conversion devices 120 rather than including a filter capacitor C 1 in each of the electric power conversion devices 120 . Additionally, one or more of the electric power conversion devices 120 may be replaced with a current-parking switching regulator with a split inductor or a conventional electric power conversion device.

FIG. 1C illustrates an electric power conversion system 160 including an electric power conversion device 180 that is implemented as a current-parking switching regulator with a split inductor, in accordance with one embodiment. Compared with the electric power conversion device 120 shown in FIG. 1A , the electric power conversion device 180 includes a first inductor L 11 that is coupled in series with a second inductor L 2 to form a split inductor. Splitting the inductor reduces losses due to parasitic capacitance CPA on the downstream side of the first inductor L 11 .

The electric power conversion device 180 may be one phase of a multi-phase switching regulator. The electric power conversion device 180 is configured to provide a desired output voltage level (V L ) at the load 170 by converting power received from an electric power source 108 . The electric power conversion device 180 includes a current control mechanism and a voltage control mechanism. The current control mechanism is coupled to the electric power source 108 and the controller 165 may be configured to generate control signals in the same manner as the controller 105 and is operable to control the average of the current I L11 flowing through the inductor L 11 and the current I L2 flowing through the inductor L 2 . For example, as illustrated, the switching mechanisms and M 12 are configured and operable in the same manner as previously described switching mechanisms M 1 and M 2 , respectively. Similarly, the switching mechanisms M 13 and M 14 are configured and operable in the same manner as previously described switching mechanisms M 3 and M 4 , respectively. The capacitor C 11 performs substantially the same function as the capacitor C 1 .

Using two different inductors L 11 and L 2 to form a split inductor reduces the switching energy so that the bulk of the parasitic capacitance falls between the inductor L 11 and L 2 , shown as a first parasitic capacitance CPA. In one embodiment, L 11 is a 0.5 μH 30 A first inductor on a printed circuit board (e.g., a discrete component) and the second inductor L 2 is a 1 nH inductor in the package that enclosed the load 170 . The first parasitic capacitance CPA includes the capacitance of the vias, board traces, and package traces between the first inductor L 11 and the second inductor L 2 . The first parasitic capacitance CPA may be approximately 490 pF. The second parasitic capacitance CPB consists primarily of the drain capacitance of the switching mechanisms M 13 and M 14 and may be approximately 10 pF. If CPB=10 pF and V L =1V, then E P is 1.0 pJ and, at a switching frequency of 500 MHz, 5 mW is dissipated charging and discharging CPB.

The switching frequency of 500 MHz, allows use of a 0.5 μF capacitor (implemented as a distributed array of smaller capacitors in some embodiments) for the capacitor C 11 . The 1 nH inductance of the second inductor L 2 can be formed by integrating a ferrite bead around the traces or bumps carrying the current I L2 or by simply running a trace a suitable distance from the ground return (making the second inductor L 2 a planar air-core inductor). The resonant frequency of the tank circuit formed by L 2 and the first parasitic capacitance CPA is f r =230 MHz. Thus, as long as the switching frequency of the switching mechanisms M 13 and M 14 is high compared to f r , the capacitance of the first parasitic capacitance CPA is effectively isolated from the switching node V L . Because the first parasitic capacitance CPA is located between the first inductor L 11 and the second inductor L 2 , CPA is isolated and is lossless. Any excess current is stored in the split inductor formed by the first inductor L 11 and the second inductor L 2 .

FIG. 2 illustrates a flowchart 200 of a method for regulating the voltage level provided to the load 110 or 170 , in accordance with one embodiment. At step 205 , a current source is configured to generate a current. At step 210 , a voltage control mechanism is configured to provide a portion of the current to the load to regulate the voltage level at the load. At step 215 , if the voltage level at the load 110 or 170 is not greater than a maximum voltage level, the regulation of the voltage level provided to the load is complete. Otherwise, at step 220 , the voltage control mechanism is switched to decouple the current source from the load 110 or 170 and to couple the current source to a current sink to reduce the voltage level at the load. In one embodiment, the current source is at least the inductor L 1 or L 11 and the current sink is ground.

More illustrative information will now be set forth regarding various optional architectures and features with which the foregoing framework may or may not be implemented, per the desires of the user. It should be strongly noted that the following information is set forth for illustrative purposes and should not be construed as limiting in any manner. Any of the following features may be optionally incorporated with or without the exclusion of other features described.

FIG. 3A illustrates a downstream controller 310 of a current-parking switching regulator within an electric power conversion system 300 , in accordance with one embodiment the downstream controller 310 is a bang-bang control circuit that is configured to hold the voltage level at the load 110 , V L , within a specified voltage range between a minimum voltage level (Vmin) and a maximum voltage level (Vmax). For example, when the nominal voltage level of V L is 1 volt and a 20 mV ripple is specified, Vmin is specified as 0.99V and Vmax is specified as 1.01V.

As shown, in FIG. 3A , in one embodiment the downstream controller 310 includes a voltage divider comprising the resistors R 1 and R 2 that are configured to produce a scaled version of V L , scaled voltage V LS . The downstream controller 310 also includes the comparators 312 and 314 , a set-reset flip-flop 315 , and a pre-driver sub-circuit 320 . The scaled voltage V LS =KV L and is within the common-mode range of the comparators 312 and 314 . In one embodiment K=0.5 so that V LS =0.5V L . The comparator 312 compares V LS to a scaled version of Vmax, Vmax_s=KVmax and the comparator 314 compares V LS to a scaled version of Vmin, Vmin_s=KVmin. When V LS is less than Vmin_s, the output of the comparator 314 goes high (TRUE) and the S (set) input to the set-reset flip-flop 315 is asserted, so the Q output that generates the D signal is high (TRUE). When V LS is greater than Vmax_s, the output of the comparator 312 goes high (TRUE) and the R (reset) input to the set-reset flip-flop 315 is asserted, so the Q output that generates the D signal is low (FALSE).

The pre-driver sub-circuit 320 is configured to generate signals coupled to the gates of the switching mechanisms M 3 and M 4 that enable and disable the switching mechanisms M 3 and M 4 . When D is high, the switching mechanism M 4 is enabled and the switching mechanism M 3 is disabled. When D is low, the switching mechanism M 13 is enabled and the switching mechanism M 4 is disabled. When the switching mechanism M 3 is enabled, the current source inductor L 1 ) is coupled to the current sink (i.e., ground) and when the switching mechanism M 3 is disabled the current source is decoupled or isolated from the current sink. When the switching mechanism M 4 is enabled, the current source is coupled to the load 110 and when the switching mechanism M 4 is disabled the current source is decoupled or isolated from the load 110 .

As shown in FIG. 3A , in one embodiment the switching mechanism M 3 is a N-type planar MOS transistor and the switching mechanism M 4 is a P-type planar MOS transistor. The signals generated by the pre-driver sub-circuit 320 are configured to prevent overlap current and overvoltage on the drains of the switching mechanisms M 3 and M 4 . Specifically, only one of the switching mechanisms M 3 and M 4 is enabled at a time.

The switching mechanism M 3 is disabled before the switching mechanism M 4 is enabled to ensure a “dead-time” when both switching mechanisms M 3 and M 4 are disabled. The parasitic capacitance of the drains of the switching mechanisms M 3 and M 4 is charged by the current I L1 during the dead-time and the switching mechanism M 4 is enabled when the voltage across the parasitic capacitance reaches V L , so that current does not flow from the load 110 to the inductor L 1 . The dead-time between when the switching mechanism M 3 is disabled and the switching mechanism M 4 is enabled is controlled to allow the inductor L 1 to charge the drain of the switching mechanism M 4 to V L before the switching mechanism M 4 is enabled. The dead-time also ensures that switching mechanism M 3 is disabled when the switching mechanism M 4 is enabled to avoid shoot-through current from the load 110 through the switching mechanisms M 4 and M 3 to ground.

Similarly the dead-time between when the switching mechanism M 4 is disabled and the switching mechanism M 3 is enabled is controlled to keep the drain of the switching mechanism M 4 from being charged too high by I L1 before the switching mechanism M 3 is enabled. The dead-time between when the switching mechanism M 4 is disabled and the switching mechanism M 3 is enabled also ensures that switching mechanism M 4 is disabled when the switching mechanism M 3 is enabled to avoid shoot-through current from the load 110 through the switching mechanisms M 4 and M 3 to ground.

When power is initially applied to the power conversion system 300 V L is zero and many nanoseconds are needed to charge the capacitor C 1 to a voltage between Vmin and Vmax. The downstream controller circuit 310 may be configured to operate using an auxiliary supply voltage V ST that is turned on at startup, before the upstream controller 305 begins to generate the current I L1 . The switching mechanism M 4 is enabled to charge C 1 when the current source starts up. In one embodiment, the auxiliary supply voltage is not used for the downstream controller 310 and the downstream controller 310 is configured to disable the switching mechanism M 3 and enable the switching mechanism M 4 until V L reaches Vmax.

FIG. 3B illustrates waveforms 330 corresponding to the current-parking switching regulator using the downstream controller 310 shown in FIG. 3A , in accordance with one embodiment. The top waveform is the signal D, the output of the set-reset flip-flop 415 shown in FIG. 3A , and the bottom waveform is V L . The bottom waveform is a saw-tooth bounded between Vmin (0.99V) and Vmax (1.01V). The slope of the downward ramps of V L is given by I L /C 1 , while the slope of the upward ramps of V L is given by (I L1 −I L )/C 1 , where I L is the current provided to the load 110 . The frequency of the saw-tooth varies as the current demand of the load 110 varies and as the current I L1 varies.

The frequency is at a maximum when the portion of the current I L1 that is provided to the load 110 (i.e., the load current I L ) is high (corresponding to a steep downward slope of V L ), but when I L is also a small fraction of I L1 (corresponding to a steep upward slope). The frequency is low, resulting in lower power loss due to switching, when I L is low and when I L is also a high fraction of I L1 .

The waveforms 330 correspond to a current I L1 of 30 A and a current I L of 20 A for the first 10 ns. At 10 ns the load current I L is abruptly changed from 20 A to 25 A. During the first 10 ns the downstream controller 310 generates the signal D having a period of approximately 3 ns with D asserted for 2 ns and negated for 1 ns. After the first 10 ns, the period increases to approximately 5 ns with D asserted for 4 ns and negated for 1 ns.

FIG. 3C illustrates another a downstream controller 360 of a current-parking switching regulator within an electric power conversion system 350 , in accordance with one embodiment. Like the downstream controller 310 , the downstream controller 360 is a bang-bang control circuit that is configured to hold the voltage level at the load 110 , V L , within a specified voltage range between Vmin and Vmax.

As shown, in FIG. 3C , in one embodiment the downstream controller 360 includes a voltage divider comprising the resistors R 1 and R 2 that are configured to produce a scaled version of V L , scaled voltage V LS . The downstream controller 360 includes a single comparator 362 and the pre-driver sub-circuit 320 . A scaled reference voltage, Vref_s, is centered within a range bounded by Vmin_s and Vmax_s. The downstream controller 360 includes a feedback circuit comprising the resistors 143 and R 4 that are configured to provide positive feedback to the comparator 362 by coupling the output D to the feedback circuit to vary the input. V F , to the comparator 362 . The comparator 362 compares V LS to V F , producing hysteresis in the signal D. When V LS is less than V F , signal D, the output of the comparator 362 , goes high (TRUE). When V LS is greater than V F , signal D, the output of the comparator 362 , goes low (FALSE). The voltage generated at the non-inverting input of the comparator 362 varies in response to changes in the voltage level of D. The resistors R 3 and R 4 set the hysteresis of the comparator 362 so that the voltage at the positive input of the comparator 362 (i.e., V F ) is equal to Vmin_s when D is low and equal to Vmax_s when D is high. Thus, R 3 and R 4 set the ripple of the voltage control mechanism.

FIG. 4A illustrates yet another downstream controller 410 of a current-parking switching regulator within an electric power conversion system 400 , in accordance with one embodiment. The downstream controller 410 is configured to control the voltage control mechanism using pulse width modulation (PWM). As shown in FIG. 4A , in one embodiment, the downstream controller 410 includes the voltage divider comprising the resistors R 1 and R 2 that are configured to produce a scaled version of V L , scaled voltage V LS . The downstream controller 410 also includes a single comparator 412 , a set-reset flip-flop 415 , and the pre-driver sub-circuit 320 . The scaled voltage V LS =KV L and is within the common-mode range of the comparator 412 . The comparator 412 compares V LS to a scaled version of Vmax, Vmax_s=KVmax. When V LS is less than Vmax_s, the output of the comparator 412 is low and the R (reset) input to the set-reset flip-flop 415 is negated, such that the Q output is dependent on the S input. When V LS is greater than Vmax_s, the output of the comparator 412 goes high and the R input to the set-reset flip-flop 415 is asserted, so the Q output is low (FALSE). The R input has priority over the S input (i.e., when both R and S are asserted Q is low).

The S input of the set-reset flip-flop 415 is coupled to an oscillator 425 that produces a sequence of narrow pulses at a constant frequency. Each of the pulses asserts the S input of the set-reset flip-flop 415 , causing the D signal to go high as long as the R input of the set-reset flip-flop 415 is not asserted. When the D signal is high, the switching mechanism M 4 is enabled and the switching mechanism M 3 is disabled. When the D signal is low, the switching mechanism M 3 is enabled and the switching mechanism M 4 is disabled.

When V LS is greater than Vmax_s, the comparator 412 asserts the R input of the set-reset flip-flop 415 , disabling the switching mechanism M 4 and enabling the switching mechanism M 3 to reduce the voltage level of V L . The frequency of the oscillator 425 should be selected to keep the voltage ripple of V L within the range bounded be Vmin and Vmax under worst-case operating conditions. Compared with the downstream controllers 310 and 360 , the downstream controller 410 may have higher power loss due to switching because the downstream controller 410 operates at a high frequency than necessary when V L is within the voltage range bounded by Vmin and Vmax.

FIG. 4B illustrates waveforms 450 corresponding to the current-parking switching regulator using the downstream controller 410 shown in FIG. 4A , in accordance with one embodiment. The frequency of the oscillator 425 is 500 MHz (i.e., 2 ns period). The top waveform is the signal D, the output of the set-reset flip-flop 415 shown in FIG. 4A , and the bottom waveform is V L . The waveforms 450 correspond to a current I L1 of 30 A and a current I L of 20 A for the first 10 ns. At 10 ns the load current I L is abruptly changed from 20 A to 25 A. During the first 10 ns the downstream controller 410 generates the signal D, transitioning (or keeping) the signal D low whenever V L is greater than Vmax.

After the first 10 ns, the magnitude of the voltage ripple of V L changes (i.e., increases) while the frequency of the saw-tooth waveform of V L does not necessarily change. Because the rising slope of V L reaches Vmax with differing amounts of time before the next pulse of the oscillator 425 , the minimum voltage reached by V L varies. In the worst case, the entire 2 ns period of the oscillator 425 occurs during a falling slope of V L producing a large voltage ripple as shown following 18 ns. The downstream controller 410 may produce a higher voltage ripple and have higher power loss due to switching compared with the downstream controllers 310 and 360 .

FIG. 5A illustrates another flowchart 500 of a method for regulating the voltage level provided to the load 110 or 170 using the downstream controller 310 , 360 , or 410 (or the controller 105 or 165 that includes a downstream controller), in accordance with one embodiment. At step 502 , the switching mechanisms in the voltage control mechanism are initialized. Specifically, the switching mechanism M 3 is disabled and the switching mechanism M 4 is enabled. In another embodiment, the auxiliary supply voltage V ST is provided to the downstream controller, and step 502 is replaced with a step that provides the auxiliary supply to the downstream controller before the electric power 108 source provides a supply voltage to the current source.

At step 505 , the upstream controller 305 (or controller 105 or 165 ) configures the current control mechanism to generate the current I L1 through the inductor L 1 . The current control mechanism may be configured to provide a current that is greater than an average current needed by the load 110 or 170 . At step 510 , the downstream controller configures the voltage control mechanism to provide a portion of the current to the load 110 or 170 to regulate the voltage level, V L at the load 110 . At step 515 , the downstream controller determines if V L is greater than Vmax, and, if so, then at step 525 , the voltage control mechanism is switched to decouple the current source from the load 110 or 170 and to couple the current source (i.e., the inductor L 1 ) to a current sink (i.e., ground) to provide a lesser portion of I L1 to the load 110 or 170 to reduce V L . After step 525 , the downstream controller returns to step 515 .

If, at step 515 , V L is not greater than Vmax, then at step 530 , the downstream controller determines if V L is less than Vmin. If V L is less than Vmin, then at step 535 the downstream controller configures the voltage control mechanism to couple the current source to the load 110 or 170 and to decouple the current source from the current sink to provide a greater portion of I L1 to the load 110 or 170 to increase V L . After step 535 , the downstream controller returns to step 515 . Otherwise, when V L is not greater than Vmax and is not less than Vmin (i.e., V L is within the range bounded by Vmin and Vmax), then the downstream controller returns to step 515 .

The portion of the current that is provided to the load 110 or 170 is determined by the signal D that is generated by the downstream controller 310 , 360 , or 410 . The signal D that alternately enables the first switching mechanism M 4 allowing a portion of the current to flow to the load 110 or 170 while disabling a second switching mechanism M 3 and then enables the second switching mechanism to pull the inductor L 1 (or L 2 for the electric power conversion device 180 ) to ground while disabling the second switching mechanism to isolate the load 110 or 170 from the inductor in response to a current transient, the downstream controller 310 , 360 , or 410 quickly increases or decreases the portion of the current that is provided to the load 110 or 170 and maintains the voltage level V L within a predetermined range bounded by Vmin and Vmax. Specifically, when V L is greater than Vmax, D goes low to divert current away from load 170 and when V L is less than Vmin, D goes high to source current to the load 170 .

FIG. 5B illustrates a system 550 including a current-parking switching regulator driving two loads, loads 110 -A and 110 -B, with independently controller voltages V LA and V LB using a shared current source, according to one embodiment. As shown in FIG. 5B , the single inductor L 1 provides the current I L1 , a portion of which is provided to each of the two loads 110 -A and 110 -B. In one embodiment, the portions of the current I L1 are not provided to the two loads 110 -A and 110 -B simultaneously. A filter capacitor C 1 A is coupled to the load 110 -A and a filter capacitor C 1 B is coupled to the load 110 -B. A separate downstream controller 510 -A and 510 -B and respective switching mechanism M 4 A and M 4 B is associated with each of the loads. However, the switching mechanism M 3 may be shared by the downstream controllers 510 -A and 510 -B. The downstream controllers 510 -A and 510 -B may each be one of the downstream controllers 310 , 360 , and 410 .

The D output signals, DA and DB that are generated by the downstream controllers 510 -A and 510 -B are combined to give operating priority to the load 110 -A. In one embodiment, load 110 -A is the most critical or highest-current load. When DA is high, the switching mechanism M 4 A is enabled, coupling the current source to the load 110 -A and the switching mechanisms M 4 B and M 3 are both disabled. When the switching mechanism M 4 A is enabled the current I L1 is provided to capacitor C 1 A and V LA ramps up while V LB ramps down. When DA is low DB may be high, enabling the switching mechanism M 4 B and disabling the switching mechanisms M 4 A and M 3 . When the switching mechanism M 4 B is enabled current I L1 is provided to capacitor C 1 B and V LB ramps up while V LA ramps down. When DA and DB are both low, the switching mechanism M 3 is enabled and both switching mechanisms M 4 A and M 4 B are disabled causing the current. L L1 to be “parked” in the inductor L 1 while both V LA and V LB ramp down. The effective duty factor of the system 550 is the duty factor of the signal formed by the logical “OR” of DA and DB, shown in FIG. 5B as the signal D.

FIG. 5C illustrates a system 560 including a current-parking switching regulator, according to one embodiment. The current-parking switching regulator in the system 560 may be one of the electric power conversion devices 120 and 180 shown in FIGS. 1A and 1C , respectively, or one of the current-parking switching regulators shown in FIGS. 3A, 3C, and 4A .

The electric power source 108 is coupled to the current control mechanism and the voltage control mechanism of the current-parking switching regulator with the inductor L 1 an alternate embodiment, the electric power source 108 is coupled to the current control mechanism and the voltage control mechanism of the current-parking switching regulator with the inductors L 11 and L 2 . The upstream controller 305 is configured to generate a current through the inductor L 1 . The downstream controller 510 may be one of the downstream controllers 310 , 360 , and 410 and is configured to regulate the voltage level at the load, i.e., circuit 580 . In one embodiment, the downstream controller 510 is configured to maintain the voltage level at the circuit 580 within a predetermined range bounded by Vmin and Vmax.

The inductor L 1 is positioned outside of a package 570 that encloses the circuit 580 . A second inductor L 2 (not shown) may be positioned inside of the package 570 , reducing the second parasitic capacitance CPB compared with the first parasitic capacitance CPA, as described in conjunction with FIG. 1C . The second inductor L 2 , the switching mechanisms M 3 and 414 (or M 13 and M 14 ), and the capacitor C 1 (or C 11 ) may be fabricated as part of the die 575 that includes the circuit 580 . In one embodiment, the second inductor L 2 a planar air-core inductor and the switching mechanisms M 3 and M 4 (or M 13 and M 14 ) are planar MOS transistors. Although a single phase of the current-parking switching regulator with a split inductor is shown in FIG. 5B , multiple phases of the current-parking switching regulator with a split inductor or a combination of one or more current-parking switching regulators (with or without a split inductor) may be used with one or more conventional electric power conversion devices to provide power to the circuit 580 .

FIG. 6 illustrates an exemplary system 600 in which the various architecture and/or functionality of the various previous embodiments may be implemented. As shown, a system 600 is provided including at least one central processor 601 that is connected to a communication bus 602 . The communication bus 602 may be implemented using any suitable protocol, such as PCI (Peripheral Component Interconnect), PCI-Express, AGP (Accelerated Graphics Port), HyperTransport, or any other bus or point-to-point communication protocol(s). The system 600 also includes a main memory 604 . Control logic (software) and data are stored in the main memory 604 which may take the form of random access memory (RAM).

The system 600 also includes input devices 612 , a graphics processor 606 , and a display 608 , i.e., a conventional CRT (cathode ray tube), LCD (liquid crystal display), LED (light emitting diode), plasma display or the like user input may be received from the input devices 612 , e.g., keyboard, mouse, touchpad, microphone, and the like. In one embodiment, the graphics processor 606 may include a plurality of shader modules, a rasterization module, etc. Each of the foregoing modules may even be situated on a single semiconductor platform to form a graphics processing unit (GPU).

In the present description, a single semiconductor platform may refer to a sole unitary semiconductor-based integrated circuit or chip. It should be noted that the term single semiconductor platform may also refer to multi-chip modules with increased connectivity which simulate on-chip operation, and make substantial improvements over utilizing a conventional central processing unit (CPU) and bus implementation. Of course, the various modules may also be situated separately or in various combinations of semiconductor platforms per the desires of the user. One or more of the systems 550 and 500 shown in FIGS. 5A and 5B , respectively, may be incorporated in the system 600 to provide power to one or more of the chips.

The system 600 may also include a secondary storage 610 . The secondary storage 610 includes, for example, a hard disk drive and/or a removable storage drive, representing a floppy disk drive, a magnetic tape drive, a compact disk drive, digital versatile disk (DVD) drive, recording device, universal serial bus (USB) flash memory. The removable storage drive reads from and/or writes to a removable storage unit in a well-known manner. Computer programs, or computer control logic algorithms, may be stored in the main memory 604 and/or the secondary storage 610 . Such computer programs, when executed, enable the system 600 to perform various functions. The main memory 604 , the storage 610 , and/or any other storage are possible examples of computer-readable media.

In one embodiment, the architecture and/or functionality of the various previous figures may be implemented in the context of the central processor 601 , the graphics processor 606 , an integrated circuit (not shown) that is capable of at least a portion of the capabilities of both the central processor 601 and the graphics processor 606 , a chipset (i.e., a group of integrated circuits designed to work and sold as a unit for performing related functions, etc.), and/or any other integrated circuit for that matter.

Still yet, the architecture and/or functionality of the various previous figures may be implemented in the context of a general computer system, a circuit board system, a game console system dedicated for entertainment purposes, an application-specific system, and/or any other desired system. For example, the system 600 may take the form of a desktop computer, laptop computer, server, workstation, game consoles, embedded system, and/or any other type of logic. Still yet, the system 600 may take the form of various other devices including, but not limited to a personal digital assistant (PDA) device, a mobile phone device, a television, etc.

Further, while not shown, the system 600 may be coupled to a network (e.g., a telecommunications network, local area network (LAN), wireless network, wide area network (WAN) such as the Internet, peer-to-peer network, cable network, or the like) for communication purposes.

While various embodiments have been described above, it should be understood that they have been presented by way of example only, and not limitation. Thus, the breadth and scope of a preferred embodiment should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.