High-speed comparator

1. Field of the Invention

The present invention generally relates to comparators. More specifically, the present invention improves the bit-error rate of high-speed comparators that operate at low supply voltages.

2. Related Art

A comparator is designed to compare an input signal to a known reference level. The input signal can be an input voltage or an input current. Correspondingly, the known reference level can be a known voltage reference level or a known current reference level. Typically, the comparator is designed to output a logic “1” at the end of a clock cycle, when the input signal exceeds the known reference level, and to output a logic “0” at the end of the clock cycle, when the input signal is below the known reference signal. Alternatively, the comparator can be designed to operate in a converse manner. That is, the comparator can output a logic “0” at the end of the clock cycle, when the input signal exceeds the known reference level, and to output a logic “1” at the end of a clock cycle when the input signal is below the known reference level.

Comparators are basic building blocks of an Analog-to-Digital Converter (ADC). Transistors arranged to provide positive feedback are typically used to implement the comparator. Some ADC architectures, such as flash, folding and subranging ADCs, require a large number of comparators. The large number of comparators used in these ADC architectures drives the need to make the comparators capable of operating with low power supply voltages. A low power comparator reduces overall power consumption and therefore allows an ADC architecture to incorporate more comparators into its design.

Comparators are often required to operate using small input signals. Typically, the comparator can generate an output (i.e., a logic “1” or a logic “0”) more quickly when provided with a large input signal. Consequently, with a small input signal, the comparator needs more time to generate the output. A bit-error may result if the comparator does not generate the output by the end of the clock cycle. With conventional ADC architectures, clock cycles are becoming shorter and input signals are becoming smaller. Accordingly, comparators that operate at high speeds, from low supply voltages, and with low bit-error rates (BERs) are desired.

The BER of the comparator strongly depends on the bias currents of the transistors used in the comparator. The bias currents within the comparator are limited by the supply voltage of the comparator. One technique for achieving a low BER without increasing the supply voltage or extending the clock cycle of the comparator involves lowering the threshold voltage of the transistors. Fabrication of low threshold voltage transistors, however, is expensive. Further, the power consumption of an “off” transistor that has a lowered threshold voltage may become high enough to be impractical.

Another technique for achieving a low BER is to implement the comparator using thick-oxide transistors that are operated from an input/output (I/O) supply voltage. The I/O power supply is often significantly higher than the power supply provided to core transistors. Thick-oxide transistors, however, require more area within an ADC, because their minimum length is greater than that of the core transistors. Further, thick-oxide transistors have a lower transconductance for the same bias current, which has a detrimental effect on the BER of the comparator. Finally, because thick-oxide transistors are operated from the higher I/O power supply, the power consumption of the comparator will increase significantly.

Accordingly, the present invention is related to a high-speed comparator that achieves a low BER while operating from low input signals and low supply voltages that substantially obviates one or more of the disadvantages of the related art.

In one aspect, there is provided a differential comparator with improved bit-error rate performance. The differential comparator includes a first pair of transistors receiving a differential input. A second pair of transistors is coupled to the first pair of transistors. A pair of resistive elements is connected between the first pair and the second pair of transistors so as to increase bias currents shared by the first and second pairs of transistors. The increased bias currents reduce a time required by the differential comparator to transition from a meta-stable state to a stable state, thereby improving a bit-error rate of the differential comparator. The resistive elements can use linear resistors or transmission gates. Gates of either the first or second pair of transistors can provide an output.

Additional features and advantages of the invention will be set forth in the description that follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure and particularly pointed out in the written description and claims hereof as well as the appended drawings.

It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings:

FIG. 1 illustrates an exemplary conventional differential comparator.

FIG. 2 illustrates a configuration of differential input current sources of the exemplary conventional differential comparator.

FIG. 3 illustrates a portion of the exemplary conventional differential comparator in a meta-stable state at the start of a latch phase.

FIG. 4 illustrates a technique for increasing a drain current of the circuit of FIG. 3 .

FIG. 5 illustrates a differential comparator of one embodiment of the invention with an improved bit-error rate.

FIG. 6 illustrates another embodiment of the differential comparator of FIG. 5 .

FIG. 7 illustrates an alternative arrangement of the differential comparator of FIG. 5 .

FIG. 8 illustrates another embodiment of the differential comparator of FIG. 7 .

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.

FIG. 1 illustrates an exemplary conventional differential comparator 100 . The differential comparator 100 has an input including a differential-mode signal current I IN and a common-mode bias current I BIAS . A differential input current source 102 a provides an input current equal to I BIAS +I IN /2. A differential input current source 102 b provides a complementary input current equal to I BIAS −I IN /2. Together, nodes 110 a and 110 b provide an output of the differential comparator 100 .

As further shown in FIG. 1 , the differential comparator 100 includes an N-channel type metal oxide semiconductor field effect transistor (NMOSFET) 106 a and an NMOSFET 106 b . A source of the NMOSFET 106 a and a source of the NMOSFET 106 b are connected to a supply voltage V SS . The supply voltage V SS is a relatively low supply voltage. For example, V SS could be a ground or a negative supply voltage. The supply voltage V SS often represents a logic “0.” A gate of the NMOSFET 106 a is connected to a drain of the NMOSFET 106 b . Similarly, a gate of the NMOSFET 106 b is connected to a drain of the NMOSFET 106 a . This cross-attached configuration of the NMOSFETs 106 a and 106 b provides positive feedback between the NMOSFETs 106 a and 106 b . The drain of the NMOSFET 106 a is connected to a drain of an NMOSFET 112 at the node 110 a . The drain of the NMOSFET 106 b is connected to a source of the NMOSFET 112 at the node 110 b . A gate of the NMOSFET 112 is configured to receive a clock signal (clk). The NMOSFET 112 operates as a switch responsive to the clock signal clk.

The differential comparator 100 further includes a P-channel type metal oxide semiconductor field effect transistor (PMOSFET) 118 a and a PMOSFET 118 b . A drain of the PMOSFET 118 a is connected to a gate of the PMOSFET 118 b . Similarly, a drain of the PMOSFET 118 b is connected to a gate of the PMOSFET 118 a . This cross-attached configuration of PMOSFETs 118 a and 118 b provides positive feedback between the PMOSFETs 118 a and 118 b . A source of the PMOSFET 118 a and a source of the PMOSFET 118 b are both connected to a drain of a PMOSFET 122 . A source of the PMOSFET 122 is connected to a supply voltage V DD , which is a relatively high supply voltage. However, in many applications, the voltage supply V DD may not exceed 1 . 2 volts and may be as low as 1 volt. The voltage supply V DD often represents a logic “1.” A gate of the PMOSFET 122 is also configured to receive the clock signal clk. The PMOSFET 112 operates as a switch responsive to the clock signal clk.

Together, the NMOSFET 106 a , the NMOSFET 106 b and the NMOSFET 112 form an NMOS latch 104 . Similarly, the PMOSFET 118 a , the PMOSFET 118 b and the PMOSFET 122 form a PMOS latch 120 . The NMOS latch 104 and the PMOS latch 120 are arranged in a stacked configuration between the supply voltages V SS and V DD . The gate of the PMOSFET 118 b and the drain of the PMOSFET 118 a are connected to the NMOS latch 104 at the node 110 a . The gate of the PMOSFET 118 a and the drain of the PMOSFET 118 b are connected to the NMOS latch 104 at the node 110 b . Effectively, the gates of the NMOSFET 106 a and the NMOSFET 106 b are cross-attached to the drains of the PMOSFET 118 b and the PMOSFET 118 a , respectively. Similarly, the gates of the PMOSFET 118 a and the PMOSFET 118 b are cross-attached to the drains of the NMOSFET 106 b and the NMOSFET 106 a , respectively.

FIG. 2 illustrates a configuration of the differential input current sources 102 a and 102 b in more detail. The differential input current source 102 a includes a PMOSFET 202 a . The differential input current source 102 b includes a PMOSFET 202 b . A source of the PMOSFET 202 a is connected to a source of the PMOSFET 202 b . The sources of the PMOSFETs 202 a and 202 b are connected to a current source 204 . The current source 204 supplies a bias current equal to 2·I BIAS . The current source 204 is connected to the voltage supply V DD . The current source 204 supplies the bias current I BIAS to the sources of the PMOSFETs 202 a and 202 b . A drain of the PMOSFET 202 a is connected to the node 110 a . A drain of the PMOSFET 202 b is connected to the node 110 b.

A gate of the PMOSFET 202 a and a gate of the PMOSFET 202 b are connected to a differential-mode input voltage V IN . The differential-mode input voltage V IN applied to the gate of the PMOSFET 202 a and the gate of the PMOSFET 202 b provides the differential-mode signal current I IN to the node 110 a and the node 110 b . Specifically, the differential-mode input voltage V IN , in conjunction with the current source 204 , provides the input current equal to I BIAS +I IN /2. Similarly, the differential-mode input voltage, in conjunction with the current source 204 , provides the input current equal to I BIAS −I IN /2. The magnitude of the differential-mode signal current I IN is proportional to the magnitude of the differential-mode input voltage V IN applied to the gates of the PMOSFET 202 a and the PMOSFET 202 b.

The differential comparator 100 operates in two distinctive clock phases within one clock cycle. The first clock phase is a reset phase and the second clock phase is a latch phase. During the reset phase, the clock signal clk applied to the gate of the NMOSFET 112 and the PMOSFET 122 is relatively high. The NMOSFET 112 is turned on by the clock signal clk being relatively high. Turning on the NMOSFET 112 results in connecting the node 110 a to the node 110 b . In effect, the gate of the NMOSFET 106 a and the gate of the NMOSFET 106 b are connected together. A voltage at the node 110 a is therefore equal to a voltage at the node 110 b during the reset phase. This operation erases the output of the differential comparator 100 from the previous latch phase.

Also, during the reset phase, the PMOSFET 122 is turned off by the clock signal clk being relatively high. Turning the PMOSFET 112 off ensures that the PMOS latch 120 is disconnected from the voltage supply V DD during the reset phase. Disconnecting the PMOS latch 120 from the voltage supply V DD prevents excessive current flow from the voltage supply V DD to the voltage supply V SS .

The latch phase of the differential comparator 100 begins when the NMOSFET 112 is turned off and the PMOSFET 122 is turned on. Specifically, the latch phase begins when the clock signal clk is relatively low. With the PMOSFET 122 turned on, the PMOS latch 120 is connected to the voltage supply V DD . With the NMOSFET 112 turned off, the gate of the NMOSFET 106 a is no longer connected to the gate of the NMOSFET 106 b.

At the beginning of the latch phase, the differential comparator 100 is in a meta-stable state. The differential comparator 100 uses the positive feedback configuration of the NMOS latch 104 and the PMOS latch 120 to transition into one of two possible stable states during the latch phase. Which stable state the differential comparator 100 switches to is determined by the value of the differential-mode signal current I IN relative to a threshold level (i.e., the known reference level) of the differential comparator 100 .

The differential comparator 100 is assumed to have an ideal threshold level of zero. When the differential-mode signal current I IN is greater than the threshold level, a voltage at the node 110 a increases steadily. As the voltage at the node 110 a increases, a drain current of the NMOSFET 106 b increases and, in turn, a voltage at the node 110 b decreases. As the voltage at the node 110 b decreases, a drain current of the NMOSFET 106 a decreases, which further increases the voltage at the node 110 a . The voltage at the node 110 a will continue to increase until the node 110 a is pulled up to the supply voltage V DD . The voltage at the node 110 b will continue to decrease until the voltage at the node 110 b is pulled down to the supply voltage V SS . The PMOSFETs 118 a and 118 b serve to increase the speed at which the node 110 a is pulled up to the supply voltage V DD .

As a result of the positive feedback of the NMOS latch 104 and of the PMOS latch 120 , a differential-mode signal current I IN greater than the threshold level causes the node 110 a to “clip” to the supply voltage V DD and the node 110 b to “clip” to the supply voltage V SS . A logic “1” and a logic “0” are therefore output at the nodes 110 a and 110 b , respectively, at the end of the latch phase. This output state is one of the two stable states of the differential comparator 100 .

The differential comparator 100 will transition from the meta-stable state to the alternative stable state when the differential-mode signal current I IN is below the threshold level. When the differential-mode signal current I IN is below the threshold level, a voltage at the node 110 b increases steadily. As the voltage at the node 110 b increases, the drain current of the NMOSFET 106 a increases and, in turn, a voltage at the node 110 a decreases. As the voltage at the node 110 a decreases, the drain current of the NMOSFET 106 b decreases, which further increases the voltage at the node 110 b . The voltage at the node 110 b will continue to increase until the node 110 b is pulled up to the supply voltage V DD . The voltage at the node 110 a will continue to decrease until the voltage at the node 110 a is pulled down to the supply voltage V SS . The PMOSFETs 118 a and 118 b serve to increase the speed at which the node 110 b is pulled up to the supply voltage V DD .

As a result of the positive feedback of the NMOS latch 104 and of the PMOS latch 120 , a differential-mode signal current I IN below the threshold level causes the node 110 a to “clip” to the supply voltage V SS and the node 110 b to “clip” to the supply voltage V DD . A logic “0” and a logic “1” are therefore output at the nodes 110 a and 110 b , respectively, at the end of the latch phase. This output state is a second stable state of the differential comparator 100 .

As noted above, the transition of the differential comparator 100 from a meta-stable state to one of the two stable states occurs during the latch phase. Essentially, this requires the differential comparator 100 to reach a stable state in less than one clock cycle (typically, half a clock cycle). Currently, Analog-to-Digital Converters (ADCs) are being designed to operate at clock speeds between 100 MHz and 2 GHz. At a clock speed of 1 GHz, for example, the clock period is 1 ns. Therefore, at a clock speed of 1 GHz, the differential comparator 100 has only 500 ps to transition from the meta-stable state to one of the two stable states.

The speed of the differential comparator 100 is inversely related to the value of the differential-mode signal current I IN . That is, as I IN decreases, the time it takes the differential comparator 100 to complete a transition from the meta-stable state to a stable state increases. If the differential comparator 100 does not reach a stable state by the end of the latch phase, a bit-error can result. Specifically, the differential comparator does not have enough time to provide the proper output for a given differential-mode signal current I IN . In effect, the differential comparator 100 becomes inaccurate when I IN becomes small. Accordingly, the bit-error rate (BER) of the differential comparator will suffer over the course of several clock cycles.

The transition speed of the differential comparator 100 , and consequently the BER of the differential comparator 100 , is strongly dependent upon the shared drain current of the NMOSFET 106 a and the PMOSFET 118 a and the shared drain current of the NMOSFET 106 b and the PMOSFET 118 b at the start of the latch phase. When the differential comparator 100 is in the meta-stable state, a gate-source voltage of the NMOSFET 106 a is equal to a gate-source voltage of the NMOSFET 106 b . Similarly, a source-gate voltage of the PMOSFET 118 a is equal to a source-gate voltage the PMOSFET 118 b . This allows a portion of the differential comparator 100 to be represented as a half circuit. FIG. 3 illustrates a portion of the exemplary conventional differential comparator 100 in the meta-stable state, at the start of the latch phase. Specifically, the NMOSFET 106 a and the NMOSFET 106 b are represented by an NMOSFET 106 . Similarly, the PMOSFET 118 a and the PMOSFET 118 b are represented by a PMOSFET 118 . As shown in FIG. 3 , a source of the PMOSFET 118 is connected to the supply voltage V DD since the PMOSFET 122 (not shown in FIG. 3 ) is turned on during the latch phase.

In FIG. 3 , the on-resistance of the PMOSFET 122 is assumed to be negligible. Further, it is assumed that the differential-mode signal current I IN and the common-mode bias current I BIAS have been disconnected from the differential comparator 100 . FIG. 3 shows a drain current, I BER , shared by the NMOSFET 106 and the PMOSFET 118 . A relationship between the drain current I BER and the supply voltage V DD can be determined from the half circuit representation of the differential comparator 100 illustrated in FIG. 3 . It follows from FIG. 3 that:

V SG,118 ( I BER )+ V GS,106 ( I BER )= V DD ,

where V SS =0 volts, V GS,106 represents the gate-source voltage of the NMOSFET 106 and is a function of I BER , and V SG,118 represents the source-gate voltage of the PMOSFET 118 and is also a function of I BER . This equation shows that the sum of the gate-source voltages of the NMOSFET 106 and the PMOSFET 118 , and therefore the drain current I BER , is limited by the supply voltage V DD .

The drain current I BER depicted in FIG. 3 can be increased by increasing the gate-source voltages of the NMOSFET 106 and the PMOSFET 118 . Increasing the gate-source voltages of the NMOSFET 106 and the PMOSFET 118 typically entails increasing the supply voltage V DD . The overall power consumption of the differential comparator 100 will increase when V DD is increased. Alternatively, the NMOSFET 106 and the PMOSFET 118 can be fabricated to have very low threshold voltages. If the NMOSFET 106 and the PMOSFET 118 have low threshold voltages, then a lower supply voltage V DD can be used generate a desired drain current I BER . Fabrication of low threshold transistors, however, is expensive. Further, the overall power consumption of the differential comparator 100 may also increase from the use of low threshold transistors.

FIG. 4 shows a technique for increasing the drain current I BER of the NMOSFET 106 and the PMOSFET 118 without the need to increase the supply voltage V DD or to change the characteristics of the NMOSFET 106 or the PMOSFET 118 . As shown in FIG. 4 , a resistive element 402 is introduced and the connections between the NMOSFET 106 and the PMOSFET 118 are adjusted. Specifically, the resistive element 402 is placed between the drain of the NMOSFET 106 , at a node 404 , and the drain of the PMOSFET 118 , at a node 406 . Further, the gate of the NMOSFET 106 is connected to the drain of the PMOSFET 118 at the node 406 while the gate of the PMOSFET 118 is connected to the drain of the NMOSFET 106 at the node 404 .

A relationship between the drain current I BER and the supply voltage V DD can be determined from the arrangement of the NMOSFET 106 and the PMOSFET 118 depicted in FIG. 4 . It follows from FIG. 4 that:

V SG,118 ( I BER )+ V GS,106 ( I BER )= V DD +R·I BER ,

where R represents the resistance of the resistive element 402 .

This equation shows that the sum of the gate-source voltages of the NMOSFET 106 and the PMOSFET 118 now exceeds the supply voltage V DD . Therefore, the shared drain current I BER of the NMOSFET 106 and the PMOSFET 118 shown in FIG. 4 can be significantly greater than the shared drain current I BER of the NMOSFET 106 and the PMOSFET 118 shown in FIG. 3 for the same supply voltage V DD .

FIG. 5 illustrates a differential comparator 500 incorporating the resistive element 402 in the manner shown in FIG. 4 . As shown in FIG. 5 , the gate of the NMOSFET 106 b is connected to the drain of the PMOSFET 118 a at a node 406 a . The gate of the NMOSFET 106 a is connected to the drain of the PMOSFET 118 b at a node 406 b . Similarly, the gate of the PMOSFET 118 a is connected to the drain of the NMOSFET 106 b at a node 404 b . The gate of the PMOSFET 118 b is connected to the drain of the NMOSFET 106 a at a node 404 a . A resistive element 402 a is connected between the drain of the PMOSFET 118 a and the drain of the NMOSFET 106 a . Specifically, the resistive element 402 a is connected between the node 404 a and the node 406 a . Likewise, a resistive element 402 b is connected between the drain of the PMOSFET 118 b and the drain of the NMOSFET 106 b . Specifically, the resistive element 402 b is connected between the node 404 b and the node 406 b . The placement of the resistive elements 402 a and 402 b between the connections of the NMOSFETs 106 a and 106 b and the PMOSFETs 118 a and 118 b maintains the positive feedback between the NMOSFETs 106 a and 106 b and the positive feedback between the PMOSFETs 118 a and 118 b.

An output of the differential comparator 500 is provided by nodes 406 a and 406 b . The differential input current source 102 a is connected to the gate of the NMOSFET 106 a and the differential input current source 102 b is connected to the gate of the NMOSFET 106 b . The threshold level of the differential comparator 500 is assumed to be zero. The differential comparator 500 depicted in FIG. 5 behaves in a similar manner to the differential comparator 100 depicted in FIG. 1 . When the differential-mode signal current I IN is greater than the threshold level, the node 406 a will “clip” to V DD and the node 406 b will “clip” to V SS . Alternatively, when the differential-mode signal current I IN is below the threshold level, the node 406 a will “clip” to V SS and the node 406 b will “clip” to V DD .

No current flows through the resistive elements 402 a and 402 b after the differential comparator 500 has fully latched. A voltage at the gate of the NMOSFET 106 a is therefore equal to a voltage at the gate of the PMOSFET 118 a . Likewise, a voltage at the gate of the NMOSFET 106 b is equal to a voltage at the gate of the PMOSFET 118 b . This allows the gates of the NMOSFETs 106 a and 106 b to serve as the output of the differential comparator 500 or the gates of the PMOSFETs 118 a and 118 b to serve as the output.

The resistive element 402 a and the resistive element 402 b are chosen to provide a desired drain current for the NMOSFET 106 a and the PMOSFET 118 a and a desired drain current for the NMOSFET 106 b and the PMOSFET 118 b , respectively. For example, drain currents of approximately 0.1 mA can be achieved by selecting the resistive element 402 a and the resistive element 402 b to each be approximately 2 kΩ, given a voltage drop of approximately 200 mV across each resistive element. By increasing the bias currents to 0.1 mA, the differential comparator 500 can maintain the same BER when the clock speed is increased by a factor of about ten. For example, the differential comparator 500 can operate at approximately 1 GHz and can achieve the same BER as the differential comparator 100 operating at only 100 MHz. Alternatively, the differential comparator 500 can be operated at the same clock speed as the differential comparator 100 while exhibiting an improved BER.

The resistive element 402 a and the resistive element 402 b depicted in FIG. 4 can each be implemented as a linear resistor. Alternatively, the resistive element 402 a and the resistive element 402 b can each be implemented, e.g., as a transmission gate. FIG. 6 illustrates the differential comparator 500 where the resistive element 402 a and the resistive element 402 b are each implemented as transmission gates.

The resistive element 402 a includes a PMOSFET 604 a and an NMOSFET 606 a . A gate of the PMOSFET 604 a is connected to the voltage supply V SS . A gate of the NMOSFET 606 a is connected to the voltage supply V DD . A source of the PMOSFET 604 a is connected to a drain of the NMOSFET 606 a at the node 406 a . A source of the NMOSFET 606 a is connected to a drain of the PMOSFET 604 a at the node 404 a . The resistive element 402 b includes a PMOSFET 604 b and an NMOSFET 606 b . A gate of the PMOSFET 604 b is connected to the voltage supply V SS . A gate of the NMOSFET 606 b is connected to the voltage supply V DD . A source of the PMOSFET 604 b is connected to a drain of the NMOSFET 606 b at the node 406 b . A source of the NMOSFET 606 b is connected to a drain of the PMOSFET 604 b at the node 404 b.

Essentially, the source and drain terminals of the PMOSFET 604 a and the NMOSFET 606 a are connected in parallel and the gate terminals are driven by opposite phase logic signals. With the gate of the NMOSFET 606 a connected to the supply voltage V DD , the transmission gate formed by the PMOSFET 604 a and the NMOSFET 606 a is in a conducting state. Specifically, the transmission gate of the PMOSFET 604 a and the NMOSFET 606 a are biased to be in a triode region of operation when a voltage of approximately 100 mV is applied across the node 404 a and the node 406 a . When the transmission gate formed by the PMOSFET 604 a and the NMOSFET 606 a is in the conducting state, the node 404 a and the node 406 a are connected together through the parallel combination of the on-resistances of the PMOSFET 604 a and the NMOSFET 606 a . In this way, the transmission gate formed by the PMOSFET 604 a and the NMOSFET 606 a provides a bidirectional resistive connection between the node 404 a and the node 406 a.

The transmission gate formed by the NMOSFET 606 b and the PMOSFET 604 b operates in a similar manner. That is, the transmission gate formed by the PMOSFET 604 b and the NMOSFET 606 b also provides a bidirectional resistive connection between the node 404 b and the node 406 b.

FIG. 7 illustrates a differential comparator 700 that is an alternative configuration of the differential comparator 500 . The arrangement of the differential comparator 700 differs from the arrangement of the differential comparator 500 in the connections of the differential input current sources 102 a and 102 b . Specifically, in FIG. 7 , the differential input current source 102 a is connected to the drain of the NMOSFET 106 a and the differential input current source 102 b is connected to the drain of the NMOSFET 106 b . Changing the connections of the differential input current sources 102 a and 102 b in this way “flips” an output of the differential comparator 700 for a given input. That is, when I IN is greater than the threshold level, the node 406 a will now “clip” to V SS while the node 406 b will “clip” to V DD . When I IN is below the threshold level, the node 406 a will now “clip” to V DD while the node 406 b will “clip” to V SS . The inverted operation of the differential comparator 700 provides another possible implementation of the present invention.

FIG. 8 illustrates the differential comparator 700 where the resistive element 402 a and the resistive element 402 b are each implemented as transmission gates. The differential comparator 700 depicted in FIG. 8 operates in a similar manner to the differential comparator 700 depicted in FIG. 7 .

Conclusion

It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined in the appended claims. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.