Voltage level translator circuit

The present invention relates generally to electronic circuits, and more particularly relates to voltage level translator circuits.

Certain portable applications, including wireless handsets, notebook computers and personal digital assistants (PDAs), often employ circuitry which runs on two or more different voltage levels. For example, circuitry utilized with such portable applications may be configured so that a portion of the circuitry, such as, for example, input/output (IO) buffers, runs at a higher voltage level (e.g., about 3.3 volts), while another portion of the circuitry, such as, for example, core logic, runs at a substantially lower voltage level (e.g., about 1.0 volt). This difference in voltage levels often necessitates the use of a voltage level translator circuit for interfacing between the multiple voltage levels.

Conventional voltage level translator circuits have generally been found to be unreliable and/or at least partially inoperable at certain process, voltage and/or temperature (PVT) conditions, and/or to consume substantial direct current (DC) power. In certain portable applications, it is not uncommon to employ an appreciable number (e.g., hundreds) of voltage level translator circuits, and therefore the overall DC power consumption attributable to these voltage level translator circuits can be excessive. Moreover, for portable applications, power is typically supplied by a battery having a limited operating life. Consequently, in order to extend the operating life of the battery, it would be advantageous to eliminate or substantially reduce the amount of DC power consumed by the voltage level translator circuit(s).

There exists a need, therefore, for an improved voltage level translator circuit for interfacing between multiple voltage levels that does not suffer from one or more of the problems exhibited by conventional voltage level translator circuits.

The present invention meets the above-noted need by providing techniques for interfacing between multiple voltage levels in a circuit, such as, for example, between an input signal, which is referenced to a lower core supply voltage of the circuit, and an output signal, which is referenced to a higher supply voltage of the circuit, without any significant DC power consumption.

In accordance with one aspect of the invention, a voltage level translator circuit for translating an input signal referenced to a first voltage level to an output signal referenced to a second voltage level includes an input stage for receiving the input signal. The input stage includes at least one transistor device having a first threshold voltage associated therewith. The voltage level translator circuit further includes a latch circuit operative to store a signal representative of a logical state of the input signal. The latch circuit includes at least one transistor device having a second threshold voltage associated therewith, the second threshold voltage being greater than the first threshold voltage. A voltage clamp is operatively connected between the input stage and the latch circuit, the voltage clamp being configured to limit a voltage across the input stage based, at least in part, on a control signal presented thereto. The voltage level translator circuit includes a reference generator circuit for generating the control signal, a steady state value of the control signal being substantially equal to the first voltage level. The reference generator circuit is configured to adjust a voltage level of the control signal in response to the input signal.

These and other features and advantages of the present invention will become apparent from the following detailed description of illustrative embodiments thereof, which is to be read in connection with the accompanying drawings.

FIG. 1 is a schematic diagram illustrating a conventional voltage level translator circuit.

FIG. 2 is a schematic diagram depicting a voltage level translator circuit in which the methodologies of the present invention may be implemented.

FIG. 3 is a schematic diagram illustrating a voltage reference circuit suitable for use in the voltage level translator circuit shown in FIG. 2 .

FIG. 4 is a schematic diagram illustrating an exemplary voltage level translator circuit, formed in accordance with one embodiment of the present invention.

FIG. 5 is a schematic diagram illustrating an exemplary voltage reference circuit which may be used in the voltage level translator circuit shown in FIG. 4 .

FIG. 6 is a graphical representation depicting exemplary voltage levels as a function of time for certain signals corresponding to the circuits shown in FIGS. 4 and 5 , in accordance with the present invention.

FIG. 7 is a schematic diagram illustrating an exemplary voltage level translator circuit, formed in accordance with a second embodiment of the present invention.

FIG. 8 is a schematic diagram illustrating an exemplary voltage level translator circuit, formed in accordance with a third embodiment of the present invention.

The present invention will be described herein in the context of illustrative voltage level translator circuits. It should be understood, however, that the present invention is not limited to these or any other particular circuit arrangements. Rather, the invention is more generally applicable to improved techniques for interfacing between multiple voltage levels in a circuit without consuming any significant DC power. Furthermore, although implementations of the present invention are described herein with specific reference to p-type metal-oxide-semiconductor (PMOS) and n-type metal-oxide-semiconductor (NMOS) transistor devices, as may be formed using a complementary metal-oxide-semiconductor (CMOS) fabrication process, it is to be appreciated that the invention is not limited to such transistor devices and/or such a fabrication process, and that other suitable devices, such as, for example, bipolar junction transistors (BJTs), etc., and/or fabrication processes (e.g., Bipolar, BiCMOS, etc.), may be similarly employed, as will be understood by those skilled in the art.

FIG. 1 depicts a standard voltage level translator circuit 100 that can be used to translate an input signal A, referenced to a lower core supply voltage VDDCORE, to an output signal Z which is referenced to a higher supply voltage VDDIO. In many applications, the lower core supply voltage VDDCORE is typically about 1.0 volt and the higher supply voltage VDDIO is typically about 3.3 volts. It is to be appreciated, however, that the present invention is not limited to these or to any other particular voltage levels for VDDCORE and VDDIO. Furthermore, the techniques of the present invention may be similarly employed to translate an input signal referenced to the higher supply voltage VDDIO to an output signal referenced to the lower core supply voltage VDDCORE, as will be understood by those skilled in the art.

Input signal AN is a logical inversion of input signal A, such that when signal A is a logic high level, signal AN is a logic low level, and vice versa. The voltage level translator circuit 100 is powered by the higher supply voltage VDDIO and receives, as its negative voltage supply, VSS. The term “negative voltage supply” as used herein is intended to refer to a value of the voltage supply relative to the higher supply voltage VDDIO, and does not necessarily refer to a voltage less than zero volts, although using a voltage less than zero volts is contemplated by the invention.

Traditional mixed signal integrated circuit processes typically offer “high voltage” and “low voltage” transistor devices. The high voltage devices generally have a nominal threshold voltage of about 0.75 volts and are intended to operate with the higher supply voltage VDDIO (e.g., about 3.3 volts). The low voltage devices have a nominal threshold voltage which is substantially lower than the high voltage devices, such as, for example, about 0.35 volts, and are intended to operate with the lower core supply voltage VDDCORE (e.g., about 1.0 volt). In the voltage level translator circuit 100 , all of the transistor devices, namely, M 3 P 1 , M 3 P 2 , M 3 P 3 , M 3 N 1 , M 3 N 2 and M 3 N 3 , are high voltage devices.

The voltage level translator circuit 100 comprises a pair of PMOS transistors M 3 P 1 and M 3 P 2 each having a source terminal (S) connected to the positive voltage supply VDDIO, and having a gate terminal (G) of one transistor connected to a drain terminal (D) of the other transistor in a cross-coupled arrangement. Specifically, the gate terminal of M 3 P 1 is connected to the drain terminal of M 3 P 2 at node i 1 , and the gate terminal of M 3 P 2 is connected to the drain terminal of M 3 P 1 at node i 2 . It is to be appreciated that, because the MOS device is symmetrical in nature, and thus bidirectional, the assignment of source and drain designations in the MOS device is essentially arbitrary. Therefore, the source and drain regions may be referred to generally as first and second source/drain regions, respectively, where “source/drain” in this context denotes a source region or a drain region.

The voltage level translator circuit 100 further comprises NMOS transistor M 3 N 1 having a source terminal connected to the negative voltage supply VSS, a drain terminal connected to node i 2 , and a gate terminal for receiving input signal AN. Voltage level translator circuit 100 also includes NMOS transistor M 3 N 2 having a source terminal connected to the negative voltage supply VSS, a drain terminal connected to node i 1 , and a gate terminal for receiving input signal A. An output stage comprising PMOS transistor M 3 P 3 and NMOS transistor M 3 N 3 connected togther as a standard inverter, is connected to node i 1 and generates an output signal Z of the voltage level translator circuit 100 .

Under most operating conditions of the voltage level translator circuit 100 , signal A being a logic high (“1”) turns on transistor M 3 N 2 , pulling node i 1 low. Signal AN being an inversion of signal A will thus be a logic low (“0”), thereby turning off transistor M 3 N 1 . Node i 1 being at a low voltage turns on transistor M 3 P 1 , pulling node i 2 high and turning off transistor M 3 P 2 . The output signal Z being an inversion of node i 1 will be a logic high when node i 1 is low. A primary disadvantage of the voltage level translator circuit 100 , however, is that, under certain PVT conditions, such as, for example, when the temperature is low (e.g., about zero degrees Celsius), the threshold voltage of transistors M 3 N 1 and M 3 N 2 may be high, such as about 0.8 volt or higher, which is only about one tenth of a volt or less below a minimum voltage limit of the lower core supply VDDCORE (e.g., about 0.9 volt) to which input signal A is referenced. Due to voltage drops internal to the circuit, the actual voltage seen by devices M 3 N 1 and M 3 N 2 could be even lower. With less than about one tenth of a volt of overdrive, transistors M 3 N 1 and M 3 N 2 will be unacceptably slow and may even fail to turn on entirely, thus rendering the voltage level translator circuit 100 unreliable and/or inoperable.

One way to overcome this problem is to employ low voltage NMOS transistors M 1 N 1 and M 1 N 2 in place of the high voltage transistors M 3 N 1 and M 3 N 2 , respectively, as shown in the voltage level translator circuit 200 of FIG. 2 . As previously stated, low voltage transistors typically have a nominal threshold voltage (e.g., about 0.35 volt) which is substantially lower than the nominal threshold voltage of high voltage transistors, thus providing more overdrive compared to the high voltage devices. The additional few tenths of a volt of overdrive obtained by using low voltage devices is generally sufficient to ensure that the voltage level translator circuit 200 remains operable over the desired range of PVT variations. However, the voltage appearing across any two terminals of transistors M 1 N 1 and M 1 N 2 should be limited, for example, to less than an upper limit of the lower core supply VDDCORE, typically about 1.26 volts, so as to avoid damaging the devices. Since nodes i 1 and i 2 can be pulled up to the higher supply voltage VDDIO, which is nominally about 3.3 volts, transistors M 3 N 1 and M 3 N 2 are preferably added between the low voltage transistors M 1 N 1 and M 1 N 2 and the cross-coupled high voltage PMOS transistors M 3 P 1 and M 3 P 2 , respectively. Specifically, source terminals of transistors M 1 N 1 and M 1 N 2 are connected to VSS, a drain terminal of M 1 N 1 is connected to a source terminal of transistor M 3 N 1 at node i 3 , and a drain terminal of M 1 N 2 is connected to a source terminal of transistor M 3 N 2 at node i 4 . A drain terminal of M 3 N 1 is connected to the drain terminal of transistor M 3 P 1 at node i 2 , and a drain terminal of M 3 N 2 is connected to the drain terminal of transistor M 3 P 2 at node i 1 . The cross-coupled devices M 3 P 1 and M 3 P 2 may be connected in a manner similar to that shown in FIG. 1 .

In order to bias transistors M 3 N 1 and M 3 N 2 to a desired operating point so as to limit the voltage across devices M 1 N 1 and M 1 N 2 , respectively, gate terminals of M 3 N 1 and M 3 N 2 may be connected to a common reference voltage VREF. For most PVT conditions and for low speed applications (e.g., less than about 200 megahertz (MHz)), it would be sufficient to use the lower core supply VDDCORE as the reference voltage VREF. Since these devices do not switch on and off, one tenth of a volt of overdrive is generally acceptable to keep the devices turned on. However, in order to satisfy all desired PVT conditions, devices M 3 N 1 and M 3 N 2 may be required to be sized significantly large, and the voltage level translator circuit 200 may fail to operate reliably even for a moderate speed of about 200 MHz.

A solution is to employ a bias generator circuit 300 , as illustrated in FIG. 3 , for generating the reference voltage VREF, which is preferably slightly higher than VDDCORE to provide ample overdrive for devices M 3 N 1 and M 3 N 2 . As apparent from the figure, bias generator circuit 300 may include a pair of high voltage diode-connected NMOS transistor devices M 3 N 4 and M 3 N 5 . The two devices M 3 N 4 and M 3 N 5 are connected in series between VSS and VDDIO via a resistor R 1 . Reference voltage VREF may be set to a desired level by selecting an appropriate resistance value for resistor R 1 and/or appropriately sizing devices M 3 N 4 and M 3 N 5 , as will be understood by those skilled in the art. The reference voltage VREF generated by the bias generator circuit 300 essentially clamps nodes i 3 and i 4 to a high threshold voltage (e.g., about 0.75 volt) above VSS. The resistance value of resistor R 1 is chosen to be high enough (e.g., greater than about 100 kilo (K) ohms) such that the current I BIAS consumed by the bias generator circuit 300 is relatively small (e.g., micoramperes).

A filter capacitor C 1 is connected between an output node n 1 of the bias generator circuit 300 and the negative voltage supply VSS to help filter out any high frequency components that may be present in the reference voltage VREF generated by the bias generator circuit 300 . Capacitor C 1 is preferably chosen to be about 0.5 picofarad (pF). The current I BIAS in the bias generator circuit 300 and the capacitor C 1 are scaled up with the required speed of operation of the voltage level translator circuit 200 in which the bias generator circuit 300 may be employed.

Unfortunately, in addition to consuming considerable area in the integrated circuit, due at least in part to the high-valued resistor R 1 and capacitor C 1 , the bias generator circuit 300 consumes DC current, which is undesirable, particularly for portable applications in which an extended battery life is important. In certain applications, one bias generator is required for each input/output buffer, and there may be hundreds of buffers employed in a given integrated circuit device. Therefore, the overall DC current consumption attributable to the bias generator circuits can become quite significant.

FIG. 4 is a schematic diagram illustrating an exemplary voltage level translator circuit 400 , formed in accordance with the present invention. The illustrative voltage level translator circuit 400 provides a simple and robust solution that is capable of elegantly interfacing between multiple voltage levels in a circuit without any significant DC power consumption. Voltage level translator circuit 400 preferably includes an input stage 408 configured for receiving at least one input signal (e.g., signal A) which is referenced to a lower core supply voltage VDDCORE (e.g., about 1.0 volt), and a latch circuit 402 for at least temporarily storing an output signal which is referenced to a higher supply voltage VDDIO (e.g., about 3.3 volts) and is representative of a logical state of the input signal. A voltage clamp 406 is operatively coupled between the input stage 408 and the latch circuit 402 . The voltage level translator circuit 400 may also include an output stage 404 coupled to the latch circuit 402 for buffering the output signal stored in the latch circuit and for generating a buffered output signal (e.g., signal Z) of the voltage level translator circuit having substantially rail-to-rail (e.g., VSS to VDDIO) logic levels.

Like the voltage level translator circuit 200 depicted in FIG. 2 , the input stage 408 of voltage level translator circuit 300 preferably comprises a pair of low voltage NMOS devices M 1 N 1 and M 1 N 2 . The low voltage devices, having a lower threshold voltage (e.g., about 0.35 volt) compared to a high voltage device, provide the input stage with additional overdrive so as to ensure proper operation of the voltage level translator circuit 400 over a desired range of PVT variations. Each of devices M 1 N 1 and M 1 N 2 includes a source, a drain and a gate terminal. The source terminals of M 1 N 1 and M 1 N 2 are connected to a negative voltage supply, which may be VSS. The gate terminal of M 1 N 1 preferably receives input signal AN, which, as stated above, is referenced to the lower core supply voltage VDDCORE. The gate terminal of M 1 N 2 receives input signal A, which is a logical inversion of input signal AN and is similarly referenced to the lower core supply voltage VDDCORE. The drain terminals of M 1 N 1 and M 1 N 2 are connected to the voltage clamp 406 at nodes i 3 and i 4 , respectively.

Since devices M 1 N 1 and M 1 N 2 are low voltage devices, voltage clamp 406 preferably serves as a primary means for preventing the voltage at nodes i 3 and i 4 from exceeding a voltage substantially equal to a maximum upper limit of the lower core supply voltage VDDCORE, which is about 1.26 volts for this exemplary embodiment. In this manner, device M 1 N 1 and M 1 N 2 are protected from voltage overstress, which can lead to device failure. Voltage clamp 406 preferably comprises a pair of high voltage NMOS transistor devices M 3 N 1 and M 3 N 2 . As previously explained, high voltage devices typically have threshold voltages that are substantially higher compared to low voltage devices (e.g., about 0.75 volt) for a given integrated circuit process.

The voltage clamp 406 is preferably configured such that a source terminal of M 3 N 1 is connected to the drain terminal of M 1 N 1 at node i 3 , a source terminal of M 3 N 2 is connected to the drain terminal of M 1 N 2 at node i 4 , and drain terminals of M 3 N 1 and M 3 N 2 are connected to the latch circuit 402 at nodes i 2 and i 1 , respectively. Gate terminals of M 3 N 1 and M 3 N 2 are connected to bias voltage signals VBN and VB, respectively, for selectively controlling the voltages at corresponding nodes i 3 and i 4 . Unlike devices M 3 N 1 and M 3 N 2 in the voltage level translator circuit 200 depicted in FIG. 2 , the devices M 3 N 1 and M 3 N 2 in voltage clamp 406 are not biased by a common reference voltage (VREF) that is generated by a bias generator circuit which consumes DC power. Instead, each of devices M 3 N 1 and M 3 N 2 in voltage clamp 406 receives a separate bias voltage VBN and VB, respectively. Furthermore, signals VBN and VB are generated without consuming any significant DC power, as will be described in further detail below.

FIG. 5 illustrates an exemplary bias generator circuit 500 for generating the bias signals VB and VBN used by the voltage level translator circuit 400 of FIG. 4 , in accordance with one aspect of the invention. It is to be appreciated that bias generator circuit 500 may be included in the voltage clamp 406 , or, alternatively, the bias generator circuit 500 may be external to the voltage clamp 406 . Moreover, it is contemplated that two or more voltage level translator circuits 400 may share a given bias generator circuit 500 . The bias generator circuit 500 is configured to provide sufficient overdrive for devices M 3 N 1 and M 3 N 2 over a desired range of PVT variations, without consuming any significant DC power.

Bias generator circuit 500 is operative to dynamically control the respective overdrives to devices M 3 N 1 and M 3 N 2 as signals A and AN switch from one logic level to another. To accomplish this, signal VB, which is generated at node n 3 , is connected to the lower core supply voltage VDDCORE by one of a pair of low voltage PMOS transistor devices, M 1 P 1 and M 1 P 2 . For instance, when input signal A is a logic low, signal VB will be connected to VDDCORE via device M 1 P 1 , and when signal A is a logic high, signal VB will be connected to VDDCORE via device M 1 P 2 . Source terminals of devices M 1 P 1 and M 1 P 2 are connected to VDDCORE and drain terminals of M 1 P 1 and M 1 P 2 are connected to node n 3 . A gate terminal of M 1 P 1 preferably receives signal A and a gate terminal of M 1 P 2 receives signal AN. A capacitor C 1 P 1 is preferably connected to node n 3 at a first end (e.g., terminal 1 ), and receives signal A at a second end (e.g., terminal 2 ). The value of capacitor C 1 P 1 is preferably chosen to be about 0.35 pF for high-frequency operation (e.g., above about 500 MHz), or about 0.1 pF for low-frequency operation (e.g., less than about 500 MHz), although the invention is not limited to a particular value for capacitor C 1 P 1 .

Likewise, signal VBN, which is generated at node n 4 , is connected to the lower core supply voltage VDDCORE by one of a pair of low voltage PMOS transistor devices, M 1 P 3 and M 1 P 4 . Source terminals of devices M 1 P 3 and M 1 P 4 are connected to the lower core supply voltage VDDCORE and drain terminals of M 1 P 3 and M 1 P 4 are connected to node n 4 . A gate terminal of M 1 P 3 preferably receives signal AN and a gate terminal of M 1 P 4 receives signal A. A capacitor C 1 P 2 is preferably connected to node n 4 at a first end (e.g., terminal 1 ), and receives signal AN at a second end (e.g., terminal 2 ). The value of capacitor C 1 P 2 , like capacitor C 1 P 1 , is preferably chosen to be about 0.35 pF for high-frequency operation, or about 0.1 pF for low-frequency operation, although the invention is not limited to a particular value for capacitor C 1 P 2 .

It is to be appreictaed that while the bias generator circuit 500 is shown as comprising low voltage PMOS devices, namely, M 1 P 1 , M 1 P 2 , M 1 P 3 and M 1 P 4 , gated by input signals A or AN, the bias generator circuit is not limited to PMOS devices, but may alternatively include other resistive load devices, such as, for example, an NMOS device, resistor, etc. For example, PMOS devices M 1 P 1 and M 1 P 2 may be replaced by a resistor (not shown) connected between node n 3 and VDDCORE. Moreover, the PMOS devices need not be gated by input signals A and AN, but may instead be grounded gate PMOS devices, as will be understood by those skilled in the art.

A steady state value of signals VB and VBN will be substantially equal to the lower core supply voltage VDDCORE, since nodes n 3 and n 4 will be pulled up to VDDCORE either via devices M 1 P 1 or M 1 P 2 , for signal VB, or via devices M 1 P 3 and M 1 P 4 , for signal VBN. Capacitors C 1 P 1 and C 1 P 2 provide signals VB and VBN, respectively, with a momentary boost in either the positive or negative direction relative to the steady state value of VB and VBN. The voltage across capacitors C 1 P 1 and C 1 P 2 are either added to or subtracted from the steady state value of signals VB and VBN as a function of the direction of transition of signals A and AN, respectively. For example, assuming node n 3 is at the lower core supply voltage VDDCORE during steady state, when signal A transitions from a logic low to a logic high, signal VB will receive a boost in the positive direction. When signal A transitions from a logic high to a logic low, signal VB will receive a boost in the negative direction (e.g., less than the steady state value). A negative boost in signal VB is advantageous for more quickly turning off device M 3 N 2 , thereby speeding the switching of latch circuit 402 in the voltage level translator circuit 400 . Signal VBN is generated in a similar manner.

Since signal A is referenced with respect to VDDCORE, the voltage across capacitor C 1 P 1 will also be substantially equal to VDDCORE when signal A is low. When signal A transitions from a logic low to a logic high, signal VB will momentarily rise to about a few hundred millivolts higher than transporting data through VDDCORE, before returning to its steady state value of VDDCORE. When signal A is a logic high, the voltage across capacitor C 1 P 1 will be about zero, since node n 3 will also be at about VDDCORE. Therefore, when signal A transitions from a logic high to a logic low, signal VB will be pulled low momentarily as capacitor C 1 P 1 charges to VDDCORE. Signal VBN is generated in a similar manner, only based on signal AN rather than signal A. The duration of the dynamic boost in signals VB and VBN will be primarily a function of the charging and discharging times of the capacitors C 1 P 1 and C 1 P 2 , respectively. These charging and discharging times may be selectively controlled, at least in part, by sizing the devices, namely, devices M 1 P 1 , M 1 P 2 , M 1 P 3 and M 1 P 4 , as desired.

Each of capacitors C 1 P 1 and C 1 P 2 may be implemented as a PMOS device, either in an accumulation mode or in a depletion mode, although alternative structures for implementing capacitors C 1 P 1 and C 1 P 2 are similarly contemplated. A bulk terminal of each capacitor can be connected to VDDCORE to reduce integrated circuit layout area. At least a portion of the structures for generating signal VB (e.g., M 1 P 1 , M 1 P 2 and C 1 P 1 ) or signal VBN (e.g., M 1 P 3 , M 1 P 4 and C 1 P 2 ) may be shielded by metal, with the metal shield connected to VDDCORE.

With continued reference to FIG. 4 , latch circuit 402 preferably comprises a pair of high voltage PMOS transistor devices M 3 P 1 and M 3 P 2 connected in a cross-coupled arrangement, in a manner similar to the voltage level translator circuit 200 of FIG. 2 . Specifically, a gate terminal of M 3 P 1 is connected to a drain terminal of M 3 P 2 at node i 1 , and a gate terminal of M 3 P 2 is connected to a drain terminal of M 3 P 1 at node i 2 . Source terminals of M 3 P 1 and M 3 P 2 are connected to a positive voltage supply, which may be VDDIO. Latch circuit 402 is configured to at least temporarily store the voltages at nodes i 1 and i 2 , which are representative of the input signals A and AN, without consuming any significant DC power. The present invention contemplates that alternative circuitry may be employed for storing the output of the voltage translator circuit 400 , as will be understood by those skilled in the art.

The output stage 404 , which may be used to generate output signal Z, preferably comprises a high voltage NMOS transistor device M 3 N 3 and a high voltage PMOS transistor device M 3 P 3 connected in an inverter arrangement. Specifically, gate terminals of M 3 N 3 and M 3 P 3 are connected together to form an input of output stage 404 , drain terminals of M 3 N 3 and M 3 P 3 are connected together to form an output of the output stage at node n 1 , and source terminals of M 3 N 3 and M 3 P 3 are connected to voltage supplies VSS and VDDIO, respectively. The input of output stage 404 is connected to an output of the latch circuit 402 at node i 1 . The output stage 404 serves primarily to generate the output signal Z having substantially rail-to-rail logic levels and to buffer the output of the latch circuit 402 at node i 1 . Since the output stage 404 in the exemplary voltage level translator circuit 400 is inverting, its input is taken from node i 1 in order to keep the logic level of the output signal Z the same as the input signal A. It is to be appreciated that, in an alternative embodiment of the invention, when a noninverting output stage is employed, the input of the output stage would preferably be connected to node i 2 so that the logic level of output signal Z is the same as the logic level of the input signal A.

By way of example only, the operation of voltage level translator circuit 400 will be described. When signal A is a logic low and signal AN, being a logical inversion of signal A, is a logic high, device M 1 N 1 is turned on and device M 1 N 2 is turned off. Device M 1 N 1 being turned on pulls node i 2 low, provided device M 3 N 1 is fully turned on, which turns on device M 3 P 2 . With M 3 P 2 turned on, node i 1 will be pulled high, thereby turning off device M 3 P 1 . With M 1 N 2 turned off, node i 4 will also be high, provided device M 3 N 2 is fully turned on. When signal A switches from logic low to logic high, M 1 N 2 turns on and pulls node i 4 low. At the same time, signal A switching from logic low to logic high dynamically causes signal VB to rise above its steady state voltage VDDCORE, thereby providing a substantial overdrive for device M 3 N 2 and causing node i 1 to be pulled low more quickly compared to if signal VB were not boosted higher than VDDCORE. Concurrently, signal AN switching from logic high to logic low dynamically causes signal VBN to drop below its steady states voltage VDDCORE, thereby turning off device M 3 N 1 more quickly and providing more effective isolation between nodes i 2 and i 3 . This allows node i 2 to transition to a logic high faster, turning off M 3 P 2 .

Similarly, when signal A switches from logic high to logic low and signal AN switches from logic low to logic high, M 1 N 1 turns on and pulls node i 3 low, and M 1 N 2 turns off. At the same time, signal AN switching from logic low to logic high dynamically causes signal VBN to rise above its steady state voltage VDDCORE, thereby providing a substantial overdrive for device M 3 N 1 and causing node i 2 to be pulled low more quickly compared to if signal VBN were not boosted higher than VDDCORE. Concurrently, signal A switching from logic high to logic low dynamically causes signal VB to drop below its steady states voltage VDDCORE, thereby turning off device M 3 N 2 more quickly and providing more effective isolation between nodes i 1 and i 4 . This allows node i 1 to transition to a logic high faster, turning off M 3 P 1 .

FIG. 6 is a graphical representation of exemplary simulation results of certain signals in the illustrative voltage level translator circuit 400 of FIG. 4 reference circuit 500 of FIG. 5 , as a function of time (in seconds). The simulation results are provided for the illustrative voltage level translator circuit operating at a speed of about 500 MHz and under worst case PVT conditions (e.g., slow integrated circuit process, low temperature). Input signal A is represented by graph 602 , bias signal VB is represented by graph 604 , input signal AN is represented by graph 606 , bias signal VBN is represented by graph 608 , and output signal Z is represented by graph 610 . As apparent from the figure, bias signals VB and VBN change dynamically either up or down from their steady state voltages of about 0.9 volt at each edge transition (e.g., t 1 , t 2 , t 3 ) of input signals A and AN.

FIG. 7 is a schematic diagram illustrating an exemplary voltage level translator circuit 700 , formed in accordance with another embodiment of the invention. The voltage level translator circuit 700 may be essentially identical to the voltage level translator circuit 400 shown in FIG. 4 , except for the addition of two low voltage PMOS transistor devices M 1 P 5 and M 1 P 6 . Source terminals of M 1 P 5 and M 1 P 6 are preferably connected to the low core supply voltage VDDCORE, a drain terminal of M 1 P 5 is connected to node i 3 , and a drain terminal of M 1 P 6 is connected to node i 4 . A gate terminal of M 1 P 5 preferably receives input signal AN and a gate terminal of M 1 P 6 receives input signal A.

Devices M 1 P 5 and M 1 P 6 function primarily to speed up the operation of voltage level translator circuit 700 . For example, when signal A switches from a logic low to a logic high, and thus signal AN switches from a logic high to a logic low, device M 1 P 5 advantageously pulls node i 3 high, such as to about VDDCORE. Similarly, when signal A switches from a logic high to a logic low and signal AN switches from a logic low to a logic high, device M 1 P 6 pulls node i 4 high. In addition, since M 1 P 5 and M 1 P 6 are connected to VDDCORE, these devices, when active, effectively clamp the voltage at nodes i 3 and i 4 to a maximum of about VDDCORE, thereby further ensuring that devices M 1 N 1 and M 1 N 2 are never stressed beyond an acceptable limit.

It is to be appreciated that the voltage level translation techniques of the present invention described herein may be used with alternative circuit configurations for translating among other voltage levels without consuming any significant DC power, as will be understood by those skilled in the art. For example, FIG. 8 illustrates an exemplary voltage level translator circuit 800 , formed in accordance with an alternative embodiment of the invention. Voltage level translator circuit 800 is similar to the voltage level translator circuit 400 shown in FIG. 4 , except that the circuit configurations are essentially flipped upside down from one another, and voltage level translator circuit 800 employs transistor devices having polarities opposite to the polarities of the transistor devices in voltage level translator circuit 400 , as will be understood by those skilled in the art. Additionally, VDDIO and VDDCORE are preferably zero volts, VSS is preferably about −3.3 volts, and a negative lower core supply voltage VSSCORE is preferably about −1.0 volt, where VDDIO and VDDCORE are electrically isolated from one another, and VSS and VS SCORE are electrically isolated from one another.

At least a portion of the voltage level translator circuit of the present invention may be implemented in an integrated circuit. A plurality of identical die are typically formed in a repeated pattern on a surface of a semiconductor wafer. Each die includes a device described herein, and may include other structures or circuits. The individual die are cut or diced from the wafer, then packaged as an integrated circuit. One skilled in the art would know how to dice wafers and package die to produce integrated circuits. Integrated circuits so manufactured are considered part of this invention.

Although illustrative embodiments of the present invention have been described herein with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments, and that various other changes and modifications may be made therein by one skilled in the art without departing from the scope of the appended claims.