Controlled impedance amplifiers

The present invention is generally related to mass storage devices including hard disk drive (HDD) systems, and more particularly to impedance mismatches between sensing heads on a flexible printed circuit (FPC) and a differential amplifier on a separate circuit board.

In a HDD system, sensing heads adapted to read data from a magnetic storage media are typically mounted on a flexible printed circuited (FPC), and are usually electrically connected to a differential amplifier on a fixed printed circuit board as shown in FIG. 1 . The sensor preamp connection is the connection between the MR sensor and the preamp, which has an input impedance of Rin. The FPC usually has a characteristic impedance of 50-150 ohms. Due to the impedance mismatch of the FPC impedance and the input impedance of the differential amplifier, the frequency response is not very flat, and in many cases peaking is present at a high frequency. The peaking in gain response is not favorable since it degrades group delay flatness, and generates high frequency noise. The peaking may also trigger an oscillation and makes the signal read back process unstable.

The present invention achieves technical advantages by adding feedback to the input of the differential amplifier to set the amplifier input impedance close to the FPC characteristic impedance. A matched source and input impedance produces a generally flat gain frequency response. As the result of flat response and wider gain bandwidth, white noise without high frequency peaking is obtained.

Conventionally, the input impedance may be set to a desired value by adding a termination resistor (R T ) in parallel with the input as shown in FIG. 2 . However, the termination resistor attenuates the input signal but not the differential amplifier's noise. Thus, input noise referred to the sensing device increases. Providing feedback according to the present invention is superior in terms of the input referred noise since the feedback attenuates both the input signal from the sensor and the noise from the amplifier. The input referred noise to the sensor does not degrade. In general, the sensing devices are set close to ground potential, while the differential amplifier's outputs are usually close to positive supply voltage due to a nature of a differential amplifier. The feedback resistors may be connected through AC coupling capacitors to compensate the voltage differences between input and output. However, the AC coupling capacitors causes settling issues and increases physical die size. The circuit of the present invention provides a method to connect the feedback resistors directly from outputs to inputs which have different potential.

The advantages of the present invention include matched impedance results in better system level frequency response, the direct feedback connection eliminates costly on-chip capacitors, and the input impedance can be adjusted by the amount of feedback.

FIG. 1 shows a typical HDD read back system;

FIG. 2 is an example of a conventional matched impedance system with a termination resistor;

FIG. 3 shows a schematic diagram of the present invention including feedback to the input of the differential amplifier; and

FIG. 4 is a schematic drawing of the present invention shown in FIG. 3 .

Referring now to FIG. 4 in view of FIG. 3, the present invention achieves technical advantages by providing feedback from the output to the input through resistors R 5 via a direct connection. The feedback resistors R 5 have two functions, providing both AC signal feedback and preventing DC level shifting. The amount of feedback is set by the combination of the first stage gain and the resistor value of each R 5 . The input voltage, Vin, is set by a bias control circuits VB+ and VB− (comprising Q 13 and R 7 , and Q 14 and R 8 respectively), and the feedback connection of R 5 does not disturb the input voltage. In general, Vin is set to ground level, thus, the voltage drop across each R 5 is set to the lower end of R 5 to be at ground level. This can be embodied by the following method.

The output voltage of the amplifier V OUT :

V OUT =V CC −I 1 *R 1 −2 V BE

By making a replica of the I 1 *R 1 voltage drop into I 3 *R 3 (i.e. I 1 *R 1 =I 3 *R 3 ), obtain a current that conducts through resistor R 4 (I 4 ): I4 = ( V CC - 2 * V BE - I 3 * R 3 ) / R 4 = ( V CC - 2 * V BE - I 1 * R 1 ) / R 4

The voltage at the lower end of the feedback resistor R 5 , V IN , is given by:

V

IN

=V

−I

5

*R

5

The voltage drop I 5 *R 5 can be set to equal to the I 4 *R 4 voltage drop with a simple current mirror, seen to include transistors Q 7 , Q 8 , Q 2 , Q 9 , Q 10 , Q 11 , and Q 12 to establish current I 4 , and resistor ratio of R 4 and R 5 , thus:

V

IN

=V

−I

4

*R

4

Substituting equation (2) into equation (3):

V IN =V CC −I 1 *R 1 −2 *V BE −( V CC −2 *V BE −I 1 *R 1 )=0

In conclusion, the lower end voltage of the feedback resistor R 5 can be set at a ground potential with a simple resistor ratio and the current mirror circuit. As shown in the equation, the feedback circuit has no voltage supply dependency nor temperature dependency in terms of DC voltage.

Though the invention has been described with respect to a specific preferred embodiment, many variations and modifications will become apparent to those:skilled in the art upon reading the present application. It is therefore the intention that the appended claims be interpreted as broadly as possible in view of the prior art to include all such variations and modifications.