Reference voltage generating circuit and low power consumption sensor

This application claims the benefit of Taiwan Patent Application No. 109142145, filed on Dec. 1, 2020, in the Taiwan Intellectual Property Office, the disclosure of which is incorporated herein in its entirety by reference.

1. Technical Field

The present disclosure relates to a reference voltage generating circuit, in particular to a self-biased and capacitive-coupled reference voltage generating circuit that can be started-up quickly and has a high power supply rejection ratio (PSRR), and a sensor using the reference voltage generating circuit.

2. Description of the Related Art

Reference voltage generating circuits are widely used in various electronic systems to generate voltages which are independent to the process, power supply, and temperature changes, so as to meet applications of more advanced battery-less Internet of Things (IoT) devices, such as patch-type sensing systems and biomedical implants. Therefore, designing a low-power consumption, small area and calibration-free reference voltage generating circuit has become a new requirement and challenge. In addition to the requirements of temperature, process, and voltage stability, low-power circuit design often results in long start-up time and reduced power supply rejection ratio. These two design indicators (i.e. start-up time and PSRR) are also stringent tests for circuit design.

With the increasing demand for self-powered sensing chips, many papers have proposed various temperature-stable voltage reference generating circuits with power consumption in the micro-Watt (μW) level, which mainly include sub-bandgap reference voltage generating circuit architecture and complementary metal-oxide-semiconductor field effect transistor (CMOS) architecture.

Compared with the CMOS architecture, the sub-bandgap reference voltage generating circuit generally has larger power consumption and chip area, but has a better temperature coefficient (TC). The sub-bandgap reference voltage generating circuit which uses the switch to switch the bipolar junction transistor (BJT) can reduce the static power consumption to tens of nano-Watt (nW), but the complex controlling and clock circuit and the capacitors which are used for suppressing noise will significantly increase the chip area. In addition, the hybrid reference voltage generating circuit which uses a leakage current has characteristics of CMOS and BJT, while achieving the purpose of low-power consumption and low temperature coefficient, but the leakage current of the parasitic diode narrows the temperature range, thus it cannot operate at high temperatures and low temperature environment. Furthermore, the hybrid reference voltage generating circuit uses a zero threshold voltage transistor, which increases the cost of manufacturing chips and makes it more difficult to control process variation.

Although the above designs can achieve low power consumption reference voltage generating circuits, the key problem of the actual design is that low power consumption designs will also lead to a decrease in bandwidth, causing serious problems in suppressing the interference coupling between 50-60 Hz frequency bandwidth. Regarding the current design of the power supply rejection ratio (PSRR) of the reference voltage generating circuit which power consumption is less than 100 nano-Watt, the bandwidth of the current design (i.e. the bandwidth which PSRR is less than −50 db) is limited.

Another problem is that low power consumption operation (less than 100 nW operation) causes a large impedance value between the power supply and the output of the reference voltage generating circuit, which makes the required time up to tens of milliseconds or more for voltage switching. It induces problems in circuits that require fast switching or response, such as vibration energy harvesting circuits and duty-cycling communication circuits.

It can be understood from the above description that the technical problem to be solved is to provide a reference voltage generating circuit with characteristics such as low temperature variation, low process variation, voltage stability, low power consumption, small area and calibration-free, and the reference voltage generating circuit needs to be able to improve the problems of long start-up time and reduced power supply rejection ratio caused by circuit design with low power consumption.

In order to solve the aforementioned conventional problems, an embodiment of the present disclosure provides a self-biased and capacitive-coupled reference voltage generating circuit, which comprises a current source circuit and a core circuit. An input terminal of the current source circuit is connected to a first feedback node, the voltage of the first feedback node is a reference voltage, and a plurality of output terminals of the current source circuit output current sources.

The core circuit comprises a first stacked diode-connected circuit and a second stacked diode-connected circuit. The first stacked diode-connected circuit has a first transistor with a diode-connected configuration and a second transistor having a conductive type same as that of the first transistor, a connection node of the first transistor and the second transistor outputs a threshold voltage difference value, an input terminal of the first stacked diode-connected circuit, the gates of the first transistor and the second transistor are connected to one of the output terminals of the current source circuit, and an output terminal of the second transistor is connected to a ground terminal.

The second stacked diode-connected circuit has a third transistor with a diode-connected configuration and a fourth transistor having a conductive type same as that of the third transistor, a connection node of the third transistor and the fourth transistor outputs a reference voltage, an input terminal of the second stacked diode-connected circuit, the gates of the third and fourth transistors are connected to another one of the output terminals of the current source circuit, there is a second feedback node formed between an output terminal of the fourth transistor and the input terminal of the current source circuit.

According to an embodiment of the present disclosure, the current source circuit comprises a cascode current mirror circuit.

According to an embodiment of the present disclosure, the cascode current mirror circuit comprises a first output circuit, a second output circuit, and an input circuit. The first output circuit comprises a first P-type transistor and a second P-type transistor, and a source of the first P-type transistor is connected to an operating voltage. The second P-type transistor is serially connected to the first P-type transistor, and the second P-type transistor outputs a first current source of the current sources to the first stacked diode-connected circuit.

The second output circuit comprises a third P-type transistor and a fourth P-type transistor, and a source of the third P-type transistor is connected to the operating voltage. The fourth P-type transistor is serially connected to the third P-type transistor, and the fourth P-type transistor outputs a second current source of the current sources to the second stacked diode-connected circuit, and a gate of the fourth P-type transistor is connected to a gate of the second P-type transistor.

The input circuit comprises a fifth P-type transistor, a sixth P-type transistor, a fifth N-type transistor and a sixth N-type transistor. a drain of the fifth P-type transistor is connected to the operating voltage, and a gate of the fifth P-type transistor is connected to a gate of the third P-type transistor and connected to a gate of the first P-type transistor.

A gate of the sixth P-type transistor is connected to a second feedback node, and a serial connection node of the sixth P-type transistor and the fifth P-type transistor is connected to the gate of the fifth P-type transistor.

A gate of the fifth N-type transistor is connected to the gate of the fourth transistor.

A gate of the sixth N-type transistor is connected to the first feedback node, and a serial connection node of the sixth N-type transistor and the fifth N-type transistor is connected to a third feedback node, and the third feedback node is connected to the gates of the fourth P-type transistor and the second P-type transistor, and a source of the sixth N-type transistor is connected to the ground terminal.

According to an embodiment of the present disclosure, the reference voltage generating circuit further comprises a first coupling capacitor and a second coupling capacitor. Two terminals of the first coupling capacitor are coupled between the gate of the fifth N-type transistor and the serial connection node of the fifth N-type transistor and the sixth P-type transistor.

A terminal of the second coupling capacitor is coupled to the serial connection node of the fifth N-type transistor and the sixth P-type transistor, and another terminal of the second coupling capacitor is coupled between the first output circuit and the first stacked diode-connected circuit.

According to an embodiment of the present disclosure, the reference voltage generating circuit further comprises a third coupling capacitor, a terminal of the third coupling capacitor is coupled to the gates of the fourth P-type transistor and the second P-type transistor, and another terminal of the third coupling capacitor is coupled to the ground terminal, and the high voltage terminal of the third coupling capacitor is coupled to the third feedback node, and a high voltage terminal of the two terminals of the third coupling capacitor is connected to a serial connection node of the sixth N-type transistor and the fifth N-type transistor through the third feedback node.

According to an embodiment of the present disclosure, a capacitance value of the second coupling capacitor is greater than a capacitance value of the first coupling capacitor.

According to an embodiment of the present disclosure, a size and a threshold voltage of the third transistor are respectively different from those of the fourth transistor.

According to an embodiment of the present disclosure, the reference voltage generating circuit lacks of a bipolar junction transistor.

Based on the above purposes, an embodiment of the present disclosure also provide a reference voltage generating circuit with a stacked diode-connected architecture, which comprises a current source circuit, a first transistor and a second transistor. An input terminal of the current source circuit is connected to an operating voltage, and the current source circuit outputs a current source.

A drain of the first transistor is connected to an output terminal of the current source circuit. The second transistor is serially connected to the first transistor, and a source of the second transistor is connected to a ground terminal, the serial connection node of the second transistor and the first transistor outputs a reference voltage, and the gates of the second transistor and the first transistor are connected to the output terminal of the current source circuit, a thickness of a gate oxide layer of the first transistor is smaller than a thickness of a gate oxide layer of the second transistor.

According to an embodiment of the present disclosure, the first transistor and the second transistor are a same conductive type.

Based on the above purposes, an embodiment of the present disclosure also provides a low-power consumption sensor, which is suitable for battery-less Internet of Things (IoT) devices, and the low-power consumption sensor comprises one of the above-mentioned reference voltage generating circuits.

According to an embodiment of the present disclosure, the low-power consumption sensor is a patch-type sensor or a biomedical implanter.

As mentioned above, a self-biased and capacitive-coupled reference voltage generating circuit and a reference voltage generating circuit with stacked diode-connected architecture according to an embodiments of the present disclosure have the following advantages:

1. By using a reference voltage generating circuit with a stacked diode-connected architecture, the threshold voltage difference of two transistors can be used to improve the stability of temperature coefficient, and to reduce the influence of process variation and the sensitivity of power supply.

2. By using a reference voltage generating circuit with a stacked diode-connected architecture, it can increase the resistance of the current source and suppress the power interference between the operating voltage and the output reference voltage.

3. Since bipolar junction transistors are not used in the circuit architecture, power consumption and chip area can be effectively reduced.

4. By using a self-biased feedback loop, it can eliminate the additional power consumption and chip area of the conventional starting circuit, and also prevent the leakage current of the startup circuit from affecting the temperature performance of the circuit.

5. With the coupled capacitor, the start-up time can be shortened to 0.2 milliseconds, and the bandwidth can be extended to 100 Hz to suppress interference in the 50-60 Hz frequency band from the commercial power supply.

In order to make the above and other purposes, features, advantages and embodiments of the present disclosure more obvious and understandable, the description of the accompanying drawings is as follows:

FIG. 1 is a schematic diagram of a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure.

FIG. 2 is a simplified schematic diagram of a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure.

FIG. 3 is a schematic diagram of a reference voltage generating circuit with a stacked diode-connected architecture according to an embodiment of the present disclosure.

FIG. 4 is a schematic diagram of a low power consumption sensor comprising a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure.

Hereinafter, at least one embodiment of the present disclosure will be described according to FIG. 1 to FIG. 4 . The descriptions are not intended to limit the implementation of the present disclosure, but only examples of the present disclosure.

Refer to FIG. 1 , which is a schematic diagram of a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure. As shown in the drawings, the self-biased and capacitive-coupled reference voltage generating circuit 10 comprises a current source circuit 100 and a core circuit 200 . An input terminal of the current source circuit 100 is connected to the first feedback node F 1 , the voltage of the first feedback node F 1 is the reference voltage VREF, and output terminals of the current source circuit 100 outputs current sources (IP 1 , IP 3 , and IP 5 ).

The core circuit 200 comprises a first stacked diode-connected circuit and a second stacked diode-connected circuit. The first stacked diode-connected circuit has a first transistor MN 1 with a diode-connected configuration (representing a connection between the gate and the drain) and a second transistor MN 2 having a conductive type same as that of the first transistor MN 1 (for example, the conductive type of the two transistors MN 1 and MN 2 are N-type). The difference value of the threshold voltages is outputted by the connection node of the first transistor MN 1 and the second transistor MN 2 . An input terminal of the first stacked diode-connected circuit (representing the drain of the first transistor MN 1 ) and the gates of the first transistor MN 1 and the second transistor MN 2 are connected to the output terminal (representing the current source IP 1 ) of the current source circuit 100 . An output terminal of the second transistor MN 2 (representing the source of the second transistor MN 2 ) is connected to the ground terminal GND.

The second stacked diode-connected circuit has a third transistor MN 3 with a diode-connected configuration (representing a connection between the gate and the drain) and a fourth transistor MN 4 having a conductive type same as that of the third transistor MN 3 (for example, the conductive type of the two transistors MN 3 and MN 4 are N-type). The reference voltage VREF is outputted by the connection node of the third transistors MN 3 and the fourth transistor MN 4 . An input terminal of the second stacked diode-connected circuit (representing the drain of the third transistor MN 3 ), the gates of the third transistor MN 3 and the fourth transistor MN 4 are connected to another output terminal of the current source circuit 100 (representing the current source IP 3 ). There is a second feedback node F 2 between an output terminal of the fourth transistor MN 4 (representing the source of the fourth transistor MN 4 ) and the input terminal of the current source circuit 100 .

First, the first stacked diode-connected circuit disclosed above will be described in detail. The transistor current (I D ) equation operated in the subthreshold region is shown as equation (1):

Where μ n represents the carrier mobility of the N-type transistor, C ox is the gate oxide capacitance of the transistor, W and L are the channel width and the channel length of the transistor respectively, m is the slope parameter, V T is the thermal voltage, V TH is the threshold voltage of the transistor, V GS is the relative voltage difference between the gate and source of the transistor, and V DS is the relative voltage difference between the drain and source of the transistor.

Therefore, if the voltage outputted by the connection node of the first transistor MN 1 and the second transistor MN 2 is used as the reference voltage VREF′, it can be expressed by the following equation (2):

V REF ′ = ( m t m b ⁢ V T ⁢ H ⁢ b - V T ⁢ H ⁢ t ) + m t ⁢ V T ⁢ ln ⁡ ( μ t ⁢ C o ⁢ x ⁢ t ⁢ W t ⁢ L b μ b ⁢ C oxb ⁢ W b ⁢ L t ) + ( 1 - m t m b ) ⁢ V G ⁢ S ⁢ b ( 2 )

the parameter with the subscript t corresponds to the first transistor MN 1 , and the parameter with the subscript b corresponds to the second transistor MN 2 . For example, the first transistor MN 1 may be a transistor whose gate oxide layer is thinner than that of the second transistor MN 2 .

In the above equation (2), the first term represents the part of the reference voltage that is inversely proportional to the absolute temperature (complementary-to-absolute-temperature, CTAT), which means that the reference voltage is adjusted by the difference value of the threshold voltages of the first transistor MN 1 and the second transistor MN 2 .

Since the first transistor MN 1 and the second transistor MN 2 used are the same conductive type of transistors, the temperature dependence of the threshold voltage difference between the two transistors can be effectively reduced when the temperature of the operating environment is changed. Therefore, the temperature effect of the reference voltage is reduced. In addition, when the manufacturing process is changed, the threshold voltages of the first transistor MN 1 and the second transistor MN 2 are shifted in the same direction which the influence of the manufacturing process on the reference voltage can be reduced. The difference of the threshold voltages between the first transistor MN 1 and the second transistor MN 2 is only 29 mV in the three process ranges of fast (FF), typical (IT) and slow (SS) under the verification of simulation.

In the above equation (2), the second term represents the part of the reference voltage that is proportional to the absolute temperature (proportional-to-absolute-temperature, PTAT), which means that the reference voltage can be adjusted by the thermal voltage V T . The parameters W and L respectively represent the size parameters of the transistor, that is, channel width and channel length.

In the above equation (2), the third term represents the slope parameter m related to the size of the first transistor MN 1 and the second transistor MN 2 in the reference voltage. The first-order linear compensation of temperature is achieved by using an appropriate transistor ratio. Therefore, this architecture can be operated stably in the temperature range of −40° C. to 130° C.

According to an embodiment of the present disclosure, the current source circuit 100 comprises a cascode current mirror circuit. By this architecture, a more stable current source IP 5 can be generated at the first feedback node F 1 , and then the current sources IP 3 and IP 1 can be duplicated through the cascode current mirror circuit.

According to an embodiment of the present disclosure, the cascode current mirror circuit comprises a first output circuit, a second output circuit, and an input circuit. The first output circuit comprises a first P-type transistor MP 1 and a second P-type transistor MP 2 . The source of the first P-type transistor MP 1 is connected to the operating voltage VDD, and the second P-type transistor MP 2 is connected in series to the first P-type the transistor MP 1 , and the second P-type transistor MP 2 outputs the current source IP 1 to the first stacked diode-connected circuit.

The second output circuit comprises a third P-type transistor MP 3 and a fourth P-type transistor MP 4 . The source of the third P-type transistor MP 3 is connected to the operating voltage VDD, and the fourth P-type transistor MP 4 is connected in series to the third P-type transistor MP 3 , and the fourth P-type transistor MP 4 outputs the current source IP 1 to the second stacked diode-connected circuit, and the gate of the fourth P-type transistor MP 4 is connected to the gate of the second P-type transistor MP 2 .

The input circuit comprises a fifth P-type transistor MP 5 , a sixth P-type transistor MP 6 , a fifth N-type transistor MN 5 , and a sixth N-type transistor MN 6 . The drain of the fifth P-type transistor MP 5 is connected to the operating voltage VDD, and the gate of the fifth P-type transistor MP 5 is connected to the gate of the third P-type transistor MP 3 and the gate of the first P-type transistor MP 1 .

The gate of the sixth P-type transistor MP 6 is connected to the second feedback node F 2 , and the serial connection node of the sixth P-type transistor MP 6 and the fifth P-type transistor MP 5 is connected to the gate of the fifth P-type transistor MP 5 .

The gate of the fifth N-type transistor MN 5 is connected to the gate of the fourth transistor MN 4 .

The gate of the sixth N-type transistor MN 6 is connected to the first feedback node F 1 , and the serial connection node of the sixth N-type transistor MN 6 and the fifth N-type transistor MN 5 is connected to the third feedback node F 3 , and the third feedback node F 3 is connected to the gate of the fourth P-type transistor MP 4 and connected to the gate of the second P-type transistor MP 2 (that is, there is a replicated feedback path that feeds back to the fourth P-type transistor MP 4 and the second P-type transistor MP 2 ), and the source of the sixth N-type transistor MN 6 is connected to the ground GND. The path between connection node of the fifth N-type transistor MN 5 and the sixth N-type transistor MN 6 , the gate of the second P-type transistor MP 2 , and the gate of the fourth P-type transistor MP 4 is formed as an architecture of the feedback path. Similar, the path between the connection node of the first transistor MN 1 and the second transistor MN 2 , and the gate of the sixth P-type transistor MP 6 is also formed a feedback path.

In the above circuit architecture, the feedback of the self-biased voltage ensures the startup of the reference voltage generating circuit 10 . When the power is turned on, the two points (A and B) in the low voltage state will quickly starts up the circuit, and then the voltage of B point will quickly decrease through the cross-coupling loop, thus avoiding the zero current state. In the design of the low power consumption reference voltage generating circuit 10 , a traditional startup circuit will increase additional power consumption. The use of a cross-coupling loop not only eliminates the startup circuit, but also prevents the leakage current generated by the startup circuit from affecting the temperature coefficient performance of the circuit.

Based on the design in FIG. 1 , the following reference voltage equation (3) can be obtained:

V R ⁢ E ⁢ F = ( m N ⁢ ⁢ 1 m N ⁢ ⁢ 2 ⁢ V THN ⁢ ⁢ 2 - V THN ⁢ ⁢ 1 ) + m N ⁢ ⁢ 1 ⁢ V T ⁢ ln ⁡ ( 2 ⁢ μ N ⁢ ⁢ 1 ⁢ C oxN ⁢ ⁢ 1 ⁢ W N ⁢ ⁢ 1 ⁢ L N ⁢ ⁢ 2 μ N ⁢ ⁢ 2 ⁢ C oxN ⁢ ⁢ 2 ⁢ W N ⁢ ⁢ 2 ⁢ L N ⁢ ⁢ 1 ) + ( m N ⁢ ⁢ 3 m N ⁢ ⁢ 4 ⁢ V THN ⁢ ⁢ 4 - V THN ⁢ ⁢ 3 ) + m N ⁢ ⁢ 3 ⁢ V T ⁢ ln ⁡ ( μ N ⁢ ⁢ 3 ⁢ C oxN ⁢ ⁢ 3 ⁢ W N ⁢ ⁢ 3 ⁢ L N ⁢ ⁢ 4 μ N ⁢ ⁢ 4 ⁢ C oxN ⁢ ⁢ 4 ⁢ W N ⁢ ⁢ 4 ⁢ L N ⁢ ⁢ 3 ) ( 3 )

the parameters with subscripts N 1 , N 2 , N 3 , and N 4 correspond to the first transistor MN 1 , the second transistor MN 2 , the third transistor MN 3 , and the fourth transistor MN 4 , respectively.

According to the above equation (3), a reference voltage independent of temperature can be obtained by adjusting the size of each transistor and the threshold voltage difference.

The power supply sensitivity is described below:

The suppression capability of the power supply variation is one of the decisive parameters for the performance of the reference voltage generating circuit. However, while reducing power consumption, it usually decreases the capability of suppressing the changes of the power supply.

Due to the influence of the slope parameter m, the sensitivity of the power supply is not zero, and the output resistance of the current source needs to be increased to increase the influence of the power supply change. Therefore, the above-mentioned stacked cascade current mirror architecture is used to increase the impedance value. The following describes the equation (4) of the power supply rejection ratio (power supply rejection ratio, PSRR):

the parameter g m is transconductance. In actual situations, the sizes of the two transistors have different designs for temperature compensation, it results in that the slope parameters m of the two transistors are different. That is, the parameter g mt of the first transistor MN 1 will not be equal to the parameter g mb of the second transistor MN 2 , so that the power supply sensitivity is not zero under actual conditions.

It can select the slope parameters of the first transistor MN 1 and the second transistor MN 2 . For example, the slope parameter of the first transistor MN 1 is 90% of the slope parameter of the second transistor MN 2 . When the resistance of the above-mentioned current source is the same, the sensitivity of the power supply can be increased by 20 dB.

The following describes the improvement of the bandwidth of the power supply rejection ratio and the start-up time:

According to an embodiment of the present disclosure, the reference voltage generating circuit 10 further comprises a first coupling capacitor C 1 and a second coupling capacitor C 2 , Two terminals of the first coupling capacitor C 1 are respectively coupled between the gate of the fifth N-type transistor MN 5 and a serial connection node of the fifth N-type transistor MN 5 and the sixth P-type transistor MP 6 .

Two terminals of the second coupling capacitor C 2 are respectively coupled between the serial connection node of the fifth N-type transistor MN 5 and the sixth P-type transistor MP 6 and the serial connection node of the first output circuit and the first stacked diode-connected circuit.

It can be seen from FIG. 1 , between node C to node D and node E, the first coupled capacitor C 1 and the second coupled capacitor C 2 are added respectively. It can couple the rapid changing signal of the power supply voltage to the core circuit 200 so that the start-up time can be effectively shorten to less than 1 millisecond.

Refer to FIG. 2 , which is a simplified schematic diagram of a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure. As shown in the drawings, the resistor RS in the current source circuit 100 corresponds to the fifth P-type transistor MP 5 and the sixth P-type transistor MP 6 , and the resistor R 1 corresponds to the third P-type transistor MP 3 and the fourth P-type transistor, and the resistor R 2 corresponds to the first P-type transistor MP 1 and the second P-type transistor MP 2 , and C 1 and C 2 are equivalent capacitors corresponding to R 1 and R 2 .

Next, from a more intuitive point of view, the first coupled capacitor C 1 and the second coupled capacitor C 2 respectively create leading and lagging paths from the power supply to VREF, and it can be seen from FIG. 2 that the paths are symmetrical and the attenuation of power disturbances is similar. Therefore, when the second coupled capacitor C 2 changes with respect to the first coupled capacitor C 1 , the attenuation of the power disturbance is changed by the two capacitors, which affects the bandwidth of the power supply suppression ratio.

According to an embodiment of the present disclosure, the reference voltage generating circuit 10 further comprises a third coupled capacitor C 3 . Two terminals of the third coupled capacitor C 3 are coupled between the gate of the fourth P-type transistor MP 4 and the ground terminal GND. The high voltage terminal of the third coupled capacitor C 3 is connected to the third feedback node F 3 , and coupled to the serial connection node of the sixth N-type transistor MN 6 and the fifth N-type transistor MN 5 through the third feedback node F 3 .

The addition of the aforementioned third coupled capacitor C 3 , for example, it can have 1.2 pico-farads (pF), which will effectively improve the stability of the transition voltage of node B after the reference voltage generating circuit 10 is activated.

According to an embodiment of the present disclosure, the capacitance value of the second coupled capacitor C 2 is greater than the capacitance value of the first coupled capacitor C 1 . For example, the first coupled capacitor C 1 is 45 femto-farads (fF), and the second coupled capacitor C 2 is 450 femto-farads (fF).

The different capacitances between the first coupled capacitor C 1 and the second coupled capacitor C 2 causes a phase difference of attenuation of the power disturbance between the two capacitors, and the characteristic of notch is generated when the phase difference is 180 degrees. It extends the bandwidth of the power supply suppression ratio to 100 Hz, and suppresses interference from the 50 to 60 Hz frequency band of commercial power supplies.

According to an embodiment of the present disclosure, the third transistor MN 3 and the fourth transistor MN 4 have different sizes and threshold voltages. In this way, if it wants to output different reference voltages VREF under the same operating voltage VDD, only the threshold voltage and size of the first transistor MN 1 and the second transistor MN 2 need to be adjusted.

According to an embodiment of the present disclosure, the reference voltage generating circuit 10 does not comprise a bipolar junction transistor (i.e. lacking of the BJT). The circuit architecture disclosed by an embodiment of the present disclosure can be enabled by 180 nm CMOS process, and the total chip area is only 5900 square micrometers, and the power consumption is 1.8 nano-Watts. Therefore, the chip area and power consumption can be reduced.

Refer to FIG. 3 , which is a schematic diagram of a reference voltage generating circuit of a stacked diode-connected architecture according to an embodiment of the present disclosure. As shown in the drawings, an embodiment of the present disclosure also provides a reference voltage generating circuit 20 with a stacked diode-connected architecture, which comprises a current source circuit ID, a first transistor MN 1 and a second transistor MN 2 . An input terminal of the current source circuit ID is connected to an operating voltage VDD, and the current source circuit ID outputs a current source.

The drain of the first transistor MN 1 is connected to an output terminal of the current source circuit ID.

The second transistor MN 2 is connected in series to the first transistor MN 1 , and the source of the second transistor MN 2 is connected to the ground terminal GND, and the serial connection node of the second transistor MN 2 and the first transistor MN 1 outputs a reference voltage VREF, and the gates of the second transistor MN 2 and the first transistor MN 1 are connected to the output terminal of the current source circuit ID.

The above-mentioned reference voltage generating circuit 20 is a modification of an embodiment of the reference voltage generating circuit 10 and more simplified circuit architecture can be obtained.

According to an embodiment of the present disclosure, the first transistor MN 1 and the second transistor MN 2 are of the same type. Similar to the foregoing embodiments, when the temperature is changed, the threshold voltages of the same type of transistor are changed in the same direction, and the impact on the output voltage can be effectively reduced.

Refer to FIG. 4 , which is a low power consumption sensor comprising a self-biased and capacitive-coupled reference voltage generating circuit according to an embodiment of the present disclosure. As shown in the drawings, an embodiment of the present disclosure also provides a low power consumption sensor 400 , which is suitable for battery-less IoT devices. The low power consumption sensor comprises the reference voltage generating circuit 10 or 20 described above.

According to an embodiment of the present disclosure, the low power consumption sensor may be a patch sensor or a biomedical implanter. Such sensors or implanters are mostly battery-less application devices, which require low power consumption, small area, and calibration-free reference voltage generating circuits 10 or 20 provided from an embodiments of the present disclosure.

The above description is only exemplary, not restrictive. Any equivalent modification or alteration that does not deviate from the spirit and scope of the present disclosure shall be included in the scope of the appended patent application claims.