Resonant DC-DC converter

This application claims priority to China Patent Application No. 201910860676.0, filed on Sep. 11, 2019. The entire contents of the above-mentioned patent applications are incorporated herein by reference for all purposes.

The present disclosure relates to a power conversion system, and more particularly to a power conversion system capable of turning off the switches at zero current and turning on the switches at zero voltage.

In conventional applications of non-isolated step-down DC-DC converters with high output current, a two-phase buck circuit with parallel configuration is employed. Please refer to FIG. 1 and FIG. 2 . FIG. 1 is a schematic circuit diagram illustrating a symmetric two-phase buck circuit with extended duty ratio. FIG. 2 is a schematic circuit diagram illustrating an asymmetric two-phase buck circuit with extended duty ratio. These two buck circuits both utilize capacitors to double the duty ratio, thereby reducing the voltage stress on switches and the effective value of switching current greatly. Consequently, the voltage seconds on inductors are reduced, and the size of inductors is greatly decreased. In addition, the switches in every phase of the buck circuit are driven by two signals having 180 degrees out of phase with respect to each other. Therefore, the frequency of the output voltage ripple is doubled, and the output voltage ripple is greatly reduced.

However, the two conventional circuits shown in FIG. 1 and FIG. 2 are both hard switching circuits, which cause the high switching loss and low switching frequency of the DC-DC converter. Moreover, the power conversion density of the DC-DC converter cannot be further improved.

Therefore, there is a need of providing a power conversion system to obviate the drawbacks encountered from the prior arts.

It is an objective of the present disclosure to provide a power conversion system. A resonance is generated among the resonant capacitors and inductors through controlling switches on and off. Accordingly, the switches can be turned off at zero current and can be turned on at zero voltage. Consequently, the switching loss is greatly reduced, and the energy conversion efficiency is improved.

It is another objective of the present disclosure to provide a power conversion system. With regard to high-power applications, a plurality of power conversion circuits connected in parallel and interleaved with each other is utilized to enlarge the load capacity of the power conversion system. Through the phase shifts among the control signals, the plurality of power conversion circuits are allowed to be connected in parallel and to be interleaved with each other. Meanwhile, the current ripples at the input and output sides of the power conversion system are reduced. Consequently, the filtering components with small size can be employed, and the size of the power conversion system is decreased.

In accordance with an aspect of the present disclosure, there is provided a power conversion system. The power conversion system includes N power conversion circuits, wherein N is an integer larger than or equal to 1. Each of the N power conversion circuits includes an input, an output, two switching power conversion units and at least one resonant capacitor. The input is configured to receive an input voltage. The output is configured to output an output voltage. The two switching power conversion units are cascaded coupled between the input and the output. Each of the two switching power conversion units includes a plurality of switches and a winding. A part of the plurality of switches is connected in series with each other and is coupled among the inputs of the N power conversion circuits. The plurality of switches operates periodically according to a switching period. The two windings of the two switching power conversion units have the same turn. A dotted terminal of one winding is electrically coupled to an undotted terminal of the other winding, and the dotted and undotted terminals of the two windings are coupled to the output. The two windings are magnetically coupled to each other to form a transformer. The at least one resonant capacitor is serially coupled between the input and the output. The resonant capacitor has a DC voltage proportional to the input voltage. In one switching period, the resonant capacitor stores an energy or outputs the stored energy to the output as the corresponding switch of the switching power conversion unit is turned on or off. A resonance is generated between the resonant capacitor and a resonant inductor, and the generated resonance has a resonant frequency and a resonant period.

FIG. 1 and FIG. 2 are schematic circuit diagrams illustrating conventional two-phase buck circuits with extended duty ratio;

FIG. 3 is a schematic circuit diagram illustrating a power conversion circuit according to a first embodiment of the present disclosure;

FIG. 4 is a schematic oscillogram showing the waveforms of currents in FIG. 3 corresponding to the variation of the switch state, wherein the resonant period is equal to the switching period;

FIG. 5A and FIG. 5B are schematic simplified circuit diagrams of the power conversion circuit of FIG. 3 during different periods of time in the switching period;

FIG. 6 is a schematic oscillogram showing the waveforms of currents in FIG. 3 corresponding to the variation of the switch state, wherein the resonant period is larger than the switching period;

FIG. 7 and FIG. 8 are schematic circuit diagrams illustrating different variants of the power conversion circuit of FIG. 3 ;

FIG. 9 is a schematic circuit diagram illustrating a power conversion circuit according to a second embodiment of the present disclosure;

FIG. 10 is a schematic oscillogram showing the waveforms of currents in FIG. 9 corresponding to the variation of the switch state;

FIG. 11A and FIG. 11B are schematic simplified circuit diagrams of the power conversion circuit of FIG. 9 during different periods of time in the switching period;

FIG. 12 and FIG. 13 are schematic circuit diagrams illustrating different variants of the power conversion circuit of FIG. 9 ;

FIG. 14 is a schematic circuit diagram illustrating a power conversion system according to the first embodiment of the present disclosure;

FIG. 15 is a schematic circuit diagram illustrating a power conversion system according to the second embodiment of the present disclosure;

FIG. 16 is a schematic circuit diagram illustrating the power conversion system including a plurality of power conversion circuits connected in parallel and interleaved with each other according to the first embodiment of the present disclosure; and

FIG. 17 is a schematic circuit diagram illustrating the power conversion system including a plurality of power conversion circuits connected in parallel and interleaved with each other according to the second embodiment of the present disclosure.

The present disclosure will now be described more specifically with reference to the following embodiments. It is to be noted that the following descriptions of preferred embodiments of this disclosure are presented herein for purpose of illustration and description only. It is not intended to be exhaustive or to be limited to the precise form disclosed.

FIG. 3 is a schematic circuit diagram illustrating a power conversion circuit according to a first embodiment of the present disclosure. As shown in FIG. 3 , in the first embodiment of the present disclosure, the power conversion system includes one power conversion circuit 1 a , and the power conversion circuit 1 a has a symmetric circuit configuration. The power conversion circuit 1 a includes an input, an output, two cascaded switching power conversion units and two resonant capacitors C 1 and C 2 . The input is configured to receive an input voltage Vin, the output is configured to output an output voltage V O , and the ratio of the input voltage Vin to the output voltage V O is 4:1. The two cascaded switching power conversion units are coupled between the input and the output. In this embodiment, one of the two switching power conversion units includes a first switch S 11 , a second switch S 22 , a third switch S 32 and a winding T 1 . The first switch S 11 , the second switch S 22 and the third switch S 32 operate periodically according to a switching period and have a switching frequency. One terminal of the winding T 1 is electrically connected between the second switch S 22 and the third switch S 32 , and the other terminal of the winding T 1 is electrically connected to a positive electrode of the output. The other switching power conversion unit includes a first switch S 12 , a second switch S 21 , a third switch S 31 and a winding T 2 . The first switch S 12 , the second switch S 21 and the third switch S 31 operate periodically according to a switching period and have a switching frequency. One terminal of the winding T 2 is electrically connected between the second switch S 21 and the third switch S 31 , and the other terminal of the winding T 2 is electrically connected to the positive electrode of the output. One terminal of the resonant capacitor C 1 is electrically connected between the first switch S 11 and the second switch S 21 , and the other terminal of the resonant capacitor C 1 is electrically connected between the second switch S 22 and the third switch S 32 . One terminal of the resonant capacitor C 2 is electrically connected between the first switch S 12 and the second switch S 22 , and the other terminal of the resonant capacitor C 2 is electrically connected between the second switch S 21 and the third switch S 31 .

The first switch S 11 , the second switch S 22 and the third switch S 31 are controlled on and off by the same control signal. The first switch S 12 , the second switch S 21 and the third switch S 32 are controlled on and off by the same control signal. These two control signals are 180 degrees out of phase with respect to each other.

The turns of the windings T 1 and T 2 are the same. The dotted terminal of the winding T 1 and the undotted terminal of winding T 2 are connected together to form the positive electrode of the output of the power conversion circuit 1 a , or the undotted terminal of the winding T 1 and the dotted terminal of winding T 2 are connected together to form the positive electrode of the output of the power conversion circuit 1 a . The windings T 1 and T 2 are wound around the same magnetic core pillar to form a close-coupling transformer T. There is an air gap on the magnetic core of the transformer T. The resonance is generated among the resonant capacitors C 1 and C 2 and the resonant inductor through controlling the plurality of switches on and off. Accordingly, the switches can be turned off at zero current and can be turned on at zero voltage. The resonant inductor is for example but not limited to a leakage inductance of the transformer T or a parasitic inductance of wires. Further, for turning the switches on at zero voltage, the inductance value of the windings T 1 and T 2 may be controlled to make the magnetizing inductance of windings small enough and make the magnetizing current large enough.

FIG. 4 is a schematic oscillogram showing the waveforms of currents in FIG. 3 corresponding to the variation of the switch state. FIG. 5A and FIG. 5B are schematic simplified circuit diagrams of the power conversion circuit of FIG. 3 during different periods of time in the switching period. In particular, FIG. 5A is a schematic equivalent circuit diagram of the power conversion circuit while the switches S 11 , S 22 and S 31 being on, and FIG. 5B is a schematic equivalent circuit diagram of the power conversion circuit while the switches S 12 , S 21 and S 32 being on. Regarding the equivalent circuits of the transformer T shown in FIG. 5A and FIG. 5B , since the dotted terminal of the winding T 1 and the undotted terminal of the winding T 2 are electrically connected, the windings T 1 and T 2 can be equivalent to two ideal windings T 1 ′ and T 2 ′ connected in series. The equivalent magnetizing inductance Lm 1 is connected to the two ends of the serial ideal windings T 1 ′ and T 2 ′ in parallel. The equivalent leakage inductance is divided into the resonant inductors Lr 1 and Lr 2 , and the resonant inductors Lr 1 and Lr 2 are serially connected to two ends of the parallel branch, which includes the equivalent magnetizing inductance Lm 1 and the serial ideal windings T 1 ′ and T 2 ′, respectively. As shown in FIGS. 4, 5A and 5B , D represents the duty ratio, Ts represents the switching period, iLr 1 represents the resonant current flowing through the resonant inductor Lr 1 , and iLr 2 represents the resonant current flowing through the resonant inductor Lr 2 . The resonant current iLr 1 includes a load current io 1 and a magnetizing current iLm 1 . The resonant current iLr 2 includes a load current io 2 and the magnetizing current iLm 1 . The magnetizing current iLm 1 is determined by the voltage and magnetizing inductance Lm 1 applied on the two ends of the windings T 1 and T 2 . The resonant current iLr 1 is generated from the resonance between the resonant capacitor Cb and the resonant inductor Lr 1 , and the resonant current iLr 2 is generated from the resonance between the resonant capacitor Cb and the resonant inductor Lr 2 .

Cb=C 1+ C 2  (1),

Lr 1= Lk 1+ Leq 1  (2),

Lr 2= Lk 2+ Leq 2  (3),

where Lk 1 and Lk 2 are the leakage inductance of the transformer T, and Leq 1 and Leq 2 are the parasitic inductance of wires or/and the additional serial inductor. Since the dotted terminal of the winding T 1 and the undotted terminal of the winding T 2 are electrically connected, the turns of winding T 1 and T 2 are same, and the load currents flowing through the ideal winding T 1 ′ and T 2 ′ are equal, i.e., io 1 =io 2 .

During a period from the time t 0 to the time t 1 , the first switch S 11 , the second switch S 22 and the third switch S 31 are in the “on” state, and the first switch S 12 , the second switch S 21 and the third switch S 32 are in the “off” state. The equivalent circuit of the power conversion circuit 1 a in this period is shown in FIG. 5A . The resonant capacitor C 1 is charged by the input voltage Vin, and the resonant capacitor C 2 is discharged to the output. The voltage vLm 1 on the two ends of the equivalent magnetizing inductance Lm 1 is equal to Vin/2. The magnetizing current iLm 1 is determined by the voltage vLm 1 and the equivalent magnetizing inductance Lm 1 . A resonance is generated among the resonant inductors Lr 1 and Lr 2 and the resonant capacitor Cb, and the resonant currents iLr 1 and iLr 2 are generated accordingly. In this embodiment, when the resonant currents iLr 1 and iLr 2 equal the magnetizing currents iLm 1 and −iLm 1 respectively, i.e., when the load currents io 1 and io 2 both equal zero, the first switch S 11 , the second switch S 22 and the third switch S 31 are turned off for realizing the zero current switching (ZCS) of switches. Consequently, the turning-off loss of switches is reduced, and the energy conversion efficiency of the power conversion circuit 1 a is improved. In addition, in this period, the terminal voltage of the first switch S 12 equals Vin/2, the terminal voltage of the second switch S 21 equals Vin, and the terminal voltage of the third switch S 32 equals Vin/2. The voltage on the two ends of the resonant capacitors C 1 and C 2 equals a DC voltage superposed by an AC resonant voltage, and the typical value of the DC voltage equals Vin/2. Taking the parameter distribution of components into consideration, the ratio value of the DC voltage to the input voltage is between 0.4 and 0.6. The amplitude of the AC resonant voltage is determined by the resonant inductance value, the resonant capacitance value, the switching frequency and the magnitude of load.

During a period from the time t 1 to the time t 2 , all the switches are turned off. The magnetizing current iLm 1 flowing through the windings T 1 and T 2 flows continuously for extracting the charges on the parasitic capacitors of the first switch S 12 , the second switch S 21 and the third switch S 32 , thereby reducing the terminal voltages of the first switch S 12 , the second switch S 21 and the third switch S 32 . When the terminal voltage of the first switch S 12 , the second switch S 21 or the third switch S 32 is reduced to be smaller than 50% of the corresponding initial voltage (i.e., the terminal voltage of the switch at the time t 1 ), the first switch S 12 , the second switch S 21 and the third switch S 32 are turned on. Consequently, the turning-on loss of switches is reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 a are improved. In another embodiment, the inductance values of the windings T 1 and T 2 may be controlled to make the equivalent magnetizing inductance Lm 1 of the windings T 1 and T 2 small enough and make the magnetizing current iLm 1 flowing through the magnetizing inductance Lm 1 large enough. Accordingly, the charges on the parasitic capacitors of the first switch S 12 , the second switch S 21 and the third switch S 32 can be extracted exhaustively, and the terminal voltages of the first switch S 12 , the second switch S 21 and the third switch S 32 are decreased to zero. Therefore, when the body diodes of the first switch S 12 , the second switch S 21 and the third switch S 32 are on, the first switch S 12 , the second switch S 21 and the third switch S 32 are turned on, thus the zero voltage switching (ZVS) of switches are realized. The turning-on loss of switches can be further reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 a are improved.

During a period from the time t 2 to the time t 3 , the first switch S 12 , the second switch S 21 and the third switch S 32 are in “on” state and the first switch S 11 , the second switch S 22 and the third switch S 31 are in “off” state. The equivalent circuit of the power conversion circuit 1 a in this period is shown in FIG. 5B . The resonant capacitor C 2 is charged by the input voltage Vin, and the resonant capacitor C 1 is discharged to the output. The voltage vLm 1 on the two ends of the equivalent magnetizing inductance Lm 1 is equal to −Vin/2. The magnetizing current iLm 1 is determined by the voltage vLm 1 and the equivalent magnetizing inductance Lm 1 . A resonance is generated among the resonant inductors Lr 1 and Lr 2 and the resonant capacitor Cb, and the resonant currents iLr 1 and iLr 2 are generated accordingly. In this embodiment, when the resonant currents iLr 1 and iLr 2 equal the magnetizing currents −iLm 1 and iLm 1 , i.e., when the load currents io 1 and io 2 both equal zero, the first switch S 12 , the second switch S 21 and the third switch S 32 are turned off for realizing the zero current switching (ZCS) of switches. Consequently, the turning-off loss of switches is reduced, and the energy conversion efficiency of the power conversion circuit 1 a is improved. In addition, in this period, the terminal voltage of the first switch S 11 equals Vin/2, the terminal voltage of the second switch S 22 equals Vin, and the terminal voltage of the third switch S 31 equals Vin/2. The voltage on the two ends of the resonant capacitors C 1 and C 2 equals a DC voltage superposed by an AC resonant voltage, and the typical value of the DC voltage equals Vin/2. Taking the parameter distribution of components into consideration, the ratio value of the DC voltage to the input voltage is between 0.4 and 0.6. The amplitude of the AC resonant voltage is determined by the resonant inductance value, the resonant capacitance value, the switching frequency and the magnitude of load.

During a period from the time t 3 to the time t 4 , all the switches are turned off. The magnetizing current iLm 1 flowing through the windings T 1 and T 2 flows continuously for extracting the charges on the parasitic capacitors of the first switch S 11 , the second switch S 22 and the third switch S 31 , thereby reducing the terminal voltages of the first switch S 11 , the second switch S 22 and the third switch S 31 . In an embodiment, when the terminal voltage of the first switch S 11 , the second switch S 22 or the third switch S 31 is reduced to be smaller than 50% of the corresponding initial voltage (i.e., the terminal voltage of the switch at the time t 3 ), the first switch S 11 , the second switch S 22 and the third switch S 31 are turned on, and the next switching period begins. Consequently, the turning-on loss of switches is reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 a are improved. In another embodiment, the inductance values of the windings T 1 and T 2 may be controlled to make the equivalent magnetizing inductance Lm 1 of the windings T 1 and T 2 small enough and make the magnetizing current iLm 1 flowing through the magnetizing inductance Lm 1 large enough. Accordingly, the charges on the parasitic capacitors of the first switch S 11 , the second switch S 22 and the third switch S 31 can be extracted exhaustively, and the terminal voltages of the first switch S 11 , the second switch S 22 and the third switch S 31 are decreased to zero. Therefore, when the body diodes of the first switch S 11 , the second switch S 22 and the third switch S 31 are on, the first switch S 11 , the second switch S 22 and the third switch S 31 are turned on, thus the zero voltage switching (ZVS) of switches are realized. The turning-on loss of the switches can be further reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 a are improved.

In above embodiments, during the period from the time t 0 to the time t 1 and the period from the time t 2 to the time t 3 , the resonant currents iLr 1 and iLr 2 flows through the resonant inductors Lr 1 and Lr 2 simultaneously, and the frequency of the resonant currents iLr 1 and iLr 2 is equal to the resonant frequency. During the period from the time t 1 to the time t 2 and the period from the time t 3 to the time t 4 , all the switches are turned off. The magnetizing current iLm 1 flowing through the windings T 1 and T 2 charges and discharges the parasitic capacitors of the switches, and the frequency of the magnetizing current iLm 1 is equal to the switching frequency. In this embodiment, the loss of turning off the switches are reduced through the zero current switching, and a resonant period of the resonance among the resonant capacitors C 1 and C 2 and the resonant inductors Lr 1 and Lr 2 is substantially equal to the switching period.

In another embodiment, the capacitance value of the resonant capacitors C 1 and C 2 is large, and the inductance value of the resonant inductors Lr 1 and Lr 2 is small. Therefore, when the resonant currents iLr 1 and iLr 2 are larger than the magnetizing currents iLm 1 and −Lm 1 respectively, i.e., when the load currents io 1 and io 2 are larger than zero, the corresponding switches can be turned off. Although the turning-off current is larger than zero, the loss caused by the non-zero current switching can be ignored actually due to the small inductance values of the resonant inductors Lr 1 and Lr 2 . However, for taking the loss and the energy conversion efficiency into consideration, the switching period Ts is preferable to be larger than or equal to 0.5 times of resonant period Tr. The equation for calculating the turning-off loss is shown as follows.

where P Lr is the turning-off loss, I Lr is the resonant current iLr 1 or iLr 2 while turning off, I Lm is the magnetizing current while turning off, and fsw is the switching frequency of the switches of the power conversion circuit 1 a.

Since the inductance value of the resonant inductors Lr 1 and Lr 2 is small, the turning-off loss P Lr is small, which would not affect the conversion efficiency of the power conversion circuit 1 a . The corresponding oscillogram is shown in FIG. 6 . At the time t 1 , the resonant current iLr 1 is larger than the magnetizing current iLm 1 , the resonant current iLr 2 is larger than the magnetizing current −iLm 1 , and the first switch S 11 , the second switch S 22 and the third switch S 31 are turned off at this moment. At the time t 3 , the resonant current iLr 1 is larger than the magnetizing current −iLm 1 , the resonant current iLr 2 is larger than the magnetizing current iLm 1 , and the first switch S 12 , the second switch S 21 and the third switch S 32 are turned off at this moment. In this embodiment, the switching period is smaller than the resonant period.

In an embodiment, as shown in FIG. 7 , the third switches S 31 and S 32 can be replaced by the diodes D 1 and D 2 respectively, and the diodes D 1 and D 2 work as free-wheeling diodes. The switching period is smaller than or equal to the resonant period. The equivalent circuit and current waveforms of this embodiment can be derived from that of the above-mentioned embodiments, thus are omitted herein.

In above embodiments, the resonant inductor includes the leakage inductance generated from the coupling between the two windings T 1 and T 2 of the transformer T and the parasitic inductance of wires. Taking the switching period and the capacitance value of the resonant capacitors C 1 and C 2 into consideration, the coupling coefficient of the windings T 1 and T 2 is preferably but not limited to be larger than 0.9. In addition, in an embodiment, the resonant inductor includes an additional inductor. As shown in FIG. 8 , the additional inductor may be disposed at the positions A, B and C. In particular, one additional inductor is disposed at the position A, or two additional inductors with the same inductance value are disposed at two positions B respectively, or two additional inductors with the same inductance value are disposed at two positions C respectively. Moreover, a plurality of additional inductors is disposed in at least two positions of the positions A, B and C.

FIG. 9 is a schematic circuit diagram illustrating a power conversion circuit according to a second embodiment of the present disclosure. As shown in FIG. 9 , in the second embodiment of the present disclosure, the power conversion system includes one power conversion circuit 1 b , and the power conversion circuit 1 b has an asymmetric circuit configuration. The power conversion circuit 1 b includes an input, an output, two cascaded switching power conversion units and a resonant capacitor Cb. The input is configured to receive an input voltage Vin, the output is configured to output an output voltage Vo, and the ratio of the input voltage Vin to the output voltage Vo is 4:1. The two cascaded switching power conversion units are coupled between the input and the output. In this embodiment, one of the two switching power conversion units includes a first switch M 11 , a second switch M 21 and a winding T 3 . The first switch M 11 and the second switch M 21 are coupled to each other in series. The first switch M 11 and the second switch M 21 operate periodically according to a switching period and have a switching frequency. One terminal of the winding T 3 is electrically connected between the resonant capacitor Cb and the second switch M 21 , and the other terminal of the winding T 3 is electrically connected to a positive electrode of the output. The other switching power conversion unit includes a first switch M 12 , a second switch M 22 and a winding T 4 . The first switch M 12 and the second switch M 22 are coupled to each other in series. The first switch M 12 and the second switch M 22 operate periodically according to a switching period and have a switching frequency. One terminal of the winding T 4 is electrically connected between the first switch M 12 and the second switch M 22 , and the other terminal of the winding T 4 is electrically connected to the positive electrode of the output. One terminal of the resonant capacitor Cb is electrically connected between the first switches M 11 and M 12 , and the other terminal of the resonant capacitor Cb is electrically connected between the second switch M 21 and the winding T 3 .

The first switch M 11 and the second switch M 22 are controlled on and off by the same control signal. The first switch M 12 and the second switch M 21 are controlled on and off by the same control signal. These two control signals are 180 degrees out of phase with respect to each other.

The turns of the windings T 3 and T 4 are the same. The dotted terminals of the winding T 3 and the undotted terminals of winding T 4 are connected together so as to form the positive electrode of the output of the power conversion circuit 1 b . The windings T 3 and T 4 are wound around the same magnetic core pillar to form a close-coupling transformer T. There is an air gap on the magnetic core of the transformer T. The resonance is generated between the resonant capacitor Cb and the resonant inductor through controlling the plurality of switches on and off. Accordingly, the switches can be turned on and off at zero current and can be turned on at zero voltage. The resonant inductor is for example but not limited to a leakage inductance of the transformer T or a parasitic inductance of wires. Further, for turning the switches on at zero voltage, the inductance value of the windings T 1 and T 2 may be controlled to make the magnetizing inductance small enough and make the magnetizing current large enough.

FIG. 10 is a schematic oscillogram showing the waveforms of currents in FIG. 9 corresponding to the variation of the switch state. FIG. 11A and FIG. 11B are schematic simplified circuit diagrams of the power conversion circuit of FIG. 9 during different periods of time in the switching period. In particular, FIG. 11A is a schematic equivalent circuit diagram of the power conversion circuit while the switches M 11 and M 22 being on, and FIG. 11B is a schematic equivalent circuit diagram of the power conversion circuit while the switches M 12 and M 21 being on. The equivalent circuit of the transformer T is similar to that shown in FIG. 5A and FIG. 5B , thus is omitted herein. As shown in FIGS. 10, 11A and 11B , D represents the duty ratio, Ts represents the switching period, iLr 3 represents the resonant current flowing through the resonant inductor Lr 3 , and iLr 4 represents the resonant current flowing through the resonant inductor Lr 4 . The resonant current iLr 3 includes a load current io 3 and a magnetizing current iLm 2 . The resonant current iLr 4 includes a load current io 4 and the magnetizing current iLm 2 . The magnetizing current iLm 2 is determined by the magnetizing inductance Lm 2 and the voltage applied on the two ends of the windings T 3 and T 4 . The resonant current iLr 3 is generated from the resonance between the resonant capacitor Cb and the resonant inductor Lr 3 , and the resonant current iLr 4 is generated from the resonance between the resonant capacitor Cb and the resonant inductor Lr 4 .

Lr 3= Lk 3+ Leq 3  (5),

Lr 4= Lk 4+ Leq 4  (6),

where Lk 3 and Lk 4 are the leakage inductance of the transformer T, and Leq 3 and Leq 4 are the parasitic inductance of wires or/and the additional serial inductor. Since the dotted terminals of the winding T 3 and the undotted terminals of winding T 4 that have the same turn are electrically connected together, the load currents flowing through the ideal winding T 3 ′ and T 4 ′ are equal, i.e., io 3 =io 4 .

During a period from the time t 0 to the time t 1 , the first switch M 11 and the second switch M 22 are in “on” state and the first switch M 12 and the second switch M 21 are in “off” state. The equivalent circuit of the power conversion circuit 1 b in this period is shown in FIG. 11A . The resonant capacitor Cb is charged by the input voltage Vin. The voltage vLm 2 on the two ends of the equivalent magnetizing inductance Lm 2 is equal to Vin/2. The magnetizing current iLm 2 is determined by the voltage vLm 2 and the equivalent magnetizing inductance Lm 2 . A resonance is generated among the resonant inductors Lr 3 and Lr 4 and the resonant capacitor Cb, and the resonant currents iLr 3 and iLr 4 are generated accordingly. When the resonant currents iLr 3 and iLr 4 equal the magnetizing currents iLm 2 and −iLm 2 respectively, i.e., when the load currents io 3 and io 4 both equal zero, the first switch M 11 and the second switch M 22 are turned off for realizing the zero current switching (ZCS) of switches. Consequently, the turning-off loss of switches is reduced, and the energy conversion efficiency of the power conversion circuit 1 b is improved. In addition, in this period, the terminal voltage of the first switch M 12 equals Vin/2, and the terminal voltage of the second switch M 21 equals Vin/2. The voltage on the two ends of the resonant capacitor Cb equals a DC voltage superposed by an AC resonant voltage, and the typical value of the DC voltage equals Vin/2. Taking the parameter distribution of components into consideration, the ratio value of the DC voltage to the input voltage is between 0.4 and 0.6. The amplitude of the AC resonant voltage is determined by the resonant inductance value, the resonant capacitance value, the switching frequency and the magnitude of load.

During a period from the time t 1 to the time t 2 , all the switches are turned off. The magnetizing current iLm 2 flowing through the windings T 3 and T 4 flows continuously for extracting the charges on the parasitic capacitors of the first switch M 12 and the second switch M 21 , thereby reducing the terminal voltages of the first switch M 12 and the second switch M 21 . In an embodiment, when the terminal voltage of the first switch M 12 and the second switch M 21 are reduced to be smaller than 50% of the corresponding initial voltages (i.e., the terminal voltages of the switches at the time t 1 ) respectively, the first switch M 12 and the second switch M 21 are turned on. Consequently, the turning-on loss of switches is reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 b are improved. In another embodiment, the inductance values of the windings T 3 and T 4 may be controlled to make the equivalent magnetizing inductance Lm 2 of the windings T 3 and T 4 small enough and make the magnetizing current iLm 2 flowing through the magnetizing inductance Lm 2 large enough. Accordingly, the charges on the parasitic capacitors of the first switch M 12 and the second switch M 21 can be extracted exhaustively, and the terminal voltages of the first switch M 12 and the second switch M 21 are decreased to zero. Therefore, when the body diodes of the first switch M 12 and the second switch M 21 are on, the first switch M 12 and the second switch M 21 are turned on, thus the zero voltage switching (ZVS) of switches are realized. The turning-on loss of switches can be further reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 b are improved.

During a period from the time t 2 to the time t 3 , the first switch M 12 and the second switch M 21 are in “on” state, and the first switch M 11 and the second switch M 22 are in “off” state. The equivalent circuit of the power conversion circuit 1 b in this period is shown in FIG. 11B . The resonant capacitor Cb is discharged to the output. The voltage vLm 2 on the two ends of the equivalent magnetizing inductance Lm 2 is equal to −Vin/2. The magnetizing current iLm 2 is determined by the voltage vLm 2 and the equivalent magnetizing inductance Lm 2 . The resonant capacitor Cb is discharged to the output through the winding T 4 of the transformer T. Similar to that during the period from the time t 0 to the time t 1 , a resonance is generated among the resonant inductors Lr 3 and Lr 4 and the resonant capacitor Cb, and the resonant currents iLr 3 and iLr 4 are generated accordingly. When the resonant currents iLr 3 and iLr 4 equal the magnetizing currents −iLm 2 and iLm 2 respectively, i.e., when the load currents io 3 and io 4 both equal zero, the first switch M 12 and the second switch M 21 are turned off for realizing the zero current switching (ZCS) of switches. Consequently, the turning-off loss of switches is reduced, and the energy conversion efficiency of the power conversion circuit 1 b is improved. In addition, in this period, the terminal voltage of the first switch M 11 equals Vin/2, the terminal voltage of the second switch M 22 equals Vin/2. The voltage on the two ends of the resonant capacitor Cb equals a DC voltage superposed by an AC resonant voltage, and the typical value of the DC voltage equals Vin/2. Taking the parameter distribution of components into consideration, the ratio value of the DC voltage to the input voltage is between 0.4 and 0.6. The amplitude of the AC resonant voltage is determined by the resonant inductance value, the resonant capacitance value, the switching frequency and the magnitude of load.

During a period from the time t 3 to the time t 4 , all the switches are turned off. The magnetizing current iLm 2 flowing through the windings T 3 and T 4 flows continuously for extracting the charges on the parasitic capacitors of the first switch M 11 and the second switch M 22 , thereby reducing the terminal voltages of the first switch M 11 and the second switch M 22 . In an embodiment, when the terminal voltages of the first switch M 11 and the second switch M 22 are reduced to be smaller than 50% of the initial voltages (i.e., the terminal voltages of the switch at the time t 3 ) respectively, the first switch M 11 and the second switch M 22 are turned on, and the next switching period begins. Consequently, the turning-on loss of switches is reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 b are improved. In another embodiment, the inductance values of the windings T 3 and T 4 may be controlled to make the equivalent magnetizing inductance Lm 2 of the windings T 3 and T 4 small enough and make the magnetizing current iLm 2 flowing through the magnetizing inductance Lm 2 large enough.

Accordingly, the charges on the parasitic capacitors of the first switch M 11 and the second switch M 22 can be extracted exhaustively, and the terminal voltages of the first switch M 11 and the second switch M 22 are decreased to zero. Therefore, when the body diodes of the first switch M 11 and the second switch M 22 are on, the first switch M 11 and the second switch M 22 are turned on, thus the zero voltage switching (ZVS) of switches are realized. The turning-on loss of the switches can be further reduced, and the energy conversion efficiency and power density of the power conversion circuit 1 b are improved.

During the period from the time t 0 to the time t 1 and the period from the time t 2 to the time t 3 , the resonant currents iLr 3 and iLr 4 flows through the windings T 3 and T 4 simultaneously, and the frequency of the resonant currents iLr 3 and iLr 4 is equal to the resonant frequency. During the period from the time t 1 to the time t 2 and the period from the time t 3 to the time t 4 , all the switches are turned off. The magnetizing current iLm 2 flowing through the windings T 3 and T 4 charges and discharges the parasitic capacitors of the switches, and the frequency of the magnetizing current iLm 2 is equal to the switching frequency. In this embodiment, the resonant period is substantially equal to the switching period.

In another embodiment, the switching period is smaller than the resonant period. The capacitance value of the resonant capacitor Cb is large, and the inductance value of the resonant inductor Lr is small. Therefore, when the resonant current is larger than the corresponding magnetizing current, the corresponding switches are turned off, and the turning-off loss can be ignored.

In an embodiment, as shown in FIG. 12 , the second switches M 21 and M 22 can be replaced by the diodes D 3 and D 4 , and the diodes D 3 and D 4 work as free-wheeling diodes. The switching period is smaller than or equal to the resonant period. The equivalent circuit and current waveforms of this embodiment can be derived from that of the above-mentioned embodiments, thus are omitted herein.

In above embodiments, the resonant inductor includes the leakage inductance generated from the coupling between the two windings T 3 and T 4 of the transformer T and the parasitic inductance of wires. Taking the switching period and the capacitance value of the resonant capacitor Cb into consideration, the coupling coefficient of the windings T 3 and T 4 is preferably but not limited to be larger than 0.9. In addition, in an embodiment, the resonant inductor includes an additional inductor. As shown in FIG. 13 , the additional inductor may be disposed at the positions D, E and F. In particular, one additional inductor is disposed at the position D, or two additional inductors with the same inductance value are disposed at two positions E respectively, or two additional inductors with the same inductance value are disposed at two positions F respectively. Moreover, a plurality of additional inductors is disposed in at least two positions of the positions D, E and F.

FIG. 14 is a schematic circuit diagram illustrating a power conversion system according to the first embodiment of the present disclosure. As shown in FIG. 14 , the power conversion system 2 a includes at least a symmetric power conversion circuit 1 a and a controller 10 . The controller 10 outputs two control signals PWM 1 and PWM 2 . The two control signals PWM 1 and PWM 2 are 180 degrees out of phase with respect to each other. The control signal PWM 1 is configured to control on and off of the first switch S 11 , the second switch S 22 and the third switch S 31 . The control signal PWM 2 is configured to control on and off of the first switch S 12 , the second switch S 21 and the third switch S 32 .

FIG. 15 is a schematic circuit diagram illustrating a power conversion system according to the second embodiment of the present disclosure. As shown in FIG. 15 , the power conversion system 2 b includes at least an asymmetric power conversion circuit 1 b and a controller 10 . The controller 10 outputs two control signals PWM 1 and PWM 2 . The two control signals PWM 1 and PWM 2 are 180 degrees out of phase with respect to each other. The control signal PWM 1 is configured to control on and off of the first switch M 11 and the second switch M 22 . The control signal PWM 2 is configured to control on and off of the first switch M 12 and the second switch M 21 .

In order to satisfy the requirement of high-power applications, in an embodiment, the present disclosure utilizes a plurality of power conversion circuits connected in parallel and interleaved with each other to enlarge the load capacity of the power conversion system. As shown in FIG. 16 and FIG. 17 , the power conversion systems 100 and 200 includes N symmetric power conversion circuits and N asymmetric power conversion circuits respectively, where N is an integer larger than 1. (In the embodiments shown in FIG. 1 to FIG. 15 , N equals 1, namely the power conversion system includes only one power conversion circuit). All the power conversion circuits have the same circuit configuration and substantially identical circuit parameters. The specific circuit configurations of the power conversion circuits are similar to that of the power conversion circuits 1 a and 1 b shown in above embodiments, and the detailed description thereof is omitted herein. The inputs of the N power conversion circuits are all connected in parallel, and the outputs of the N power conversion circuits are all connected in parallel. The inputs share at least one input capacitor CM after being connected in parallel, or the inputs are connected to N input capacitors CM respectively before being connected in parallel. Similarly, the outputs share at least one output capacitor Co after being connected in parallel, or the outputs are connected to N output capacitors Co respectively before being connected in parallel.

In addition, the power conversion system 100 and 200 further includes a controller 10 . As shown in the symmetric power conversion circuit of FIG. 16 , the controller 10 outputs N sets of control signals (PWM 11 , PWM 12 ), (PWM 21 , PWM 22 ) . . . (PWMN 1 , PWMN 2 ). The control signals PWM 11 , PWM 21 . . . PWMN 1 are 360/2N degree out of phase with respect to each other in sequence. The control signals PWM 11 and PWM 12 are utilized to control the power conversion circuit 101 , the control signals PWM 21 and PWM 22 are utilized to control the power conversion circuit 102 , and so forth, and the control signals PWMN 1 and PWMN 2 are utilized to control the power conversion circuit 10 N. In another embodiment, the angle of the N sets of control signals PWM 11 , PWM 21 . . . PWMN 1 being out of phase with respect to each other in sequence may be any angle between (360/2N−20) and (360/2N+20) degrees.

As shown in the asymmetric power conversion circuit of FIG. 17 , when N is odd, the controller 10 outputs Y sets of control signals (PWM 11 , PWM 12 ), (PWM 21 , PWM 22 ) . . . (PWMY 1 , PWMY 2 ). Y equals N, and the control signals PWM 11 , PWM 21 . . . PWMY 1 are 360/N degrees out of phase with respect to each other in sequence. In another embodiment, the angle of the Y sets of control signals PWM 11 , PWM 21 . . . PWMY 1 being out of phase with respect to each other in sequence may be any angle between (360/N−20) and (360/N+20) degrees. The control signals PWM 11 and PWM 12 are utilized to control the power conversion circuit 201 , the control signals PWM 21 and PWM 22 are utilized to control the power conversion circuit 202 , and so forth, and the control signals PWMY 1 and PWMY 2 are utilized to control the power conversion circuit 20 N. When N is even, the controller 10 output Y sets of control signals (PWM 11 , PWM 12 ), (PWM 21 , PWM 22 ) . . . (PWMY 1 , PWMY 2 ). Y equals N/2, and the control signals PWM 11 , PWM 21 . . . PWMY 1 are 360/N degrees out of phase with respect to each other in sequence. In another embodiment, the angle of the N sets of control signals PWM 11 , PWM 21 . . . PWMY 1 being out of phase with respect to each other in sequence may be any angle between (360/N−20) and (360/N+20) degrees. The control signals PWM 11 and PWM 12 are utilized to control the power conversion circuits 201 and 20 (Y+1), the control signals PWM 21 and PWM 22 are utilized to control the power conversion circuits 202 and 20 (Y+2), and so forth, and the control signals PWMY 1 and PWMY 2 are utilized to control the power conversion circuits 20 Y and 20 N.

The interleaved and parallel circuit configuration not only enlarge the load capacity of the power conversion system but also reduce the current ripples of the input and output of the power conversion system through controlling the controls signals to be output of phase with respect to each other. Therefore, the size of the input and output filters is decreased, the size of the power conversion system is decreased, and the power density of the power conversion system is improved.

In addition, the switches mentioned in the above embodiments may be MOS, SiC or GaN, and the resonance and the interleaved and parallel connection would not be affected.

From the above descriptions, the present disclosure provides a power conversion system. By replacing the inductors with the transformer and connecting the windings in special ways, a resonance is generated among the resonant capacitors and inductors through controlling switches on and off. Accordingly, the switches are turned off at zero current and are turned on at zero voltage. Consequently, the switching loss is greatly reduced, and the energy conversion efficiency is improved. The resonant inductor may be a leakage inductance of the transformer, a parasitic inductance of wires or an additional inductor. In addition, with regard to high-power applications, a plurality of power conversion circuits are connected in parallel and are interleaved with each other so as to enlarge the load capacity of the power conversion system. Through the phase shifts among the control signals, the plurality of power conversion circuits are allowed to be connected in parallel and to be interleaved with each other. Meanwhile, the current ripples at the input and output sides of the power conversion system are reduced. Consequently, the filtering components with small size can be employed, and the size of the power conversion system is decreased.

While the disclosure has been described in terms of what is presently considered to be the most practical and preferred embodiments, it is to be understood that the disclosure needs not be limited to the disclosed embodiment.