Filter module

This application claims priority from Japanese Patent Application No. 2019-107983 filed on Jun. 10, 2019. The content of this application is incorporated herein by reference in its entirety.

1. Field of the Disclosure

The present disclosure relates to a filter module.

2. Description of the Related Art

In the fifth generation mobile communication system (5G), Band n77 in the frequency band of 3.3 to 4.2 GHz, Band n78 in the frequency band of 3.3 to 3.8 GHz, and Band n79 in the frequency band of 4.4 to 5.0 GHz, for example, are defined. As a band pass filter used in these frequency bands, the use of a filter device including multiple parallel resonance circuits, each of which is constituted by an inductor and a capacitor, is being discussed (International Publication No. 2011/114851, for example). This filter device is implemented as the following multilayer radio-frequency filter. A predetermined conductive pattern is formed on each of plural insulating layers, and the conductive patterns formed on different insulating layers are connected to each other by via-conductors.

The filter device is mounted on a common module substrate, together with other circuit components, such as a radio-frequency switch and a low-noise amplifier. The circuit components mounted on the module substrate are electromagnetically shielded by a shield member. The inventor of this application has discovered that, when the filter device is mounted on the module substrate and is electromagnetically shielded by the shield member, the characteristics of the filter device may deviate from those of a filter device alone. This may fail to obtain target filter characteristics.

It is an object of the present disclosure to provide a filter module that can exhibit target filter characteristics in the state in which a filter device is mounted on a module substrate and is electromagnetically shielded by a shield member.

According to an aspect of the present disclosure, there is provided a filter module including a module substrate, a filter device, and a shield member. The filter device has a top surface and a bottom surface opposed to each other and first and second side surfaces opposed to each other. A ground terminal and first and second signal terminals are formed on the bottom surface. The filter device is mounted on the module substrate such that the bottom surface faces the module substrate. The shield member includes first and second side wall portions. The first and second side wall portions respectively face the first and second side surfaces so as to shield the filter device. The filter device includes plural parallel resonance circuits therein. Each of the plural parallel resonance circuits includes an inductor and a capacitor connected in parallel with each other. The inductors of the plural parallel resonance circuits are arranged in a direction parallel with the first side surface and the bottom surface. Each of the inductors of the plural parallel resonance circuits has a ground-side end portion and a signal-side end portion positioned at opposite sides. Each of the inductors extends upward from the ground-side end portion, which is electrically connected to the ground terminal, then extends in a direction from the first side surface toward the second side surface, and then extends in a direction toward the bottom surface. The signal-side end portion of the inductor of one of the plural parallel resonance circuits is electrically connected to the first signal terminal, while the signal-side end portion of the inductor of another one of the plural parallel resonance circuits is electrically connected to the second signal terminal. A gap between the first side surface and the first side wall portion is smaller than a gap between the second side surface and the second side wall portion.

According to another aspect of the present disclosure, there is provided a filter module including a module substrate, a filter device, and a shield member. The filter device has a top surface and a bottom surface opposed to each other and first and second side surfaces opposed to each other. A ground terminal and first and second signal terminals are formed on the bottom surface. The filter device is mounted on the module substrate such that the bottom surface faces the module substrate. The shield member includes first and second side wall portions. The first and second side wall portions respectively face the first and second side surfaces so as to shield the filter device. The filter device includes plural parallel resonance circuits therein. Each of the plural parallel resonance circuits includes an inductor and a capacitor connected in parallel with each other. The plural parallel resonance circuits are arranged in a direction parallel with the first side surface and the bottom surface. Each of the inductors of the plural parallel resonance circuits has a ground-side end portion and a signal-side end portion positioned at opposite sides. Each of the inductors extends upward from the ground-side end portion, which is electrically connected to the ground terminal, then extends in a direction from the first side surface toward the second side surface, and then extends in a direction toward the bottom surface. The signal-side end portion of the inductor of one of the plural parallel resonance circuits is electrically connected to the first signal terminal, while the signal-side end portion of the inductor of another one of the plural parallel resonance circuits is electrically connected to the second signal terminal. A gap between the second side surface and the second side wall portion is smaller than a gap between the first side surface and the first side wall portion, and is about 0.5 mm or larger.

As a result of arranging the positional relationship between the filter device and the shield member as described above, target filter characteristics can be obtained.

Other features, elements, characteristics and advantages of the present disclosure will become more apparent from the following detailed description of preferred embodiments of the present disclosure with reference to the attached drawings.

FIG. 1 is an equivalent circuit diagram of a filter device mounted on a filter module according to a first embodiment;

FIG. 2 is an exploded perspective view of the filter device according to the first embodiment;

FIGS. 3A, 3B, and 3C are respectively a plan view, a front view, and a bottom view of the filter device according to the first embodiment;

FIG. 4 is a block diagram of the filter module according to the first embodiment;

FIGS. 5A and 5B are respectively a perspective view and a sectional view of the filter module according to the first embodiment;

FIG. 6 is a graph illustrating the results of measuring the transmission coefficient S 21 of a circuit constituted by the filter device and a low-noise amplifier;

FIG. 7A is a schematic diagram illustrating the positional relationship between the filter device and a shield member;

FIG. 7B is a graph illustrating the simulation results of the transmission coefficient S 21 of the circuit constituted by the filter device and the low-noise amplifier;

FIG. 8A is a schematic diagram illustrating the positional relationship between the filter device and the shield member;

FIG. 8B is a graph illustrating the simulation results of the transmission coefficient S 21 of the circuit constituted by the filter device and the low-noise amplifier;

FIG. 9A is an equivalent circuit diagram of the filter device;

FIG. 9B is a graph illustrating the calculation results of the transmission coefficient S 21 in the path from a first signal terminal to a second signal terminal in the equivalent circuit diagram in FIG. 9A ;

FIG. 10A is an equivalent circuit diagram of the filter device;

FIG. 10B is a graph illustrating the calculation results of the transmission coefficient S 21 in the path from the first signal terminal to the second signal terminal in the equivalent circuit diagram in FIG. 10A ;

FIGS. 11A and 11B are graphs illustrating the simulation results obtained by using an electromagnetic simulator regarding the transmission coefficient S 21 in the path from the first signal terminal to the second signal terminal of the filter device;

FIG. 12 is a sectional view of a filter module according to a second embodiment; and

FIGS. 13A and 13B are graphs illustrating the simulation results obtained by using an electromagnetic simulator regarding the transmission coefficient S 21 in the path from a first signal terminal to a second signal terminal of a filter device according to the second embodiment.

A filter module according to a first embodiment will be discussed below with reference to FIGS. 1 through 11B .

FIG. 1 is an equivalent circuit diagram of a filter device 20 mounted on the filter module according to the first embodiment. The filter device 20 includes first through fourth parallel resonance circuits 21 , 22 , 23 , and 24 . The first parallel resonance circuit 21 includes an inductor L 1 and a capacitor C 1 connected in parallel with each other. The second parallel resonance circuit 22 includes an inductor L 2 and a capacitor C 2 connected in parallel with each other. The third parallel resonance circuit 23 includes an inductor L 4 and a capacitor C 4 connected in parallel with each other. The fourth parallel resonance circuit 24 includes an inductor L 5 and a capacitor C 5 connected in parallel with each other. The filter device 20 is a band pass filter for the 5G Band n77 frequency band, for example.

One terminal of each of the four parallel resonance circuits 21 through 24 is connected to a ground terminal TG via a common inductor L 3 . The terminal connected to the ground terminal TG will be called a ground-side terminal. The other terminals of the four parallel resonance circuits 21 through 24 positioned opposite to the ground-side terminals are called signal-side terminals. The signal-side terminal of the first parallel resonance circuit 21 is connected to a first signal terminal T 1 . A capacitor C 3 is connected between the signal-side terminal of the first parallel resonance circuit 21 and that of the second parallel resonance circuit 22 . The signal-side terminal of the fourth parallel resonance circuit 24 is connected to a second signal terminal T 2 . A capacitor C 6 is connected between the signal-side terminal of the third parallel resonance circuit 23 and that of the fourth parallel resonance circuit 24 . Series-connected two capacitors C 7 are connected between the first signal terminal T 1 and the second signal terminal T 2 . The inductors L 1 , L 2 , L 4 , and L 5 of the four parallel resonance circuits 21 through 24 are inductively coupled with each other.

FIG. 2 is an exploded perspective view of the filter device 20 according to the first embodiment. FIG. 3A is a plan view of the filter device 20 . FIG. 3B is a front view of the filter device 20 . FIG. 3C is a bottom view of the filter device 20 .

The outer shape of the filter device 20 is a substantially rectangular parallelepiped. The shape of the filter device 20 as viewed from above is substantially rectangular, and the long sides are about twice as long as the short sides. The filter device 20 is surface-mounted on a module substrate, for example. It is assumed that the surface of the filter device 20 facing the module substrate is a bottom surface 27 and the surface opposed to the bottom surface 27 is a top surface 28 . It is also assumed that two opposing side surfaces are first and second side surfaces 25 and 26 . The first and second side surfaces 25 and 26 connect the long sides of the bottom surface 27 and those of the top surface 28 . The direction from the bottom surface 27 to the top surface 28 is called a height direction.

Thirteen dielectric layers 100 A, 100 B, . . . , 100 M are stacked on each other in this order from the bottom surface 27 in the height direction. When the plural dielectric layers 100 A, 100 B, . . . , 100 M are not distinguished from each other, they are simply called the dielectric layers 100 . Low-temperature co-fired ceramic (LTCC), for example, is used for the dielectric layers 100 . Instead of an LTCC substrate, a glass substrate, a dielectric substrate made of an epoxy resin or a liquid crystal polymer may be used as the dielectric layers 100 .

At least one conductive pattern is formed on each of some of the dielectric layers 100 . Via-conductors are used for connecting the conductive patterns on different dielectric layers 100 .

The first signal terminal T 1 , the ground terminal TG, and the second signal terminal T 2 constituted by conductive patterns are formed on the outward surface (bottom surface 27 ) of the bottommost dielectric layer 100 A. Three conductive patterns 101 A, 101 B, and 101 C are formed on the third dielectric layer 100 C. A ground conductor 101 D constituted by a conductive pattern is formed on the fourth dielectric layer 100 D. The conductive pattern 101 A serves as one electrode of the capacitor C 1 , while the conductive pattern 101 C serves as one electrode of the capacitor C 4 . The two conductive patterns 101 A and 101 C formed on the third dielectric layer 100 C are opposed to the ground conductor 101 D with the dielectric layer 100 D interposed therebetween so as to form the capacitors C 1 and C 4 , respectively.

The conductive pattern 101 A serving as one electrode of the capacitor C 1 is connected to the first signal terminal T 1 by a via-conductor 102 A. The conductive pattern 101 C serving as one electrode of the capacitor C 4 is connected to the second signal terminal T 2 by a via-conductor 102 D. The ground conductor 101 D is connected to the ground terminal TG by via-conductors 102 E and 102 F, via the conductive pattern 101 B disposed on the third dielectric layer 100 C, and by via-conductors 102 B and 102 C. Parasitic inductance of the via-conductors 102 B, 102 C, 102 E, and 102 F is represented as the inductor L 3 ( FIG. 1 ).

Two conductive patterns 101 E and 101 F are formed on the fifth dielectric layer 100 E. The conductive pattern 101 E serves as one electrode of the capacitor C 2 , while the conductive pattern 101 F serves as one electrode of the capacitor C 5 . The two conductive patterns 101 E and 101 F are opposed to the ground conductor 101 D with the dielectric layer 100 E interposed therebetween so as to form the capacitors C 2 and C 5 , respectively.

Two conductive patterns 101 G and 101 H are formed on the sixth dielectric layer 100 F. The two conductive patterns 101 G and 101 H respectively are opposed to the two conductive patterns 101 E and 101 F with the dielectric layer 100 F interposed therebetween so as to form the capacitors C 3 and C 6 .

One conductive pattern 1011 is formed on the seventh dielectric layer 100 G. The two conductive patterns 101 G and 101 H formed on the sixth dielectric layer 100 F are opposed to the conductive pattern 1011 with the dielectric layer 100 G interposed therebetween so as to form the series-connected two capacitors C 7 .

Four conductive patterns 101 J, 101 K, 101 L, and 101 M are formed on the tenth dielectric layer 100 J. Four conductive patterns 101 J, 101 K, 101 L, and 101 M having the same shape as those on the tenth dielectric layer 100 J are also formed on each of the eleventh and twelfth dielectric layers 100 K and 100 L. The three conductive patterns 101 J disposed on the three dielectric layers 100 J, 100 K, and 100 L are overlaid on each other as viewed from above. Each set of the three conductive patterns 101 K, the three conductive patterns 101 L, and the three conductive patterns 101 M is also formed in a similar manner.

A via-conductor 102 G connects the vicinities of the end portions of the three conductive patterns 101 J close to the first side surface 25 to the ground conductor 101 D. A via-conductor 102 H connects the vicinities of the end portions of the three conductive patterns 101 J close to the second side surface 26 to the conductive pattern 101 A. The via-conductor 102 H is also connected to the conductive pattern 101 G somewhere halfway. The via-conductor 102 G, the conductive patterns 101 J, and the via-conductor 102 H form the inductor L 1 . That is, the inductor L 1 extends upward from the end portion connected to the ground conductor 101 D (this end portion will be called the ground-side end portion), then extends along the short side of the dielectric layers 100 from the first side surface 25 toward the second side surface 26 , and then extends toward the bottom surface 27 . The end portion (signal-side end portion) of the inductor L 1 opposite to the ground-side end portion is connected to the conductive pattern 101 A.

The inductor L 2 is constituted by a via-conductor 102 I, the conductive patterns 101 K, and a via-conductor 102 J. The via-conductor 102 I extends upward from the ground conductor 101 D. The conductive patterns 101 K extend from the via-conductor 102 I along the short side of the dielectric layers 100 from the first side surface 25 to the second side surface 26 . The via-conductor 102 J extends from the conductive patterns 101 K toward the bottom surface 27 . The inductor L 4 is constituted by a via-conductor 102 K, the conductive patterns 101 L, and a via-conductor 102 L. The via-conductor 102 K extends upward from the ground conductor 101 D. The conductive patterns 101 L extend from the via-conductor 102 K along the short side of the dielectric layers 100 from the first side surface 25 to the second side surface 26 . The via-conductor 102 L extends from the conductive patterns 101 L toward the bottom surface 27 . The inductor L 5 is constituted by a via-conductor 102 M, the conductive patterns 101 M, and a via-conductor 102 N. The via-conductor 102 M extends upward from the ground conductor 101 D. The conductive patterns 101 M extend from the via-conductor 102 M along the short side of the dielectric layers 100 from the first side surface 25 to the second side surface 26 . The via-conductor 102 N extends from the conductive patterns 101 M toward the bottom surface 27 .

The inductors L 1 , L 2 , L 4 , and L 5 are disposed in a direction parallel with both the first side surface 25 and the bottom surface 27 , that is, along the long sides of the dielectric layers 100 . The direction in which the inductors L 1 , L 2 , L 4 , and L 5 are arranged will be called the arrangement direction of the inductors L 1 through L 5 . The via-conductors 102 G, 102 I, 102 K, and 102 M of the inductors L 1 , L 2 , L 4 , and L 5 connected to the ground conductor 101 D are disposed closer to the first side surface 25 than the other via-conductors 102 H, 102 J, 102 L, and 102 N.

In one example, the dimension Lx ( FIG. 3A ) along the long sides of the filter device 20 is about 1.6 mm, and the dimension Ly ( FIG. 3A ) along the short sides of the filter device 20 is about 0.8 mm. The gap Gy 1 ( FIG. 3A ) between the first side surface 25 and the ground conductor 101 D is about 0.075 mm. The gap Gy 2 ( FIG. 3A ) between the second side surface 26 and the ground conductor 101 D is also about 0.075 mm.

FIG. 4 is a block diagram of a filter module 50 according to the first embodiment. The filter module 50 includes two antenna terminals 41 and 42 and two output terminals 45 and 46 . The two antenna terminals 41 and 42 are respectively connected to two filter devices 20 and 30 via a radio-frequency switch 35 . The filter device 20 is a band pass filter that allows signals of the 5G Band n77 frequency band to pass therethrough. The filter device 30 is a band pass filter that allows signals of the 5G Band n79 frequency band to pass therethrough.

Signals passing through the filter devices 20 and 30 are respectively inputted into low-noise amplifiers 37 and 38 . The signals amplified in the low-noise amplifiers 37 and 38 are respectively inputted into the output terminals 45 and 46 via a radio-frequency switch 36 and are outputted therefrom to the outside.

FIGS. 5A and 5B are respectively a perspective view and a sectional view of the filter module 50 . The filter devices 20 and 30 are surface-mounted on the mounting surface of a module substrate 51 . A region 54 where the low-noise amplifiers 37 and 38 ( FIG. 4 ) and other elements are mounted is reserved on the mounting surface of the module substrate 51 .

The module substrate 51 includes a ground plane 52 therein. The ground terminal TG of the filter device 20 is electrically connected to the ground plane 52 . Likewise, the ground terminal of the filter device 30 is also electrically connected to the ground plane 52 . A shield member 55 shields the filter devices 20 and 30 , the low-noise amplifiers 37 and 38 ( FIG. 4 ), and other elements mounted on the module substrate 51 . The shield member 55 is electrically connected to the ground plane 52 . The shield member 55 and the ground plane 52 electromagnetically shield the circuit components mounted on the module substrate 51 .

The shield member 55 is formed of a conductive film, such as a metal film, which covers the surface of a resin that seals multiple circuit components mounted on the module substrate 51 . The shield member 55 includes four side wall portions raised upward from the mounting surface of the module substrate 51 and a flat portion parallel with the module substrate 51 . The filter device 20 is mounted on the module substrate 51 such that the first side surface 25 faces one side wall portion of the shield member 55 . The side wall portion facing the first side surface 25 of the filter device 20 will be called a first side wall portion 55 A. The side wall portion located opposite to the first side wall portion 55 A will be called a second side wall portion 55 B. The second side wall portion 55 B faces the second side surface 26 of the filter device 20 with the filter device 30 and other circuit components interposed therebetween.

A gap G 1 between the first side surface 25 of the filter device 20 and the first side wall portion 55 A of the shield member 55 is smaller than a gap G 2 between the second side surface 26 and the second side wall portion 55 B.

In one example, the height H 1 from the bottom surface of the module substrate 51 to the bottom surface of the filter device 20 is about 0.12 mm. The height H 2 of the filter device 20 is about 0.325 mm. The height H 3 from the top surface of the filter device 20 to the flat portion of the shield member 55 is about 0.305 mm.

Before describing advantages of the first embodiment, the issue to be handled when the filter device 20 is mounted on the module substrate 51 and is electromagnetically shielded by the shield member 55 will be discussed below with reference to FIG. 6 .

FIG. 6 is a graph illustrating the results of measuring the transmission coefficient S 21 of a circuit constituted by the filter device 20 and the low-noise amplifier 37 ( FIG. 4 ). The horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. The broken line in FIG. 6 indicates the transmission coefficient S 21 of the circuit without the shield member 55 . The solid line in FIG. 6 indicates the transmission coefficient S 21 of the circuit with the shield member 55 .

In both the cases indicated by the broken line and the solid line, the transmission coefficient S 21 becomes high in the frequency band of about 3.2 to 4.2 GHz, which is the frequency band of 5G Band n77. In the frequency band of about 5 to 7 GHz, which is higher than the pass band, two attenuation poles where the transmission coefficient S 21 exhibits the local minimum values are observed. It is found that, when the shield member 55 is attached to the circuit, the positions of these two attenuation poles of the transmission coefficient S 21 are shifted. The example in FIG. 6 shows that the distance between the two attenuation poles is increased and the transmission coefficient S 21 in the frequency band between the two attenuation poles is higher compared with that in the circuit without the shield member 55 . That is, the attenuation characteristics in the attenuation band outside the pass band are degraded for the circuit with the shield member 55 .

The reason why the attenuation poles of the transmission coefficient S 21 are shifted when the shield member 55 is attached may be explained as follows. When the shield member 55 is attached, multiple inductors within the filter device 20 are coupled with each other due to mutual induction via the shield member 55 . This changes the circuit constants of the resonance circuits. The shifting amount of the attenuation poles is different depending on the positional relationship between the inductors and the shield member 55 .

Simulations are conducted with an electromagnetic simulator to observe how the attenuation poles of the transmission coefficient S 21 are changed when the positional relationship between the filter device 20 and the shield member 55 is varied. The simulation results will be discussed below with reference to FIGS. 7A through 8B .

FIG. 7A is a schematic diagram illustrating the positional relationship between the filter device 20 and the shield member 55 . In the example in FIG. 7A , the gap G 2 between the second side surface 26 of the filter device 20 and the second side wall portion 55 B of the shield member 55 is smaller than the gap G 1 between the first side surface 25 and the first side wall portion 55 A. This configuration is realized as a result of rotating the filter device 20 shown in FIGS. 5A and 5B by 180 degrees in a plan view. In this configuration, the side wall portion closer to the filter device 20 is the second side wall portion 55 B, while the side wall portion closer to the filter device 30 is the first side wall portion 55 A.

FIG. 7B is a graph illustrating the simulation results of the transmission coefficient S 21 of the circuit constituted by the filter device 20 and the low-noise amplifier 37 ( FIG. 4 ). The horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. The broken line in FIG. 7B indicates the transmission coefficient S 21 of the circuit without the shield member 55 . The solid line in FIG. 7B indicates the transmission coefficient S 21 of the circuit with the shield member 55 when the filter device 20 and the shield member 55 are arranged to have a gap G 2 of about 0.8 mm.

In the circuit without the shield member 55 , one attenuation pole is observed. In the circuit with the shield member 55 , two attenuation poles are observed. One of the two attenuation poles is located on the higher frequency side than the attenuation pole for the circuit without the shield member 55 , while the other attenuation pole is located on the lower frequency side than the attenuation pole for the circuit without the shield member 55 .

FIG. 8A is a schematic diagram illustrating the positional relationship between the filter device 20 and the shield member 55 . In the example in FIG. 8A , the gap G 1 between the first side surface 25 of the filter device 20 and the first side wall portion 55 A of the shield member 55 is smaller than the gap G 2 between the second side surface 26 and the second side wall portion 55 B.

FIG. 8B is a graph illustrating the simulation results of the transmission coefficient S 21 of the circuit constituted by the filter device 20 and the low-noise amplifier 37 ( FIG. 4 ). The horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. The broken line in FIG. 8B indicates the transmission coefficient S 21 of the circuit without the shield member 55 . The thin solid line in FIG. 8B indicates the transmission coefficient S 21 of the circuit with the shield member 55 when the filter device 20 and the shield member 55 are arranged to have a gap G 1 of about 0.6 mm. The thick solid line in FIG. 8B indicates the transmission coefficient S 21 of the circuit with the shield member 55 when the filter device 20 and the shield member 55 are arranged to have a gap G 1 of about 0.8 mm.

In the circuit without the shield member 55 , two attenuation poles are observed. In the circuit with the shield member 55 having a gap G 1 of about 0.6 mm, the attenuation poles have disappeared and the transmission coefficient S 21 in the frequency band of about 6 to 7 GHz is increased. In the circuit with the shield member 55 having a gap G 1 of about 0.8 mm, only one attenuation pole is observed, and the transmission coefficient S 21 in the frequency band of about 6 GHz drops.

The simulation results shown in FIG. 7A through 8B show that the transmission coefficient S 21 in the frequency band of about 6 to 7 GHz considerably varies depending on the positional relationship between the shield member 55 and the filter device 20 .

An explanation will be given, with reference to FIGS. 9A through 10B , of the reason why the transmission coefficient S 21 varies depending on the positional relationship between the shield member 55 and the filter device 20 .

FIG. 9A is an equivalent circuit diagram of the filter device 20 . FIG. 9B is a graph illustrating the calculation results of the transmission coefficient S 21 in the path from the first signal terminal T 1 to the second signal terminal T 2 in the equivalent circuit diagram in FIG. 9A . It is assumed that mutual inductance M 12 between the inductors L 1 and L 2 and mutual inductance M 45 between the inductors L 4 and L 5 will be changed depending on the positional relationship between the shield member 55 and the filter device 20 . The transmission coefficient S 21 when the mutual inductance M 12 and the mutual inductance M 45 are relatively small and that when the mutual inductance M 12 and the mutual inductance M 45 are relatively large are calculated.

In FIG. 9B , the horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. The solid line in FIG. 9B indicates the calculation result when the mutual inductance M 12 and the mutual inductance M 45 are relatively small. The broken line in FIG. 9B indicates the calculation result when the mutual inductance M 12 and the mutual inductance M 45 are relatively large.

FIG. 9B shows that, when the mutual inductance M 12 and the mutual inductance M 45 are large, the distance between the two attenuation poles is increased. This is similar to the case in which the attenuation poles of the transmission coefficient S 21 are shifted when the shield member 55 is attached to the circuit, as shown in FIG. 7B . It may thus be assumed that the shield member 55 disposed near the second side surface 26 of the filter device 20 functions to increase the mutual inductance M 12 and the mutual inductance M 45 .

FIG. 10A is an equivalent circuit diagram of the filter device 20 . FIG. 10B is a graph illustrating the calculation results of the transmission coefficient S 21 in the path from the first signal terminal T 1 to the second signal terminal T 2 in the equivalent circuit diagram in FIG. 10A . It is assumed that mutual inductance M 14 between the inductors L 1 and L 4 and mutual inductance M 25 between the inductors L 2 and L 5 will be changed depending on the positional relationship between the filter device 20 and the shield member 55 . The transmission coefficient S 21 when the mutual inductance M 14 and the mutual inductance M 25 are relatively small and that when the mutual inductance M 14 and the mutual inductance M 25 are relatively large are calculated.

In FIG. 10B , the horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. The solid line in FIG. 10B indicates the calculation result when the mutual inductance M 14 and the mutual inductance M 25 are relatively small. The broken line in FIG. 10B indicates the calculation result when the mutual inductance M 14 and the mutual inductance M 25 are relatively large.

FIG. 10B shows that, when the mutual inductance M 14 and the mutual inductance M 25 are large, the two attenuation poles have disappeared. The disappearance of the attenuation poles in this configuration is similar to that of the transmission coefficient S 21 for the circuit with the shield member 55 having a gap G 1 of about 0.6 mm indicated by the thin solid line in FIG. 8B . It may thus be assumed that the shield member 55 disposed near the first side surface 25 of the filter device 20 functions to increase the mutual inductance M 14 and the mutual inductance M 25 .

Simulations are conducted with an electromagnetic simulator to observe the transmission coefficient S 21 of the filter device 20 when the gap G 1 ( FIG. 5B ) is varied. The simulation results will be discussed below with reference to FIGS. 11A and 11B . In the simulations, the gap G 1 is set to be smaller than the gap G 2 ( FIG. 5B ), and the gap G 1 is varied, while the gap G 2 is fixed. The pass band of the filter device 20 is about 3.3 to 4.2 GHz, which corresponds to G5 Band n77.

FIGS. 11A and 11B are graphs illustrating the simulation results of the transmission coefficient S 21 in the path from the first signal terminal T 1 to the second signal terminal T 2 ( FIGS. 1 and 2 ) of the filter device 20 . The horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. FIG. 11B is an enlarged graph illustrating the frequency range of about 5.5 to 7.5 GHz of the graph in FIG. 11A .

The dimensions of the gap G 1 are indicated near the individual lines in FIG. 11B . The two attenuation poles are observed in each line in FIG. 11B . When the shield member 55 is arranged such that the gap G 1 is smaller than the gap G 2 , the attenuation pole of the higher frequency side is shifted to the lower frequency range from about 7 GHz at which the attenuation pole is located when the shield member 55 is not disposed. The attenuation pole of the lower frequency side is slightly shifted to the higher frequency range but does not exceed 6 GHz. The transmission coefficient S 21 is decreased in the frequency range of about 6 to 6.5 GHz compared with that without the shield member 55 . In particular, when the gap G 1 is about 0.1 to 0.3 mm, the transmission coefficient S 21 is reduced to about −60 dB or lower in the frequency range of about 6 to 6.5 GHz.

Advantages of the first embodiment will be described below.

In the first embodiment, as a result of arranging the filter device 20 and the shield member 55 such that the gap G 1 is smaller than the gap G 2 ( FIG. 5B ), the attenuation characteristics in the frequency band of about 6 to 6.5 GHz are considerably improved, in other words, the attenuation is considerably increased, compared with when the shield member 55 is not provided. Especially in the frequency band of 6 to 6.5 GHz, harmonic waves of a lower frequency band are likely to be superposed on a signal in the frequency band of 6 to 6.5 GHz as noise. The filter module 50 of the first embodiment can effectively be used particularly in the environments where high noise is likely to occur in the frequency band of about 6 to 6.5 GHz. With the configuration of the first embodiment, it is possible to obtain a high-performance filter device 20 having a pass band of the 5G Band n77 frequency band and having an attenuation band in the range of about 6 to 6.5 GHz.

To make the gap G 1 to be narrower than the gap G 2 , it is preferable that no circuit components be mounted between the first side surface 25 of the filter device 20 and the opposing first side wall portion 55 A of the shield member 55 . The circuit components, such as the filter device 30 ( FIG. 5A ) and the low-noise amplifiers 37 and 38 ( FIG. 4 ), are disposed between the second side surface 26 and the opposing second side wall portion 55 B of the shield member 55 .

The gap Gy 1 ( FIG. 3A ) between the first side surface 25 of the filter device 20 and the ground conductor 101 D is about 0.075 mm. When the gap G 1 is about 0.1 to 0.3 mm, the gap between the first side wall portion 55 A and the ground conductor 101 D is about 0.175 to 0.375 mm. From the viewpoint of the ground conductor 101 D of the filter device 20 , it is preferable that the gap between the first side wall portion 55 A and the ground conductor 101 D be about 0.175 to 0.375 mm.

A modified example of the first embodiment will be described below.

The filter device 20 of the filter module 50 according to the first embodiment includes the four parallel resonance circuits 21 , 22 , 23 , and 24 . However, the filter device 20 may include any multiple number of parallel resonance circuits.

Second Embodiment

A filter module 50 according to a second embodiment will be described below with reference to FIGS. 12, 13A, and 13B . The same configuration as that of the filter module 50 of the first embodiment will not be explained.

FIG. 12 is a sectional view of the filter module 50 according to the second embodiment. In the first embodiment, the gap G 1 is set to be smaller than the gap G 2 , as shown in FIG. 5B . In the second embodiment, the gap G 2 is set to be smaller than the gap G 1 . In other words, the second side surface 26 of the filter device 20 is disposed close to the second side wall portion 55 B of the shield member 55 , and no circuit components are mounted therebetween.

Simulations are conducted with an electromagnetic simulator to observe the transmission coefficient S 21 in the path from the first signal terminal T 1 to the second signal terminal T 2 ( FIGS. 1 and 2 ) of the filter device 20 . FIGS. 13A and 13B are graphs illustrating the simulation results. The horizontal axis indicates the frequency expressed by GHz, and the vertical axis indicates the transmission coefficient S 21 expressed by dB. FIG. 13B is an enlarged graph illustrating the frequency range of about 5.5 to 7.5 GHz of the graph in FIG. 13A .

When the gap G 2 is smaller than about 0.5 mm, the attenuation pole of the lower frequency side is shifted to the lower frequency range and the attenuation pole of the higher frequency side is shifted to the higher frequency range from those of the transmission coefficient S 21 when the shield member 55 is not provided. The reason for this may be that the mutual inductance M 12 between the inductors L 1 and L 2 and the mutual inductance M 45 between the inductors L 4 and L 5 are increased by the provision of the shield member 55 , as shown in FIGS. 9A and 9B .

When the gap G 2 is about 0.5 mm or larger, the provision of the shield member 55 does not significantly change the transmission coefficient S 21 from that when the shield member 55 is not provided. That is, the transmission coefficient S 21 is not considerably influenced by the shield member 55 . It is thus preferable to set the gap G 2 between the second side surface 26 of the filter device 20 and the second side wall portion 55 B of the shield member 55 to be about 0.5 mm or larger.

Advantages of the second embodiment will be described below.

In the second embodiment, as a result of arranging the filter device 20 and the shield member 55 such that the gap G 2 is smaller than the gap G 1 and that the gap G 2 is about 0.5 mm or larger, the transmission coefficient S 21 of the filter device 20 is less influenced by the provision of the shield member 55 . The gap Gy 1 ( FIG. 3A ) between the first side surface 25 of the filter device 20 and the ground conductor 101 D is about 0.075 mm. When the gap G 2 is about 0.5 mm or larger, the gap between the second side wall portion 55 B and the ground conductor 101 D is about 0.575 mm or larger. From the viewpoint of the ground conductor 101 D of the filter device 20 , it is preferable that the gap between the second side wall portion 55 B and the ground conductor 101 D be about 0.575 mm or larger.

The disclosure is not limited to the above-described embodiments. The configurations described in the different embodiments may partially be replaced by or combined with each other. Similar advantages obtained by similar configurations in plural embodiments are not repeated in the individual embodiments.

While preferred embodiments of the disclosure have been described above, it is to be understood that variations, improvements, combinations, and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the disclosure. The scope of the disclosure, therefore, is to be determined solely by the following claims.