Converter and power conversion device manufactured using the same

The present invention relates to a converter and a power conversion device manufactured using the converter, and particularly to a converter configured to convert an alternating-current (AC) voltage into first to third direct-current (DC) voltages, and a power conversion device manufactured using the converter.

Japanese Patent Laying-Open No. 2011-78296 (PTD 1) discloses a converter including four transistors and four diodes, and configured to convert an AC voltage into a high voltage, a low voltage and an intermediate voltage. According to this converter, among four diodes, each of two diodes performing a reverse recovery operation is formed of a wide band gap semiconductor, thereby reducing recovery loss. Also, each of two diodes not performing a reverse recovery operation is formed of a semiconductor other than such a wide band gap semiconductor, thereby reducing cost.

Patent Document

PTD 1: Japanese Patent Laying-Open No. 2011-78296

Technical Problem

In the conventional converter, however, four transistors have been formed of the same type of semiconductor, still leading to higher loss and higher cost.

Therefore, a main object of the present invention is to provide a converter with reduced loss and cost, and a power conversion device manufactured using the converter.

Solution to Problem

A converter according to the present invention is configured to convert an AC voltage applied to an input terminal into a first DC voltage, a second DC voltage and a third DC voltage, and output the first DC voltage, the second DC voltage and the third DC voltage through a first output terminal, a second output terminal and a third output terminal, respectively. The converter includes: a first transistor having a first electrode and a second electrode that are connected to the first output terminal and the input terminal, respectively; a second transistor having a first electrode and a second electrode that are connected to the input terminal and the second output terminal, respectively; a first diode and a second diode connected in anti-parallel to the first transistor and the second transistor, respectively; and a first bidirectional switch connected between the input terminal and the third output terminal. The first DC voltage is higher than the second DC voltage, and the third DC voltage is an intermediate voltage between the first DC voltage and the second DC voltage. The first bidirectional switch includes a third transistor, a fourth transistor, a third diode, and a fourth diode. Each of the first diode, the second diode, the third transistor, and the fourth transistor is formed of a wide band gap semiconductor. Each of the third diode, the fourth diode, the first transistor, and the second transistor is formed of a semiconductor other than the wide band gap semiconductor.

Advantageous Effects of Invention

In the converter according to the present invention, the first and second diodes performing a reverse recovery operation, and the third and fourth transistors switching a current each are formed of a wide band gap semiconductor, so that the switching loss and the recovery loss can be reduced. Also, the third and fourth diodes not performing a reverse recovery operation, and the first and second transistors not switching a current each are formed of a semiconductor other than a wide band gap semiconductor, so that the cost can be reduced.

FIG. 1 is a circuit block diagram showing the configuration of a converter according to the first embodiment of the present invention.

FIG. 2 is a time chart showing waveforms of four PWM signals that control four transistors, respectively, shown in FIG. 1 .

FIG. 3 is a circuit diagram for illustrating a current flowing through a converter shown in FIG. 1 .

FIG. 4 is a time chart showing the current flowing through the converter shown in FIG. 1 .

FIG. 5 is a time chart for illustrating switching loss in each of two types of transistors shown in FIG. 1 .

FIG. 6 is a block diagram showing the configuration of a semiconductor module included in the converter shown in FIG. 1 .

FIG. 7 is a circuit block diagram showing the configuration of an uninterruptible power supply device including the converter shown in FIG. 1 .

FIG. 8 is a circuit block diagram showing a modification of the first embodiment.

FIG. 9 is a circuit block diagram showing another modification of the first embodiment.

FIG. 10 is a circuit block diagram showing still another modification of the first embodiment.

FIG. 11 is a circuit block diagram showing the configuration of a converter according to the second embodiment of the present invention.

FIG. 12 is a circuit block diagram showing the configuration of a converter according to the third embodiment of the present invention.

FIG. 13 is a circuit block diagram showing the configuration of an inverter included in an uninterruptible power supply device according to the fourth embodiment of the present invention.

FIG. 14 is a time chart showing waveforms of four PWM signals controlling four transistors, respectively, shown in FIG. 13 .

FIG. 15 is a circuit diagram for illustrating a current flowing through the inverter shown in FIG. 13 .

FIG. 16 is a time chart showing the current flowing through the inverter shown in FIG. 13 .

FIG. 17 is a block diagram showing the configuration of a semiconductor module included in the inverter shown in FIG. 13 .

FIG. 18 is a circuit block diagram showing a modification of the fourth embodiment.

FIG. 19 is a circuit block diagram showing another modification of the fourth embodiment.

FIG. 20 is a circuit block diagram showing still another modification of the fourth embodiment.

FIG. 21 is a circuit block diagram showing the configuration of an inverter according to the fifth embodiment of the present invention.

FIG. 22 is a circuit block diagram showing the configuration of an inverter according to the sixth embodiment of the present invention.

First Embodiment

FIG. 1 is a circuit block diagram showing the configuration of a converter according to the first embodiment of the present invention. In FIG. 1 , this converter includes an input terminal T 0 , output terminals T 1 to T 3 , transistors Q 1 to Q 4 , and diodes D 1 to D 4 .

Input terminal T 0 receives an AC voltage VAC of a commercial frequency, for example, from a commercial AC power supply 10 . A battery B 1 has a positive electrode and a negative electrode that are connected to output terminals T 1 and T 3 , respectively. A battery B 2 has a positive electrode and a negative electrode that are connected to output terminals T 3 and T 2 , respectively. Each of batteries B 1 and B 2 stores DC power. Batteries B 1 and B 2 are charged with the same DC voltage. Voltages on output terminals T 1 , T 2 and T 3 are defined as DC voltages V 1 , V 2 , and V 3 , respectively, which leads to conditions of V 1 >V 3 >V 2 and V 3 =(V 1 +V 2 )/2. This converter serves to convert AC voltage VAC, which has been applied to input terminal T 0 , into DC voltages V 1 to V 3 , and outputs the converted DC voltages V 1 to V 3 through output terminals T 1 to T 3 . In addition, when output terminal T 3 is grounded, DC voltages V 1 , V 2 and V 3 are to be set at a positive voltage, a negative voltage and 0V, respectively.

Each of transistors Q 1 and Q 2 is an insulated gate bipolor transistor (IGBT) formed using Si (silicon) that is a semiconductor other than a wide band gap semiconductor. The rated current in each of transistors Q 1 and Q 2 is 300 A, for example.

Each of transistors Q 3 and Q 4 is an N-channel MOS transistor formed using SiC (silicon carbide) that is a wide band gap semiconductor. The rated current in each of transistors Q 3 and Q 4 is 500 A, for example.

Each of diodes D 1 and D 2 is a Schottky barrier diode formed using SiC (silicon carbide) that is a wide band gap semiconductor. The rated current in each of diodes D 1 and D 2 is, for example, 600 A that is greater than the rated current in each of transistors Q 1 to Q 4 and diodes D 3 and D 4 .

Each of diodes D 3 and D 4 is formed using Si (silicon) that is a semiconductor other than a wide band gap semiconductor. The rated current in each of diodes D 3 and D 4 is 450 A, for example.

In this way, transistors Q 1 and Q 2 are different in specification from transistors Q 3 and Q 4 , and diodes D 1 and D 2 are different in specification from diodes D 3 and D 4 , the reason for which will be described later.

Transistor Q 1 has: a collector (the first electrode) connected to output terminal T 1 (the first output terminal); and an emitter (the second electrode) connected to input terminal T 0 . Diode D 1 has: an anode connected to input terminal T 0 ; and a cathode connected to output terminal T 1 . In other words, diode D 1 is connected in anti-parallel to transistor Q 1 .

Transistor Q 2 has: a collector (the first electrode) connected to input terminal T 0 ; and an emitter (the second electrode) connected to output terminal T 2 (the second output terminal). Diode D 2 has: an anode connected to output terminal T 2 ; and a cathode connected to input terminal T 0 . In other words, diode D 2 is connected in anti-parallel to transistor Q 2 .

Transistors Q 3 and Q 4 have: drains (the second electrodes) that are connected to each other; and sources (the first electrodes) that are connected to input terminal T 0 and output terminal T 3 (the third output terminal), respectively. Diodes D 3 and D 4 have: cathodes that are connected to drains of transistors Q 3 and Q 4 , respectively; and anodes that are connected to input terminal T 0 and output terminal T 3 , respectively. In other words, diodes D 3 and D 4 are connected in anti-parallel to transistors Q 3 and Q 4 , respectively. Transistors Q 3 , Q 4 and diodes D 3 , D 4 form the first bidirectional switch.

Then, the operation of this converter will be hereinafter described. Transistors Q 1 to Q 4 have gates to which PWM signals ϕ 1 to ϕ 4 , respectively, are supplied. FIGS. 2( a ) to 2( e ) each show a method of generating PWM signals ϕ 1 to ϕ 4 , and waveforms. Specifically, FIG. 2( a ) shows waveforms of a sinusoidal wave command value signal CM, a positive-side triangular wave carrier signal CA 1 and a negative-side triangular wave carrier signal CA 2 . FIGS. 2( b ), 2( c ), 2( d ), and 2( e ) show the waveforms of PWM signals ϕ 1 , ϕ 3 , ϕ 4 , and ϕ 2 , respectively.

In FIGS. 2( a ) to 2( e ) , the frequency of sinusoidal wave command value signal CM is a commercial frequency, for example. The phase of sinusoidal wave command value signal CM is the same as the phase of AC voltage VAC of the commercial frequency, for example. Carrier signals CA 1 and CA 2 show the same cycle and the same phase. The cycles of carrier signals CA 1 and CA 2 are sufficiently smaller than the cycle of sinusoidal wave command value signal CM.

The high-low levels of sinusoidal wave command value signal CM and positive-side triangular wave carrier signal CA 1 are compared with each other. When the level of sinusoidal wave command value signal CM is higher than the level of positive-side triangular wave carrier signal CAL PWM signals ϕ 1 and ϕ 4 are set at an “H” level and an “L” level, respectively. When the level of sinusoidal wave command value signal CM is lower than the level of positive-side triangular wave carrier signal CAL PWM signals ϕ 1 and ϕ 4 are set at an “L” level and an “H” level, respectively.

Accordingly, in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals 41 and 44 are alternately set at an “H” level in synchronization with carrier signal CA 1 , and thus, transistors Q 1 and Q 4 are alternately turned on. Furthermore, in a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 1 and ϕ 4 are fixed at an “L” level and an “H” level, respectively, and thus, transistor Q 1 is fixed in the OFF state while transistor Q 4 is fixed in the ON state.

The high-low levels of sinusoidal wave command value signal CM and negative-side triangular wave carrier signal CA 2 are compared with each other. When the level of sinusoidal wave command value signal CM is higher than the level of positive-side triangular wave carrier signal CA 2 , PWM signals ϕ 2 and ϕ 3 are set at an “L” level and an “H” level, respectively. When the level of sinusoidal wave command value signal CM is lower than the level of positive-side triangular wave carrier signal CA 2 , PWM signals ϕ 2 and ϕ 3 are set at an “H” level and an “L” level, respectively.

Therefore, in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals ϕ 2 and ϕ 3 are fixed at an “L” level and an “H” level, respectively, and thus, transistor Q 2 is fixed in the OFF state while transistor Q 3 is fixed in the ON state. Furthermore, in a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 2 and ϕ 3 are alternately set at an “H” level in synchronization with carrier signal CA 2 , and thus, transistors Q 2 and Q 3 are alternately turned on.

The ratio between the time in which the PWM signal is set at an “H” level within one cycle and the time of one cycle of the PWM signal is referred to as a duty ratio. In a time period during which the level of sinusoidal wave command value signal CM is positive, the duty ratio of PWM signal ϕ 1 is maximized in the vicinity of a positive peak (90 degrees) of sinusoidal wave command value signal CM. Also, the duty ratio of PWM signal ϕ 1 decreases with increasing distance from the peak, and reaches 0 in the vicinity of 0 degree and 180 degrees. The duty ratio of PWM signal ϕ 1 is fixed at 0 in a time period during which sinusoidal wave command value signal CM is negative. PWM signal ϕ 4 is a complementary signal of PWM signal ϕ 1 .

The duty ratio of PWM signal ϕ 2 is fixed at 0 in a time period during which the level of sinusoidal wave command value signal CM is positive. The duty ratio of PWM signal ϕ 2 is maximized in the vicinity of a negative peak (270 degrees) of sinusoidal wave command value signal CM. Also, the duty ratio of PWM signal ϕ 2 decreases with increasing distance from the peak, and reaches 0 in the vicinity of 180 degrees and 360 degrees. PWM signal ϕ 3 is a complementary signal of PWM signal ϕ 2 .

Then, the current flowing through each of transistors Q 1 to Q 4 and diodes D 1 to D 4 during the operation of the converter will be hereinafter described. As shown in FIG. 3 , the current flowing from output terminal T 1 into input terminal T 0 is defined as I 1 ; the current flowing from input terminal T 0 into output terminal T 2 is defined as I 2 ; the current flowing from output terminal T 3 into input terminal T 0 is defined as I 3 ; and the current flowing from input terminal T 0 into output terminal T 3 is defined as I 4 .

FIGS. 4( a ) to 4( i ) are time charts each showing the operation of the converter. Specifically, FIG. 4( a ) shows waveforms of sinusoidal wave command value signal CM, positive-side triangular wave carrier signal CA 1 and negative-side triangular wave carrier signal CA 2 . FIGS. 4( b ), 4( d ), 4( f ), and 4( h ) show waveforms of PWM signals ϕ 1 , ϕ 3 , ϕ 4 , and ϕ 2 , respectively, and FIGS. 4( c ), 4( e ), 4( g ), and 4( i ) show waveforms of currents I 1 , I 3 , I 4 , and I 2 , respectively. Among currents I 1 to I 4 , a positive current shows the current flowing through transistor Q while a negative current shows the current flowing through diode D. The figures also show the case where the power factor is 1.0, in which the phase of sinusoidal wave command value signal CM matches with the phase of AC voltage VAC.

In FIGS. 4( a ) to 4( i ) , in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals ϕ 3 and ϕ 2 are fixed at an “H” level and an “L” level, respectively, and PWM signals ϕ 1 and ϕ 4 are alternately set at an “H” level. Accordingly, transistors Q 3 and Q 2 are fixed in an ON state and an OFF state, respectively, and transistors Q 1 and Q 4 are alternately turned on. Thus, a current flows from input terminal T 0 alternately into output terminals T 1 and T 3 .

During this time period, when transistor Q 4 is turned off, current T 1 of the level corresponding to the level of AC voltage VAC flows from input terminal T 0 through diode D 1 into output terminal T 1 ; and when transistor Q 4 is turned on, current I 4 of the level complementing current I 1 flows on a passage extending from input terminal T 0 through transistors Q 3 and Q 4 to output terminal T 3 .

Since transistor Q 2 is fixed in the OFF state, a current does not flow through transistor Q 2 , so that switching loss does not occur in transistor Q 2 . Although transistor Q 1 is turned on/off, a current flows through diode D 1 and a current does not flow through transistor Q 1 , so that switching loss does not occur in transistor Q 1 . Since transistor Q 3 is fixed in the ON state, a current flows through transistor Q 3 , but switching loss does not occur in transistor Q 3 . Accordingly, during this time period, among diodes D 1 to D 4 and transistors Q 1 to Q 4 , the current flowing through diode D 1 shows the largest effective value while the greatest switching loss occurs in transistor Q 4 .

Each time transistor Q 4 is changed from the OFF state to the ON state, a reverse bias voltage is applied to diode D 1 , and this diode D 1 performs a reverse recovery operation. No current flows through diodes D 2 and D 4 during this time period.

In a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 4 and ϕ 1 are fixed at an “H” level and an “L” level, respectively, and PWM signals ϕ 2 and ϕ 3 are alternately set at an “H” level. Accordingly, transistors Q 4 and Q 1 are fixed in the ON state and the OFF state, respectively, and transistors Q 2 and Q 3 are alternately turned on, and thus, a current flows alternately from output terminals T 2 and T 3 into input terminal T 0 .

During this time period, when transistor Q 3 is turned off, current I 2 of the level corresponding to the level of AC voltage VAC flows from output terminal T 2 through diode D 2 into input terminal T 0 ; and when transistor Q 3 is turned on, current I 3 of the level complementing current I 2 flows through a passage extending from output terminal T 3 via transistors Q 4 and Q 3 to output terminal T 0 .

Since transistor Q 1 is fixed in the OFF state, a current does not flow through transistor Q 1 , and thus, switching loss does not occur in transistor Q 1 . Although transistor Q 2 is turned on/off, a current flows through diode D 2 while a current does not flow through transistor Q 2 , with the result that switching loss does not occur in transistor Q 2 . Since transistor Q 4 is fixed in the ON state, a current flows through transistor Q 4 , but switching loss does not occur in transistor Q 4 . Accordingly, during this time period, among diodes D 1 to D 4 and transistors Q 1 to Q 4 , the current flowing through diode D 2 shows the largest effective value, and the greatest switching loss occurs in transistor Q 3 .

Furthermore, each time transistor Q 3 is changed from the OFF state to the ON state, a reverse bias voltage is applied to diode D 2 , and this diode D 2 performs a reverse recovery operation. Also, no current flows through other diodes D 1 and D 3 during this time period.

In summary, a large current flows through diodes D 1 and D 2 , and these diodes D 1 and D 2 each perform a reverse recovery operation. A current flowing through diodes D 3 and D 4 is smaller than the current flowing through diodes D 1 and D 2 , and thus, these diodes D 3 and D 4 each do not perform a reverse recovery operation. No current flows through transistors Q 1 and Q 2 , so that switching loss does not occur in transistors Q 1 and Q 2 . A current flows through transistors Q 3 and Q 4 , so that switching loss occurs in transistors Q 3 and Q 4 .

Thus, as described above, as transistors Q 3 and Q 4 , an N-channel MOS transistor is employed that is formed of SiC as a wide band gap semiconductor and that has a rated current of a large value (for example, 500 A), thereby reducing switching loss. Furthermore, as transistors Q 1 and Q 2 , an IGBT is employed that is formed of Si as a semiconductor other than a wide band gap semiconductor and that has a rated current of a small value (for example, 300 A), thereby reducing cost.

As diodes D 1 and D 2 , a Schottky barrier diode is employed that is formed of SiC as a wide band gap semiconductor and that has a rated current of a large value (for example, 600 A), thereby reducing recovery loss during the reverse recovery operation. As diodes D 3 and D 4 , a diode is employed that is formed of Si as a semiconductor other than a wide band gap semiconductor and that has a rated current of a small value (for example, 450 A), thereby reduction cost.

FIG. 5( a ) is a time chart showing the switching operation of an N-channel MOS transistor formed using Si (which will be referred to as a Si transistor). FIG. 5( b ) is a time chart showing the switching operation of an N-channel MOS transistor formed using SiC (which will be referred to as a SiC transistor).

In FIGS. 5( a ) and 5( b ) , in the initial state, a gate signal (not shown) is set at an “H” level to turn on the transistor, to cause a fixed current I to flow through the transistor, in which case a drain-source voltage Vds is set at 0V. When the gate signal is lowered from an “H” level to an “L” level at a certain time to turn off the transistor, current I decreases and voltage Vds increases.

As apparent from FIGS. 5( a ) and 5( b ) , a time period Ta in the Si transistor from when current I starts to fall until when current I reaches 0 A is longer than a time period Tb in the SiC transistor from when current I starts to fall until when current I reaches 0 A. In the Si transistor, current I decreases quickly until this current I reaches a certain value. From this certain value, however, it takes long for current I to reach 0 A. The current flowing in a time period during which this current reaches 0 A from such a certain value is referred to as a tail current.

On the other hand, in the SiC transistor, current I decreases immediately and some overshoot occurs. The switching loss in the transistor is represented by a product of current I and voltage Vds, which corresponds to an area of the diagonally shaded region in the figure. Accordingly, the switching loss in the SiC transistor is smaller than the switching loss in the Si transistor.

FIG. 6 is a diagram showing the external appearance of the converter shown in FIG. 1 . In FIG. 6 , the converter includes one semiconductor module M 1 . Semiconductor module M 1 is provided on its inside with transistors Q 1 to Q 4 and diodes D 1 to D 4 . Semiconductor module M 1 is provided on its outside with an input terminal T 0 and output terminals T 1 to T 3 . Furthermore, semiconductor module M 1 is provided on its outside with four signal terminals used for supplying PWM signals ϕ 1 to ϕ 4 to the gates of transistors Q 1 to Q 4 , respectively, but these four signal terminals are not shown for simplification of illustration of the figure.

FIG. 7 is a circuit block diagram showing the configuration of an uninterruptible power supply device including the converter shown in FIG. 1 . In FIG. 7 , the uninterruptible power supply device includes an input filter 1 , a converter 2 , a DC positive bus L 1 , a DC negative bus L 2 , a DC neutral point bus L 3 , capacitors C 1 , C 2 , an inverter 3 , an output filter 4 , and a controller 5 .

Input filter 1 serves as a low pass filter to allow the AC power of a commercial frequency from commercial AC power supply 10 to flow into input terminal T 0 of converter 2 , and to prevent the signal of a carrier frequency generated in converter 2 from passing through to commercial AC power supply 10 .

DC positive bus L 1 , DC negative bus L 2 and DC neutral point bus L 3 have: one ends connected to output terminals T 1 , T 2 and T 3 , respectively, of converter 2 ; and the other ends connected to three input terminals, respectively, of inverter 3 . Capacitor C 1 is connected between buses L 1 and L 3 while capacitor C 2 is connected between buses L 3 and L 2 . Buses L 1 and L 3 are connected to the positive electrode and the negative electrode, respectively, of battery B 1 while buses L 3 and L 2 are connected to the positive electrode and the negative electrode, respectively, of battery B 2 .

As shown in FIG. 1 , converter 2 includes an input terminal T 0 , output terminals T 1 to T 3 , transistors Q 1 to Q 4 , and diodes D 1 to D 4 , and is controlled by PWM signals ϕ 1 to ϕ 4 from controller 5 .

In the normal situation in which AC power is normally supplied from commercial AC power supply 10 , converter 2 converts the AC power supplied from commercial AC power supply 10 through input filter 1 into DC power, and supplies the converted DC power to each of batteries B 1 and B 2 and also to inverter 3 . Each of batteries B 1 and B 2 stores DC power.

In other words, converter 2 is controlled by PWM signals ϕ 1 to ϕ 4 supplied from controller 5 , to generate DC voltages V 1 to V 3 based on AC voltage VAC supplied from commercial AC power supply 10 through input filter 1 , and then, applies generated DC voltages V 1 , V 2 and V 3 to DC positive bus L 1 , DC negative bus L 2 and DC neutral point bus L 3 , respectively. In addition, when output terminal T 3 is grounded, DC voltages V 1 , V 2 and V 3 are set at a positive voltage, a negative voltage and 0V, respectively. DC voltages V 1 to V 3 are smoothed by capacitors C 1 and C 2 . DC voltages V 1 to V 3 are supplied to batteries B 1 , B 2 and inverter 3 . Converter 2 is stopped at the time of power interruption during which supply of the AC power from commercial AC power supply 10 is stopped.

In the normal situation in which AC power is normally supplied from commercial AC power supply 10 , inverter 3 converts the DC power generated in converter 2 into AC power. Also, at the time of power interruption during which supply of the AC power from commercial AC power supply 10 is stopped, inverter 3 converts the DC power on each of batteries B 1 and B 2 into AC power.

In other words, in the normal situation, inverter 3 generates a three-level AC voltage based on DC voltages V 1 to V 3 supplied from converter 2 through buses L 1 to L 3 . During power interruption, inverter 3 generates a three-level AC voltage based on DC voltages V 1 to V 3 supplied from batteries B 1 and B 2 through buses L 1 to L 3 .

Output filter 4 is connected between the output terminal of inverter 3 and a load 11 . Output filter 4 serves as a low pass filter to allow the AC power of a commercial frequency included in the AC power output from inverter 3 to flow into load 11 , and also to prevent the signal of a carrier frequency generated in inverter 3 from passing through to load 11 . In other words, output filter 4 converts the output voltage of inverter 3 into a sinusoidal wave of a commercial frequency, and supplies the converted sinusoidal wave to load 11 .

Controller 5 is configured to control converter 2 and inverter 3 by supplying a PWM signal while monitoring the AC voltage from commercial AC power supply 10 , the AC voltage output to load 11 , and DC voltages V 1 to V 3 , and the like.

Then, the operation of this uninterruptible power supply device will be hereinafter described. In the normal situation in which AC power is normally supplied from commercial AC power supply 10 , the AC power from commercial AC power supply 10 is supplied to converter 2 through input filter 1 , and converted into DC power by converter 2 . The DC power generated by converter 2 is stored in batteries B 1 and B 2 and also supplied to inverter 3 and converted by this inverter 3 into AC power of a commercial frequency. The AC power generated by inverter 3 is supplied to load 11 through output filter 4 , thereby operating load 11 .

At the time of power interruption during which supply of the AC power from commercial AC power supply 10 is stopped, the operation of converter 2 is stopped, and the DC power from each of batteries B 1 and B 2 is supplied to inverter 3 and converted by inverter 3 into AC power of a commercial frequency. The AC power generated in inverter 3 is supplied to load 11 through output filter 4 so as to continue the operation of load 11 .

Accordingly, even when power interruption occurs, the operation of load 11 is continued as long as DC power is stored in each of batteries B 1 and B 2 . When supply of the AC power from commercial AC power supply 10 is resumed, the operation of converter 2 is resumed. Then, the DC power generated in converter 2 is supplied to each of batteries B 1 , B 2 and inverter 3 , so that its original state is brought back.

As described above, in the present first embodiment, a Schottky barrier diode formed of a wide band gap semiconductor is used as diodes D 1 and D 2 performing a reverse recovery operation, and also, a diode formed of a semiconductor other than a wide band gap semiconductor is used as diodes D 3 and D 4 not performing a reverse recovery operation. Accordingly, the recovery loss and the cost can be reduced.

Furthermore, an N-channel MOS transistor formed of a wide band gap semiconductor is used as transistors Q 3 and Q 4 turning on/off a current. Also, an IGBT formed of a semiconductor other than a wide band gap semiconductor is used as transistors Q 1 and Q 2 not turning on/off a current. Accordingly, the switching loss and the cost can be reduced.

In addition, SiC is used as a wide band gap semiconductor in the present first embodiment, but not limited thereto, and any other semiconductors may be used as long as such semiconductors are a wide band gap semiconductor. For example, GaN (gallium nitride) may be used as a wide band gap semiconductor.

FIG. 8 is a block diagram showing a modification of the first embodiment, which is shown as compared with FIG. 6 . In FIG. 8 , in the present modification, a converter includes a substrate BP 1 , and two semiconductor modules M 2 and M 3 mounted on its surface. Semiconductor module M 2 is provided on its inside with transistors Q 1 , Q 2 and diodes D 1 , D 2 . Semiconductor module M 2 is provided on its outside with an input terminal T 0 and output terminals T 1 , T 2 . Furthermore, semiconductor module M 2 is provided on its outside with two signal terminals (not shown) used for supplying PWM signals ϕ 1 and ϕ 2 to the gates of transistors Q 1 and Q 2 .

Semiconductor module M 3 is provided on its inside with transistors Q 3 , Q 4 and diodes D 3 , D 4 . Semiconductor module M 3 is provided on its outside with an input terminal T 0 and an output terminal T 3 . Semiconductor module M 3 is provided on its outside with two signal terminals (not shown) used for supplying PWM signals ϕ 3 and ϕ 4 to the gates of transistors Q 3 and Q 4 . Input terminal T 0 of semiconductor module M 2 and input terminal T 0 of semiconductor module M 3 are connected to each other. The same effect as that in the first embodiment can be achieved also in the present modification.

FIG. 9 is a block diagram showing another modification of the first embodiment, which is shown as compared with FIG. 6 . In FIG. 9 , in the present modification, a converter includes a substrate BP 2 , and two semiconductor modules M 4 and M 5 mounted on its surface. Semiconductor module M 4 is provided on its inside with transistors Q 1 to Q 4 . Semiconductor module M 4 is provided on its outside with an input terminal T 0 , output terminals T 1 to T 3 , and an intermediate terminal T 4 . Intermediate terminal T 4 is connected to each of drains of transistors Q 3 and Q 4 . Semiconductor module M 4 is provided on its outside with four signal terminals (not shown) used for supplying PWM signals ϕ 1 to ϕ 4 to the gates of transistors Q 1 and Q 4 .

Semiconductor module M 5 is provided on its inside with diodes D 1 to D 4 . Semiconductor module M 5 is provided on its outside with an input terminal T 0 , output terminals T 1 to T 3 , and an intermediate terminal T 4 . Intermediate terminal T 4 is connected to each of cathodes of diodes D 3 and D 4 . Terminals T 0 to T 4 of semiconductor module M 4 are connected to terminals T 0 to T 4 , respectively, of semiconductor module M 5 . The same effect as that in the first embodiment can be achieved also in the present modification.

FIG. 10 is a block diagram showing still another modification of the first embodiment, which is shown as compared with FIG. 6 . In FIG. 10 , in the present modification, a converter includes a substrate BP 3 and eight semiconductor modules M 11 to M 18 mounted on its surface. Semiconductor modules M 11 to M 14 are provided on their insides with transistors Q 1 to Q 4 , respectively. Semiconductor modules M 15 to M 18 are provided on their insides with diodes D 1 to D 4 , respectively. Each of semiconductor modules M 11 and M 15 includes terminals T 0 and T 1 . Each of semiconductor modules M 12 and M 16 includes terminals T 0 and T 2 . Terminals T 1 of semiconductor modules M 11 and M 15 are connected to each other. Terminals T 2 of semiconductor modules M 12 and M 16 are connected to each other.

Each of semiconductor modules M 13 and M 17 includes terminals T 0 and T 4 . Each of semiconductor modules M 14 and M 18 includes terminals T 3 and T 4 . Terminals T 4 of semiconductor modules M 13 and M 14 are connected to drains of transistors Q 3 and Q 4 , respectively. Terminals T 4 of semiconductor modules M 17 and M 18 are connected to cathodes of diodes D 3 and D 4 , respectively. Terminals T 0 of semiconductor modules M 11 to M 13 and M 15 to M 17 are connected to each other. Terminals T 4 of semiconductor modules M 13 , M 14 , M 17 , and M 18 are connected to each other. Terminals T 3 of semiconductor modules M 14 and M 18 are connected to each other. Furthermore, semiconductor modules M 11 to M 14 are provided on their outsides with four signal terminals (not shown) used for supplying PWM signals ϕ 1 to ϕ 4 to the gates of transistors Q 1 to Q 4 , respectively. The same effect as that in the first embodiment can be achieved also in the present modification.

Second Embodiment

FIG. 11 is a circuit diagram showing the configuration of a converter according to the second embodiment of the present invention, which is shown as compared with FIG. 1 . Referring to FIG. 11 , this converter is different from the converter in FIG. 1 in that the parallel connection body formed of transistor Q 3 and diode D 3 and the parallel connection body formed of transistor Q 4 and diode D 4 are replaced with each other.

Transistors Q 3 and Q 4 have: sources that are connected to each other; and drains that are connected to output terminal T 3 and input terminal T 0 , respectively. Diodes D 3 and D 4 have: anodes that are connected to sources of transistors Q 3 and Q 4 , respectively; and cathodes that are connected to output terminal T 3 and input terminal T 0 , respectively. In other words, diodes D 3 and D 4 are connected in anti-parallel to transistors Q 3 and Q 4 , respectively. Transistors Q 1 to Q 4 are controlled by PWM signals ϕ 1 to ϕ 4 , respectively. When AC voltage VAC is a positive voltage, transistor Q 3 is turned on while transistors Q 1 and Q 4 are alternately turned on. Furthermore, when AC voltage VAC is a negative voltage, transistor Q 4 is turned on while transistors Q 2 and Q 3 are alternately turned on.

Since other configurations and operations are the same as those in the first embodiment, the description thereof will not be repeated. The same effect as that in the first embodiment can be achieved also in the present second embodiment.

Third Embodiment

FIG. 12 is a circuit diagram showing the configuration of a converter according to the third embodiment of the present invention, which is shown as compared with FIG. 1 . Referring to FIG. 12 , this converter is different from the converter in FIG. 1 in that: the drains of transistors Q 3 and Q 4 are separated from the cathodes of diodes D 3 and D 4 ; the drain of transistor Q 3 and the cathode of diode D 4 are connected to each other; and the drain of transistor Q 4 and the cathode of diode D 3 are connected to each other.

Transistors Q 1 to Q 4 are controlled by PWM signals ϕ 1 to ϕ 4 , respectively. When AC voltage VAC is a positive voltage, transistor Q 3 is turned on while transistors Q 1 and Q 4 are alternately turned on. Also, when AC voltage VAC is a negative voltage, transistor Q 4 is turned on while transistors Q 2 and Q 3 are alternately turned on.

Since other configurations and operations are the same as those in the first embodiment, the description thereof will not be repeated. The same effect as that in the first embodiment can be achieved also in the present third embodiment.

Fourth Embodiment

FIG. 13 is a circuit block diagram showing the configuration of inverter 3 included in an uninterruptible power supply device according to the fourth embodiment of the present invention. The entire configuration of the uninterruptible power supply device is as shown in FIG. 7 . Converter 2 included in the uninterruptible power supply device corresponds to a converter shown in FIG. 1, 11 , or 12 . In FIG. 13 , this inverter 3 includes input terminals T 11 to T 13 , an output terminal T 14 , transistors Q 5 to Q 8 , and diodes D 5 to D 8 .

Input terminals T 11 to T 13 are connected to DC positive bus L 1 , DC negative bus L 2 , and DC neutral point bus L 3 , respectively, in FIG. 7 . Battery B 1 has a positive electrode and a negative electrode that are connected to input terminals T 11 and T 13 , respectively. Battery B 2 has a positive electrode and a negative electrode that are connected to input terminals T 13 and T 12 , respectively. Each of batteries B 1 and B 2 outputs a DC voltage. The output voltage of battery B 1 and the output voltage of battery B 2 are equal to each other. Accordingly, DC voltages V 1 , V 2 and V 3 are applied to input terminals T 11 , T 12 , and T 13 , respectively, which leads to conditions of V 1 >V 3 >V 2 and V 3 =(V 1 +V 2 )/2. This inverter serves to convert DC voltages V 1 to V 3 , which have been applied to input terminals T 11 to T 13 , into a three-level AC voltage V 4 , and outputs the converted three-level AC voltage V 4 through output terminal T 14 . In addition, if input terminal T 13 is grounded, DC voltages V 1 , V 2 and V 3 are to be set at a positive voltage, a negative voltage and 0V, respectively.

Each of transistors Q 5 and Q 6 is an N-channel MOS transistor formed using SiC (silicon carbide) that is a wide band gap semiconductor. The rated current in each of transistors Q 5 and Q 6 is, for example, 600 A that is larger than the rated current in each of transistors Q 7 , Q 8 and diodes D 5 to D 8 .

Each of transistors Q 7 and Q 8 is an IGBT formed using Si (silicon) that is a semiconductor other than a wide band gap semiconductor. The rated current in each of transistors Q 7 and Q 8 is 450 A, for example.

Each of diodes D 5 and D 6 is formed using Si (silicon) that is a semiconductor other than a wide band gap semiconductor. The rated current in each of diodes D 5 and D 6 is 300 A, for example.

Each of diodes D 7 and D 8 is a Schottky barrier diode formed using SiC (silicon carbide) that is a wide band gap semiconductor. The rated current in each of diodes D 7 and D 8 is 500 A, for example.

In this way, transistors Q 5 and Q 6 are different in specification from transistors Q 7 and Q 8 , and diodes D 5 and D 6 are different in specification from diodes D 7 and D 8 , the reason for which will be described later.

Transistor Q 5 has: a drain (the first electrode) connected to input terminal T 11 (the first output terminal); and a source (the second electrode) connected to output terminal T 14 (the fourth output terminal). Diode D 5 has: an anode connected to output terminal T 14 ; and a cathode connected to input terminal T 11 .

Transistor Q 6 has: a drain connected to output terminal T 14 ; and a source connected to input terminal T 12 (the second output terminal). Diode D 6 has: an anode connected to input terminal T 12 ; and a cathode connected to output terminal T 14 . In other words, diodes D 5 and D 6 are connected in anti-parallel to transistors Q 5 and Q 6 , respectively.

Transistors Q 7 and Q 8 have collectors (the first electrodes) that are connected to each other. Transistors Q 7 and Q 8 have emitters (the second electrodes) that are connected to input terminal T 13 (the third output terminal) and output terminal T 14 , respectively. Diodes D 7 and D 8 have: cathodes that are connected to collectors of transistors Q 7 and Q 8 ; and anodes that are connected to input terminal T 13 and output terminal T 14 , respectively. In other words, diodes D 7 and D 8 are connected in anti-parallel to transistors Q 7 and Q 8 , respectively. Transistors Q 7 , Q 8 and diodes D 7 , D 8 form the second bidirectional switch.

Then, the operation of this inverter will be hereinafter described. PWM signals ϕ 5 to ϕ 8 are supplied to the gates of transistors Q 5 to Q 8 , respectively. FIGS. 14( a ) to 14( e ) each show a method of generating PWM signals ϕ 5 to ϕ 8 and waveforms. Specifically, FIG. 14( a ) shows waveforms of sinusoidal wave command value signal CM, positive-side triangular wave carrier signal CA 1 and negative-side triangular wave carrier signal CA 2 , and FIGS. 14( b ), 14( c ), 14( d ), and 14( e ) show waveforms of PWM signals ϕ 5 , ϕ 8 , ϕ 7 , and ϕ 6 , respectively.

In FIGS. 14( a ) to 14( e ) , the frequency of sinusoidal wave command value signal CM is a commercial frequency, for example. Carrier signals CA 1 and CA 2 have the same cycle and the same phase. The cycles of carrier signals CA 1 and CA 2 are sufficiently smaller than the cycle of sinusoidal wave command value signal CM.

The high-low levels of sinusoidal wave command value signal CM and positive-side triangular wave carrier signal CA 1 are compared with each other. When the level of sinusoidal wave command value signal CM is higher than the level of positive-side triangular wave carrier signal CA 1 , PWM signals ϕ 5 and ϕ 7 are set at an “H” level and an “L” level, respectively. When the level of sinusoidal wave command value signal CM is lower than the level of positive-side triangular wave carrier signal CA 1 , PWM signals ϕ 5 and ϕ 7 are set at an “L” level and an “H” level, respectively.

Accordingly, in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals ϕ 5 and ϕ 7 are alternately set at an “H” level in synchronization with carrier signal CA 1 , and thus, transistors Q 5 and Q 7 are alternately turned on. Furthermore, in a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 5 and ϕ 7 are fixed at an “L” level and an “H” level, respectively, so that transistor Q 5 is fixed in an OFF state and transistor Q 7 is fixed in an ON state.

The high-low levels of sinusoidal wave command value signal CM and negative-side triangular wave carrier signal CA 2 are compared with each other. When the level of sinusoidal wave command value signal CM is higher than the level of positive-side triangular wave carrier signal CA 2 , PWM signals ϕ 6 and ϕ 8 are set at an “L” level and an “H” level, respectively. When the level of sinusoidal wave command value signal CM is lower than the level of positive-side triangular wave carrier signal CA 2 , PWM signals ϕ 6 and ϕ 8 are set at an “H” level and an “L” level, respectively.

Accordingly, in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals ϕ 6 and ϕ 8 are fixed at the “L” level and the “H” level, respectively, and thus, transistor Q 6 is fixed in the OFF state and transistor Q 8 is fixed in the ON state. Furthermore, in a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 6 and ϕ 8 are alternately set at an “H” level in synchronization with carrier signal CA 2 , and transistors Q 6 and Q 8 are alternately turned on.

The ratio between the time in which the PWM signal is set at an “H” level within one cycle and the time of one cycle of the PWM signal is referred to as a duty ratio. In a time period during which the level of sinusoidal wave command value signal CM is positive, the duty ratio of PWM signal ϕ 5 is maximized in the vicinity of a positive peak (90 degrees) of sinusoidal wave command value signal CM. Also, the duty ratio of PWM signal ϕ 5 decreases with increasing distance from the peak, and reaches 0 in the vicinity of 0 degree and 180 degrees. The duty ratio of PWM signal ϕ 5 is fixed at 0 in a time period during which sinusoidal wave command value signal CM is negative. PWM signal ϕ 7 is a complementary signal of PWM signal ϕ 5 .

The duty ratio of PWM signal ϕ 6 is fixed at 0 in a time period during which the level of sinusoidal wave command value signal CM is positive. The duty ratio of PWM signal ϕ 6 is maximized in the vicinity of a negative peak (270 degrees) of sinusoidal wave command value signal CM. Also, the duty ratio of PWM signal ϕ 6 decreases with increasing distance from the peak, and reaches 0 in the vicinity of 180 degrees and 360 degrees. PWM signal ϕ 8 is a complementary signal of PWM signal ϕ 6 .

Then, the current flowing through each of transistors Q 5 to Q 8 and diodes D 5 to D 8 during the operation of the inverter will be hereinafter described. As shown in FIG. 15 , the current flowing from input terminal T 11 into output terminal T 14 is defined as I 5 ; the current flowing from output terminal T 14 into input terminal T 12 is defined as I 6 ; the current flowing from input terminal T 13 into output terminal T 14 is defined as I 7 ; and the current flowing from output terminal T 14 into input terminal T 13 is defined as I 8 .

FIGS. 16( a ) to 16( i ) are time charts each showing the operation of an inverter. Specifically, FIG. 16( a ) shows waveforms of sinusoidal wave command value signal CM, positive-side triangular wave carrier signal CA 1 and negative-side triangular wave carrier signal CA 2 . FIGS. 16( b ), 16( d ), 16( f ), and 16( h ) show waveforms of PWM signals ϕ 5 , ϕ 8 , ϕ 7 , and ϕ 6 , respectively. FIGS. 16( c ), 16( e ), 16( g ), and 16( i ) show waveforms of currents I 5 , I 8 , I 7 , and I 6 , respectively. Among currents I 5 to I 8 , a positive current shows a current flowing through transistor Q while a negative current shows a current flowing through diode D. The figures also show the case where the power factor is 1.0.

In FIGS. 16( a ) to 16( i ) , in a time period during which the level of sinusoidal wave command value signal CM is positive, PWM signals ϕ 8 and ϕ 6 are fixed at an “H” level and an “L” level, respectively, and PWM signals ϕ 5 and ϕ 7 are alternately set at an “H” level. Accordingly, transistors Q 8 and Q 6 are fixed in the ON state and the OFF state, respectively, and transistors Q 5 and Q 7 are alternately turned on. Thus, DC voltages V 1 and V 3 alternately appear on output terminal T 14 .

During this time period, current I 5 of the level corresponding to the ON time of transistor Q 5 flows when transistor Q 5 is turned on, and current I 7 of the level complementing current I 5 in a passage of diode D 7 and transistor Q 8 flows when transistor Q 5 is turned off.

Since transistor Q 6 is fixed in the OFF state, no current flows through transistor Q 6 , so that switching loss does not occur in transistor Q 6 . Transistor Q 7 is turned on/off, but a current flows through diode D 7 while no current flows through transistor Q 7 , so that switching loss does not occur in transistor Q 7 . Since transistor Q 8 is fixed in the ON state, a current flows through transistor Q 8 , but switching loss does not occur in transistor Q 8 . Accordingly, during this time period, among transistors Q 5 to Q 8 , the current flowing through transistor Q 5 shows the largest effective value while the greatest switching loss occurs in transistor Q 5 .

Each time transistor Q 5 is changed from the OFF state to the ON state, a reverse bias voltage is applied to diode D 7 , and this diode D 7 performs a reverse recovery operation. No current flows through other diodes D 5 , D 6 and D 8 during this time period.

In a time period during which the level of sinusoidal wave command value signal CM is negative, PWM signals ϕ 7 and ϕ 5 are fixed at an “H” level and an “L” level, respectively, and PWM signals ϕ 6 and ϕ 8 are alternately set at an “H” level. Accordingly, transistors Q 7 and Q 5 are fixed in the ON state and the OFF state, respectively, and transistors Q 6 and Q 8 are alternately turned on. Thus, DC voltages V 2 and V 3 appear alternately on output terminal T 14 .

During this time period, current I 6 of the level corresponding to the ON time of transistor Q 6 flows when transistor Q 6 is turned on, and current I 7 flows through a passage of diode D 8 and transistor Q 7 when transistor Q 6 is turned off.

Since transistor Q 5 is fixed in the OFF state, a current does not flow through transistor Q 5 , so that switching loss does not occur in transistor Q 5 . Although transistor Q 8 is turned on/off, a current flows through diode D 8 while a current does not flow through transistor Q 8 , so that switching loss does not occur in transistor Q 8 . Since transistor Q 7 is fixed in the ON state, a current flows through transistor Q 7 , but switching loss does not occur in transistor Q 7 . Accordingly, during this time period, among transistors Q 5 to Q 8 , the current flowing through transistor Q 6 shows the largest effective value while the greatest switching loss occurs in transistor Q 6 .

Furthermore, each time transistor Q 6 is changed from the OFF state to the ON state, a reverse bias voltage is applied to diode D 8 , and this diode D 8 performs a reverse recovery operation. Also, no current flows through other diodes D 5 , D 6 and D 7 during this time period.

In summary, a large current flows through each of transistors Q 5 and Q 6 , and thus, switching loss occurs in transistors Q 5 and Q 6 . The current flowing through transistors Q 7 and Q 8 is smaller than the current flowing through transistors Q 5 and Q 6 , and thus, switching loss does not occur in transistors Q 7 and Q 8 .

Accordingly, as described above, as transistors Q 5 and Q 6 , an N-channel MOS transistor is employed that is formed of SiC as a wide band gap semiconductor and that has a rated current of a relatively large value (for example, 600 A), thereby reducing switching loss. Also, as transistors Q 7 and Q 8 , an IGBT is employed that is formed of Si as a semiconductor other than a wide band gap semiconductor and that has a rated current of a relatively small value (for example, 450 A), thereby reducing cost.

A current flowing through diodes D 7 and D 8 is comparable to the current flowing through transistors Q 7 and Q 8 . These diodes D 7 and D 8 each perform a reverse recovery operation. No current flows through diodes D 5 and D 6 . In addition, as is well known, diodes D 5 and D 6 are provided in order to protect transistors Q 5 and Q 6 from the voltage generated in the inductor when this inductor is used as a load.

Accordingly, as described above, as diodes D 7 and D 8 , a Schottky barrier diode is employed that is formed of SiC as a wide band gap semiconductor and that has a rated current of a value comparable to those of transistors Q 7 and Q 8 (for example, 500 A), thereby reducing the recovery loss occurring during the reverse recovery operation. As diodes D 5 and D 6 , a diode is employed that is formed of Si as a semiconductor other than a wide band gap semiconductor and that has a rated current of a relatively small value (for example, 300 A), thereby reducing cost.

FIG. 17 is a diagram showing the external appearance of inverter 3 shown in FIG. 13 . In FIG. 17 , inverter 3 includes one semiconductor module M 21 . Semiconductor module M 21 is provided on its inside with transistors Q 5 to Q 8 and diodes D 5 to D 8 . Semiconductor module M 21 is provided on its outside with input terminals T 11 to T 13 and an output terminal T 14 . Furthermore, semiconductor module M 21 is provided on its outside with four signal terminals used for supplying PWM signals ϕ 5 to ϕ 8 to the gates of transistors Q 5 to Q 8 , respectively, but these four signal terminals are not shown for simplification of illustration of the figure.

As described above, in the present fourth embodiment, an N-channel MOS transistor formed of a wide band gap semiconductor is used as transistors Q 5 and Q 6 that turn on/off a current, and an IGBT formed of a semiconductor other than a wide band gap semiconductor is used as transistors Q 7 and Q 8 that do not turn on/off a current. Accordingly, the switching loss and the cost can be reduced.

Furthermore, a Schottky barrier diode formed of a wide band gap semiconductor is used as diodes D 7 and D 8 performing a reverse recovery operation, and a diode formed of a semiconductor other than a wide band gap semiconductor is used as diodes D 5 and D 6 not performing a reverse recovery operation. Accordingly, the recovery loss and the cost can be reduced.

In addition, in the present fourth embodiment, SiC is used as a wide band gap semiconductor, but not limited thereto, and any other semiconductors may be used as long as such semiconductors are a wide band gap semiconductor. For example, GaN (gallium nitride) may be used as a wide band gap semiconductor.

FIG. 18 is a block diagram showing a modification of the fourth embodiment, which is shown as compared with FIG. 17 . In FIG. 18 , in the present modification, inverter 3 includes a substrate BP 11 , and two semiconductor modules M 22 and M 23 mounted on its surface. Semiconductor module M 22 is provided on its inside with transistors Q 5 , Q 6 and diodes D 5 , D 6 . Semiconductor module M 22 is provided on its outside with input terminals T 11 , T 12 and an output terminal T 14 . Furthermore, semiconductor module M 22 is provided on its outside with two signal terminals (not shown) used for supplying PWM signals ϕ 5 and ϕ 6 to the gates of transistors Q 5 and Q 6 .

Semiconductor module M 23 is provided on its inside with transistors Q 7 , Q 8 and diodes D 7 , D 8 . Semiconductor module M 23 is provided on its outside with an input terminal T 13 and an output terminal T 14 . Semiconductor module M 23 is provided on its outside with two signal terminals (not shown) used for supplying PWM signals ϕ 7 and ϕ 8 to the gates of transistors Q 7 and Q 8 . Output terminal T 14 of semiconductor module M 22 and output terminal T 14 of semiconductor module M 23 are connected to each other. The same effect as that in the fourth embodiment can be achieved also in the present modification.

FIG. 19 is a block diagram showing another modification of the fourth embodiment, which is shown as compared with FIG. 17 . In FIG. 19 , in the present modification, the inverter includes a substrate BP 12 , and two semiconductor modules M 24 and M 25 mounted on its surface. Semiconductor module M 24 is provided on its inside with transistors Q 5 to Q 8 . Semiconductor module M 24 is provided on its outside with input terminals T 11 to T 13 , an output terminal T 14 , and an intermediate terminal T 15 . Intermediate terminal T 15 is connected to the emitter of each of transistors Q 7 and Q 8 . Semiconductor module M 24 is provided on its outside with four signal terminals (not shown) used for supplying PWM signals ϕ 5 to ϕ 8 to the gates of transistors Q 5 to Q 8 .

Semiconductor module M 25 is provided on its inside with diodes D 5 to D 8 . Semiconductor module M 25 is provided on its outside with input terminals T 11 to T 13 , an output terminal T 14 , and an intermediate terminal T 15 . Intermediate terminal T 15 is connected to the anode of each of diodes D 7 and D 8 . Terminals T 11 to T 15 of semiconductor module M 24 are connected to terminals T 11 to T 15 , respectively, of semiconductor module M 25 . The same effect as that in the fourth embodiment can be achieved also in the present modification.

FIG. 20 is a block diagram showing still another modification of the fourth embodiment, which is shown as compared with FIG. 17 . In FIG. 20 , in the present modification, an inverter 3 includes a substrate BP 13 and eight semiconductor modules M 31 to M 38 mounted on its surface. Semiconductor modules M 31 to M 34 are provided on their insides with transistors Q 5 to Q 8 , respectively. Semiconductor modules M 35 to M 38 are provided on their insides with diodes D 5 to D 8 , respectively. Each of semiconductor modules M 31 and M 35 includes terminals T 11 and T 14 . Each of semiconductor modules M 32 and M 36 includes terminals T 12 and T 14 . Terminals T 11 of semiconductor modules M 31 and M 35 are connected to each other. Terminals T 12 of semiconductor modules M 32 and M 16 are connected to each other.

Each of semiconductor modules M 33 and M 37 includes terminals T 13 and T 15 , and each of semiconductor modules M 34 and M 38 includes terminals T 14 and T 5 . Terminals T 15 of semiconductor modules M 33 and M 34 are connected to the collectors of transistors Q 7 and Q 8 , respectively. Terminals T 15 of semiconductor modules M 37 and M 38 are connected to the cathodes of diodes D 7 and D 8 , respectively. Terminals T 13 of semiconductor modules M 33 and M 37 are connected to each other. Terminals T 15 of semiconductor modules M 33 , M 34 , M 37 , and M 38 are connected to each other. Terminals T 14 of semiconductor modules M 31 , M 32 , M 34 to M 36 , and M 38 are connected to each other. Furthermore, semiconductor modules M 31 to 34 are provided on their outsides with four signal terminals (not shown) used for supplying PWM signals ϕ 5 and ϕ 8 to the gates of transistors Q 5 to Q 8 . The same effect as that in the first embodiment can be achieved also in the present modification.

Fifth Embodiment

FIG. 21 is a circuit diagram showing the configuration of an inverter according to the fifth embodiment of the present invention, which is shown as compared with FIG. 13 . Referring to FIG. 21 , this inverter is different from inverter 3 in FIG. 13 in that the parallel connection body formed of transistor Q 7 and diode D 7 and the parallel connection body formed of transistor Q 8 and diode D 8 are replaced with each other.

Transistors Q 7 and Q 8 have: emitters that are connected to each other; and collectors that are connected to an input terminal T 13 and an output terminal T 14 , respectively. Transistors Q 5 to Q 8 are controlled by PWM signals ϕ 5 to ϕ 8 , respectively. When DC voltages V 1 and V 3 are alternately output through output terminal T 14 , transistor Q 8 is turned on while transistors Q 5 and Q 7 are alternately turned on. Furthermore, when DC voltages V 2 and V 3 are alternately output through output terminal T 14 , transistor Q 7 is turned on while transistors Q 6 and Q 8 are alternately turned on.

Since other configurations and operations are the same as those in the fourth embodiment, the description thereof will not be repeated. The same effect as that in the fourth embodiment can be achieved also in the present fifth embodiment.

Sixth Embodiment

FIG. 22 is a circuit diagram showing the configuration of an inverter according to the sixth embodiment of the present invention, which is shown as compared with FIG. 13 . Referring to FIG. 22 , this inverter is different from the inverter in FIG. 13 in that: the collectors of transistors Q 7 and Q 8 are separated from the cathodes of diodes D 7 and D 8 ; the collector of transistor Q 7 and the cathode of diode D 8 are connected to each other; and the collector of transistor Q 8 and the cathode of diode D 7 are connected to each other.

Transistors Q 5 to Q 8 are controlled by PWM signals ϕ 5 to ϕ 8 , respectively. When DC voltages V 1 and V 3 are alternately output through output terminal T 14 , transistor Q 8 is turned on while transistors Q 5 and Q 7 are alternately turned on. Furthermore, when DC voltages V 2 and V 3 are alternately output through output terminal T 14 , transistor Q 7 is turned on while transistors Q 6 and Q 8 are alternately turned on.

Since other configurations and operations are the same as those in the fourth embodiment, the description thereof will not be repeated. The same effect as that in the fourth embodiment can be achieved also in the present sixth embodiment.

It should be understood that the embodiments disclosed herein are illustrative and non-restrictive in every respect. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the meaning and scope equivalent to the terms of the claims.

T 0 , T 11 to T 13 input terminal, T 1 to T 3 , T 14 output terminal, T 4 , T 14 , T 15 intermediate terminal, Q 1 to Q 8 transistor, D 1 to D 8 diode, B 1 , B 2 battery, M 1 to M 5 , M 11 to M 18 , M 21 to M 25 , M 31 to M 38 semiconductor module, BP 1 to BP 3 , BP 11 to BP 13 substrate, 1 input filter, 2 converter, L 1 DC positive bus, L 2 DC negative bus, L 3 DC neutral point bus, C 1 , C 2 capacitor, 3 inverter, 4 output filter, 5 controller, 10 commercial AC power supply, 11 load.