Transformer-type phase shifter, phase-shift circuit, and communication terminal apparatus

This application claims the benefit of priority to Japanese Patent Application No. 2015-045452 filed on Mar. 9, 2015 and Japanese Patent Application No. 2015-006266 filed on Jan. 15, 2015 and is a Continuation Application of PCT Application No. PCT/JP2016/050183 filed on Jan. 6, 2016. The entire contents of each application are hereby incorporated herein by reference.

1. Field of the Invention

The present invention relates to a transformer-type phase shifter connected between an antenna and a feeder circuit, a phase-shift circuit, and a communication terminal apparatus including the transformer-type phase shifter or the phase-shift circuit.

2. Description of the Related Art

For example, a communication terminal apparatus typified by a mobile phone is typically required to have a high-frequency circuit that is small and highly efficient.

In a high-frequency circuit, a phase-shift circuit that rotates a phase angle in order to provide a transmission signal with a predetermined phase to the following-stage (next-stage) circuit is used as a circuit component.

If, for example, a phase-shift circuit and an antenna matching circuit are provided between a high-frequency power amplifier and an antenna in a transmission circuit, and a phase-shift amount of the phase-shift circuit is appropriately set for matching between the power amplifier and the antenna matching circuit, it is possible increase the efficiency of the power amplifier and to suppress the generation of harmonics.

In the related art, for example, a transmission-line-type phase shifter disclosed in Japanese Unexamined Patent Application Publication No. 2003-32003 is advantageous in terms of an insertion-loss characteristic and frequency dependency, but is disadvantageous in terms of a large size if the phase-shift amount is increased. In addition, because of the frequency dependency of the phase-shift amount, the transmission-line-type phase shifter is difficult to apply to a wide-band high-frequency circuit.

On the other hand, a phase-shift circuit including components such as a distributed-constant-type delay line is difficult to apply to a wide-band high-frequency circuit because the frequency dependency of the phase-shift amount is high. Accordingly, it is difficult to use such a phase-shift circuit for mobile phone terminals and the like, which are now using a wider band. In addition, unfortunately, a signal insertion loss will be increased if a large phase-shift amount is obtained.

FIG. 15 illustrates a phase-shift characteristic of a transmission line including a phase shifter using a distributed-constant-type delay line of the related art. In FIG. 15 , the horizontal axis represents the frequency, and the vertical axis represents the phase-shift amount. A characteristic line A represents a characteristic of the transmission line alone, and a characteristic line B represents a characteristic of the transmission line provided with a phase shifter. The difference in phase-shift amount between these lines corresponds to a phase-shift amount of the phase shifter. In this example, phase-shift amounts of the transmission line are −30° at a frequency of 1 GHz and −60° at a frequency of 1.9 GHz, whereas phase-shift amounts of the transmission line connected to the phase shifter are −90° at a frequency of 1 GHz and −180° at a frequency of 1.9 GHz. Therefore, phase-shift amounts of the phase shifter alone are −60° at a frequency of 1 GHz and −120° at a frequency of 1.9 GHz.

In the above manner, both the transmission line and the phase shifter such as the distributed-constant-type delay line have a high dependency on the frequency. That is, the larger the necessary phase-shift amount is, the more difficult it is to apply to wide-band use. In addition, in both the transmission line and the distributed-constant-type delay line, the insertion loss notably increases with an increase in the frequency. In the distributed-constant-type delay line, the increase in the insertion loss at high frequencies is particularly notable. Accordingly, the insertion loss is increased if the phase-shift amount is to be increased.

Preferred embodiments of the present invention provide a transformer-type phase shifter, a phase-shift circuit, and a communication terminal apparatus including the transformer-type phase shifter or the phase-shift circuit, which are advantageous in reducing the size, loss, and frequency dependency of the phase-shift amount.

A transformer-type phase shifter according to a preferred embodiment of the present invention includes a transformer including a first coil and a second coil that is magnetically coupled to the first coil with a coupling coefficient of less than 1, the transformer including a parallel inductance component and a series inductance component, and an impedance adjustment circuit including a reactance element that provides impedance matching for the transformer.

It is preferable that the impedance adjustment circuit include a first capacitance element connected in parallel to the first coil, a second capacitance element connected in parallel to the second coil, and a third capacitance element connected between the first coil and the second coil. At least one of the first capacitance element, the second capacitance element, and the third capacitance element may be defined by a stray capacitance of the transformer alone or by an actual component.

A transformer-type phase shifter according to a preferred embodiment of the present invention includes a transformer including a first coil and a second coil that is magnetically coupled to the first coil with a coupling coefficient of less than 1, the transformer being connected between a first port and a second port, a first capacitance element connected between the first port connected to the transformer and a ground, a second capacitance element connected between the second port connected to the transformer and the ground, and a third capacitance element connected between the first port and the second port connected to the transformer. At least one of the first capacitance element, the second capacitance element, and the third capacitance element may be defined by a stray capacitance of the transformer alone or may be defined by an actual component.

It is possible to shift a phase exceeding 180°, i.e., exceeding the sum of the phase difference (180°) between an input/output signal of the first coil and an input/output signal of the second coil in the transformer circuit and the phase-shift amount due to the parallel inductance component and the series inductance component in the transformer.

In addition, although the presence of the parallel parasitic inductance component and the series parasitic inductance component in the transformer causes the impedance of the transformer to be different from a predetermined value (e.g., about 50Ω), the reactance element (capacitance elements) enables the impedance to be adjusted.

It is preferable that the transformer be provided in a single stack in which a plurality of base layers are stacked and that the first coil and the second coil be defined by conductor patterns provided on the base layers. Accordingly, it becomes easy to mount a phase shifter as a single component on a communication terminal apparatus or the like because the phase shifter may be mounted on a printed wiring board or the like.

It is preferable that the first coil and the second coil have the same or substantially the same inside diameter, and the same or substantially the same outside diameter and that coil axes thereof have a co-axial relationship. Accordingly, although the first coil and the second coil have a small number of turns, that is, although the first coil and the second coil are small, a transformer with an appropriately high coupling coefficient is able to be obtained.

It is preferable that the third capacitance element mainly include a capacitance between the first coil and the second coil. Accordingly, it is unnecessary to provide a pattern to define the third capacitance element, or the third capacitance element is unnecessary as a component, thus reducing the size and cost.

It is preferable that the first capacitance element be mainly defined by a capacitance between wires of the first coil and that the second capacitance element be mainly defined by a capacitance between wires of the second coil. Accordingly, it is unnecessary to provide patterns to define first capacitance element and the second capacitance element, or the first capacitance element and the second capacitance element are unnecessary as components, thus reducing the size and cost.

It is possible that a transformer ratio between the first coil and the second coil is 1:n (where n is a value other than 1). Accordingly, the transformer is able to shift the phase and also is able to convert the impedance, thus also having the function of an impedance matching circuit between a circuit connected to the first port and a circuit connected to the second port.

A phase-shift circuit according to a preferred embodiment of the present invention includes the transformer-type phase shifter according to any one of the above-described preferred embodiments of the present invention and a phase-shift line that is connected in series to the transformer-type phase shifter with a phase-shift amount of less than about 90°, for example. This configuration enables a predetermined phase shift in a wide range exceeding about 180° in a finely adjusted manner. The phase-shift line may be defined by a substrate and a conductor pattern provided on or in the substrate.

A communication terminal apparatus according to a preferred embodiment of the present invention is a communication terminal apparatus including a feeder circuit and an antenna element that is connected to the feeder circuit, the feeder circuit including a power amplifying circuit, and the communication terminal apparatus includes the transformer-type phase shifter according to any one of the above-described preferred embodiments of the present invention or the phase-shift circuit according to the above-described preferred embodiment of the present invention between the feeder circuit and the antenna element. Accordingly, the efficiency is able to be increased or the size is able to be reduced in a transmission circuit system including an antenna.

According to a transformer-type phase shifter according to a preferred embodiment of the present invention, it is possible to obtain a large phase-shift amount of greater than about 180° in spite of a small size and a small loss. In addition, it is possible to obtain a phase-shift characteristic with a low frequency dependency of the phase-shift amount. In addition, by including the impedance adjustment circuit, it is possible to finely adjust the phase-shift amount.

According to a phase-shift circuit according to a preferred embodiment of the present invention, it is possible to easily obtain a phase-shift amount even greater than 180° while keeping the size small, the loss small, and the frequency dependency of the phase-shift amount low.

According to a communication terminal apparatus according to a preferred embodiment of the present invention, it is possible to increase efficiency and reduce the size in a transmission circuit system including the antenna.

The above and other elements, features, steps, characteristics and advantages of the present invention will become more apparent from the following detailed description of the preferred embodiments with reference to the attached drawings.

FIG. 1 is a circuit diagram of a transformer-type phase shifter 11 according to a first preferred embodiment of the present invention.

FIGS. 2A and 2B are each an equivalent circuit diagram of a transformer T.

FIG. 3 is an external appearance perspective view of the transformer-type phase shifter 11 .

FIG. 4 is a plan view of each layer in the transformer-type phase shifter 11 .

FIG. 5 is a vertical cross-sectional view of the transformer-type phase shifter 11 .

FIG. 6A is a circuit diagram illustrating the transformer-type phase shifter 11 , and FIG. 6B is an equivalent circuit diagram illustrating the transformer-type phase shifter 11 .

FIG. 7 is a Smith chart illustrating a phase-shift function of the transformer T.

FIG. 8 illustrates a frequency characteristic of a phase-shift amount of the transformer-type phase shifter 11 .

FIG. 9 illustrates a frequency characteristic of an insertion loss of the transformer-type phase shifter 11 .

FIGS. 10A and 10B are block diagrams illustrating configurations of phase-shift circuits 30 A and 30 B according to a second preferred embodiment of the present invention.

FIGS. 11A, 11B, 11C, and 11D illustrate circuit configurations of four phase-shift circuits 31 , 32 , 33 , and 34 according to the second preferred embodiment of the present invention.

FIG. 12 is a circuit diagram of a transformer-type phase shifter 13 according to a third preferred embodiment of the present invention.

FIG. 13 is a circuit diagram of a transformer-type phase shifter 14 according to a fourth preferred embodiment of the present invention.

FIG. 14 is a block diagram of a communication terminal apparatus 200 according to a fifth preferred embodiment of the present invention.

FIG. 15 illustrates a phase-shift characteristic of a transmission line including a phase shifter using a distributed-constant-type delay line of the related art.

Hereinafter, a plurality of preferred embodiments of the present invention will be described by taking some specific examples with reference to drawings. In the drawings, like numerals denote like components. In a second preferred embodiment and the following preferred embodiments, elements or features that are common to those in a first preferred embodiment will be omitted from description, and different elements or features will be described. In particular, like functions and effects obtained by like configurations will not be referred to in each of the preferred embodiments.

First Preferred Embodiment

FIG. 1 is a circuit diagram of a transformer-type phase shifter 11 according to a first preferred embodiment of the present invention. The transformer-type phase shifter 11 includes a transformer T. The transformer T includes a first coil L 1 and a second coil L 2 that is magnetically coupled to the first coil L 1 with a coupling coefficient of less than 1. The transformer-type phase shifter 11 further includes an impedance adjustment circuit including a first capacitance element C 1 , a second capacitance element C 2 , and a third capacitance element C 3 .

The first capacitance element C 1 is connected in parallel to the first coil L 1 , and the second capacitance element C 2 is connected in parallel to the second coil L 2 . The third capacitance element C 3 is connected between the first coil L 1 and the second coil L 2 .

FIGS. 2A and 2B are each an equivalent circuit diagram of the transformer T. The equivalent circuit of the transformer T can be represented in several forms. FIG. 2A illustrates an ideal transformer IT, an inductor La that is connected in series to the upstream side of the ideal transformer IT, an inductor Lb that is connected in parallel to the upstream side of the ideal transformer IT, and an inductor Lc that is connected in series to the downstream side of the ideal transformer IT.

FIG. 2B illustrates the ideal transformer IT, two inductors, the inductor La and an inductor Lc 1 , which are connected in series to the upstream side of the ideal transformer IT, and the inductor Lb that is connected in parallel to the upstream side of the ideal transformer IT.

Here, if the transformer ratio of the transformer T is denoted by 1:n, the coupling coefficient between the first coil L 1 and the second coil L 2 (see FIG. 1 ) is denoted by k, the inductance of the first coil L 1 is denoted by L 1 , and the inductance of the second coil L 2 is denoted by L 2 , the inductances of the inductors La, Lb, and Lc have the following relationship.

La: L1(1−k)

Lb: k*L1

Lc: L2(1−k)

Lc1: n 2 *L2*(1−k)

The transformer ratio of the ideal transformer corresponds to a transformer ratio according to the number of turns of the first coil L 1 and the second coil L 2 .

In any case, in the transformer T according to this preferred embodiment, since the coupling coefficient k between the first coil L 1 and the second coil L 2 is less than 1, a series inductance component and a parallel inductance component are generated.

FIG. 3 is an external appearance perspective view of the transformer-type phase shifter 11 according to this preferred embodiment, and FIG. 4 is a plan view of each layer in the transformer-type phase shifter 11 . FIG. 5 is a cross-sectional view of the transformer-type phase shifter 11 . The transformer-type phase shifter 11 includes a plurality of insulating bases S 1 to S 9 . Various conductor patterns are provided on the bases S 1 to S 9 . The phrase “various conductor patterns” refers to, not only a conductor pattern provided on the surface of the base, but also an interlayer connecting conductor. The interlayer connecting conductor includes, not only a via-conductor, but also an end-surface electrode provided on an end surface of a stack 100 .

The top surface of the base S 1 corresponds to a surface (bottom surface) on which the stack 100 is mounted. On the base S 1 , a terminal T 1 as a first port P 1 , a terminal T 2 as a second port P 2 , a ground terminal GND, and an open terminal NC are provided.

On the bases S 5 and S 4 , conductors L 1 A 1 and L 1 A 2 are provided, respectively. On the base S 3 , conductors L 1 A 3 and L 1 B 1 are provided. On the base S 2 , conductors L 1 B 2 and L 1 C are provided.

A first end of the conductor L 1 A 1 is connected to the terminal T 1 defining and functioning as the first port. A second end of the conductor L 1 A 1 is connected to a first end of the conductor L 1 A 2 via a via-conductor V 1 . A second end of the conductor L 1 A 2 is connected to a first end of the conductor L 1 A 3 via a via-conductor V 2 . A second end of the conductor L 1 A 3 is connected to a first end of the conductor L 1 B 1 . The second end of the conductor L 1 A 3 and the first end of the L 1 B 1 are connected to a first end of the conductor L 1 B 2 via a via-conductor V 3 . A second end of the conductor L 1 B 1 is connected to a second end of the conductor L 1 B 2 via a via-conductor V 4 . The second end of the conductor L 1 B 2 is connected to a first end of the conductor L 1 C. A second end of the conductor L 1 C is connected to the ground terminal GND.

Conductors L 2 A 1 and L 2 A 2 are provided on the bases S 6 and S 7 , respectively. Conductors L 2 A 3 and L 2 B 1 are provided on the base S 8 . Conductors L 2 B 2 and L 2 C are provided on the base S 9 .

A first end of the conductor L 2 A 1 is connected to the terminal T 2 defining and functioning as the second port. A second end of the conductor L 2 A 1 is connected to a first end of the conductor L 2 A 2 via a via-conductor V 5 . A second end of the conductor L 2 A 2 is connected to a first end of the conductor L 2 A 3 via a via-conductor V 6 . A second end of the conductor L 2 A 3 is connected to a first end of the conductor L 2 B 1 . The second end of the conductor L 2 A 3 and the first end of the conductor L 2 B 1 are connected to a first end of the conductor L 2 B 2 via a via-conductor V 7 . A second end of the conductor L 2 B 1 is connected to a second end of the conductor L 2 B 2 via a via-conductor V 8 . The second end of the conductor L 2 B 2 is connected to a first end of the conductor L 2 C. A second end of the conductor L 2 C is connected to the ground terminal GND.

The conductors L 1 A 1 , L 1 A 2 , L 1 A 3 , L 1 B 1 , L 1 B 2 , and L 1 C and via-conductors V 1 , V 2 , V 3 , and V 4 define the first coil L 1 . The conductors L 2 A 1 , L 2 A 2 , L 2 A 3 , L 2 B 1 , L 2 B 2 , and L 2 C and via-conductors V 5 , V 6 , V 7 , and V 8 define the second coil L 2 . Both of the first coil L 1 and the second coil L 2 preferably are rectangular or substantially rectangular helical coils, for example.

The base layers in the stack 100 may be a non-magnetic ceramic stack including low-temperature co-fired ceramics (LTCC) or the like or a resin stack including a resin material such as polyimide or a liquid crystal polymer, for example. The base layers are formed of a non-magnetic material (not magnetic ferrite) as described above and thus can be applied to a transformer and a phase shifter having a predetermined inductance and a predetermined coupling coefficient even in a high-frequency band over 700 MHz.

The above-described conductor patterns and interlayer connecting conductors are preferably made of a conductor material including Ag or Cu as a main component and having a low resistivity. If the base layers are ceramics, for example, the conductor patterns and interlayer connecting conductors are preferably formed by screen printing and firing of conductive paste including Ag or Cu as a main component. If the base layers are resin, for example, the conductor patterns and interlayer connecting conductors are preferably patterned by etching a metal foil such as an Al foil or a Cu foil.

The first coil L 1 and the second coil L 2 preferably have the same or substantially the same inside diameter and the same or substantially the same outside diameter and the same coil winding axis CA (coaxial). Accordingly, although the first coil L 1 and the second coil L 2 have a small number of turns, that is, although the first coil L 1 and the second coil L 2 are small, a transformer with an appropriately high coupling coefficient k is obtained. In addition, it is possible to generate a high capacitance between the first coil L 1 and the second coil L 2 , and the third capacitance element C 3 having predetermined capacitance is able to be provided.

FIG. 6A is a circuit diagram illustrating the transformer-type phase shifter 11 according to this preferred embodiment. Here, the first coil L 1 and the second coil L 2 define a transformer.

The first capacitance element C 1 is mainly defined by stray capacitance between conductor layers located on the bases S 2 , S 3 , S 4 , and S 5 . Similarly, the second capacitance element C 2 is mainly defined by stray capacitance between conductor layers located on the bases S 6 , S 7 , S 8 , and S 9 . In addition, the third capacitance element C 3 is mainly defined by stray capacitance between the first coil L 1 and the second coil L 2 and, in particular, is mainly defined by capacitance between the conductor L 1 A 1 and the conductor L 2 A 1 .

By reducing the distance between the plurality of conductor layers defining the first coil L 1 and the second coil L 2 , if the capacitance elements C 1 , C 2 , and C 3 are defined only of stray capacitance, it is possible to increase the self-inductance and the mutual inductance of the first coil L 1 and the second coil L 2 .

Note that at least one of the first capacitance element C 1 , the second capacitance element C 2 , and the third capacitance element C 3 may be defined by stray capacitance and actual components.

FIG. 6B is an equivalent circuit diagram illustrating the transformer-type phase shifter 11 in which the ideal transformer IT and parasitic inductance are separated from each other. The inductors La and Lc denote series parasitic inductance components, and the inductor Lb denotes a parallel parasitic inductance component.

Although the parasitic inductance components (inductors La, Lb, and Lc) cause the inductance of the transformer to be different from a predetermined value (e.g., about 50Ω), by including the capacitance elements C 1 , C 2 , and C 3 , the impedance of the transformer is adjusted to be the predetermined value. In particular, the capacitance elements C 1 and C 2 have a function of correcting the deviation in the impedance due to the parallel parasitic inductance component Lb, and the capacitance element C 3 has a function of correcting the deviation in the impedance due to the series parasitic inductance components La and Lc.

Note that the positions of the via-conductors V 3 and V 4 illustrated in FIG. 4 determine the ratio of a parallel connection portion of the conductors L 1 B 1 and L 1 B 2 in the first coil L 1 illustrated in FIG. 6A . Similarly, the positions of the via-conductors V 7 and V 8 illustrated in FIG. 4 determine the ratio of a parallel connection portion of the conductors L 2 B 1 and L 2 B 2 in the second coil L 2 illustrated in FIG. 6A . Accordingly, the inductance of the first coil L 1 is able to be finely adjusted depending on the positions of the via-conductors V 3 and V 4 , and the inductance of the second coil L 2 is able to be finely adjusted depending on the positions of the via-conductors V 7 and V 8 .

Current flows in a dispersed manner in the parallel connection portion of the conductors L 1 B 1 and L 1 B 2 , whereas current does not disperse in this manner in the conductor L 1 A 1 . Similarly, current flows in a dispersed manner in the parallel connection portion of the conductors L 2 B 1 and L 2 B 2 , whereas current does not disperse in this manner in the conductor L 2 A 1 .

Conductor portions of the first coil L 1 and the second coil L 2 , the portions being close to each other in the stacking direction, contribute the most to the coupling. That is, portions of the conductors L 1 A 1 and L 2 A 1 , which face each other in the stacking direction over the entire circumference, contribute to the coupling between the first coil L 1 and the second coil L 2 . As described above, since current does not disperse in the conductors L 1 A 1 and L 2 A 1 due to the parallel connection portion, the coupling strength between the first coil L 1 and the second coil L 2 is high.

In the above manner, by providing the parallel connection portion apart from the conductor pattern of the other coil in the stacking direction, a decrease in the coupling strength caused by the provision of the parallel connection portion is significantly reduced or prevented.

In addition, by arranging the conductors L 1 A 1 and L 2 A 1 , which are connected to the terminals T 1 and T 2 around the center in the stacking direction, and by arranging the conductors L 1 C and L 2 C, which are connected to the ground terminal GND, in the upper portion and the lower portion in the stacking direction, such effects are produced that it is possible to provide a transformer in which the first coil L 1 and the second coil L 2 share a magnetic flux without a complex configuration and to adjust the capacitance element C 3 easily.

FIG. 7 is a Smith chart illustrating a phase-shift function of the transformer T. If a transformer has a transformer ratio of 1:1 and does not include a parasitic component (if it is an ideal transformer having a transformer ratio of 1:1), both the impedance of the transformer seen from the first port P 1 and the impedance of the transformer seen from the second port P 2 are plotted at the center of the Smith chart. Note that the phase difference between the port P 1 and the port P 2 is 180°, as indicated by the semicircle arrow in FIG. 7 .

FIG. 8 illustrates a frequency characteristic of a phase-shift amount of the transformer-type phase shifter 11 according to this preferred embodiment. In FIG. 8 , the horizontal axis represents the frequency, and the vertical axis represents the phase-shift amount. The phase-shift amount is in the range of ±180°. In this example, marker m 1 represents a phase-shift amount at a frequency of 1 GHz, and marker m 2 represents a phase-shift amount at a frequency of 1.9 GHz. In the representation of FIG. 8 , each of the phase-shift amounts is a value obtained by adding 180° to a value obtained by subtracting a read value from 180° on the upper side. That is, the phase-shift amount at 1 GHz is 180°+65°=245°, and the phase-shift amount at 1.9 GHz is 180°+115°=295°.

In this manner, even if the frequency almost doubles, the phase-shift amount becomes only about 1.2 times the original (changed only by about 20%), for example.

FIG. 9 illustrates a frequency characteristic of an insertion loss of the transformer-type phase shifter 11 according to this preferred embodiment. Insertion losses are about −0.4 dB at a frequency of 1 GHz and about −0.3 dB at a frequency of 1.9 GHz, whereby a low insertion-loss characteristic is obtained. The insertion loss at a frequency of 1.9 GHz is lower than that at a frequency of 1 GHz in this example because more signal components directly pass through the third capacitance element C 3 without passing through the transformer T.

Note that the first capacitance element C 1 and the second capacitance element C 2 are not limited to capacitance between wires of coils, and conductor patterns other than coils may be formed. In addition, a capacitor may be connected as an externally attached component. Furthermore, the third capacitance element C 3 is not limited to capacitance between coils, and conductor patterns other than coils may be provided. In addition, a capacitor may be connected as an externally attached component.

Second Preferred Embodiment

FIGS. 10A and 10B are block diagrams illustrating configurations of phase-shift circuits 30 A and 30 B according to a second preferred embodiment of the present invention. Each of the phase-shift circuits 30 A and 30 B is connected between a feeder circuit 9 and an antenna 1 . The phase-shift circuit 30 A includes a transformer-type phase shifter 10 and a phase-shift line 20 that is connected in series to the transformer-type phase shifter 10 . The phase-shift circuit 30 B includes the phase-shift line 20 and the transformer-type phase shifter 10 that is connected in series to the phase-shift line 20 . The transformer-type phase shifter 10 preferably has the same basic configuration as the transformer-type phase shifter 11 described in the first preferred embodiment. The phase-shift line 20 is a phase-shift line having a phase-shift amount of less than about 90°, for example.

Each of the phase-shift circuits 30 A and 30 B has a phase difference between ports P 1 and P 2 by a phase angle obtained by adding the phase-shift amount of the transformer-type phase shifter 10 and the phase-shift amount of the phase-shift line 20 .

If the phase-shift amount obtained by fine adjustment of the deviation in the impedance due to the parasitic inductance component of the transformer by adding a capacitance element connected in parallel and a capacitance element connected in series is denoted by α, the phase-shift amount of the transformer-type phase shifter 10 is about 180°+α. If the phase-shift amount of the phase-shift line 20 is represented as β, the phase-shift amount of each of the phase-shift circuits 30 A and 30 B is about 180°+α+β.

In the above manner, by adding the phase-shift line 20 , it is possible to shift the phase even greater than about 180°, thus enabling fine adjustment of the entire phase-shift amount by using the phase-shift amount α of the transformer-type phase shifter 10 .

Note that an antenna matching circuit may be provided between each of the phase-shift circuits 30 A and 30 B and the antenna 1 in FIGS. 10A and 10B .

FIGS. 11A, 11B, 11C, and 11D illustrate circuit configurations of four phase-shift circuits 31 , 32 , 33 , and 34 . In FIGS. 11A, 11B, 11C, and 11D , phase-shift lines 21 and 22 and the like are represented as lumped-constant circuits although the phase-shift lines 21 and 22 and the like are distributed-constant circuits. Each of the phase-shift lines includes inductors Ld 1 and Ld 2 that are connected in series to a transmission line and capacitors Cd 1 and Cd 2 that are connected in shunt to the ground.

In the example of FIG. 11A , the phase-shift circuit 31 includes the transformer-type phase shifter 11 and the phase-shift line 21 . The transformer-type phase shifter 11 is the same as that illustrated in FIG. 1 in the first preferred embodiment.

In the example of FIG. 11B , the phase-shift circuit 32 includes a transformer-type phase-shifter main portion 11 B and the phase-shift line 22 . The capacitance element C 2 included in the phase-shift line 22 and the transformer-type phase-shifter main portion 11 B define a transformer-type phase shifter.

In the example of FIG. 11C , the phase-shift circuit 33 includes a transformer-type phase-shifter main portion 11 C, the capacitance element C 2 , and the phase-shift line 21 . The transformer-type phase-shifter main portion 11 C and the capacitance element C 2 , which is an externally attached component, define a transformer-type phase shifter.

In the example of FIG. 11D , the phase-shift circuit illustrated in FIG. 11A includes a single component.

Note that in the phase-shift line 20 , the phase-shift amount may be set by determining the electrical length of the transmission line (for example, an approximately 50Ω line) or may be adjusted by adding a lumped-constant element such as an inductor or a capacitor or an LC circuit.

As described above, although the phase-shift amount of each of the phase-shift circuits 30 A and 30 B illustrated in FIGS. 10A and 10B is about 180°+α+β, if the transformer included in the transformer-type phase shifter includes many parasitic components (parallel inductance component and series inductance component) (if the coupling coefficient k is small), α can be increased. However, if impedance is to be matched by the impedance adjustment circuit formed of the first capacitance elements C 1 , C 2 , and C 3 and the like, it is difficult to match the impedance in a wide band (it is difficult to collect impedance loci obtained by a frequency sweep around the center of the Smith chart). Accordingly, if the used frequency band is wide, it is preferable that α be set to a small value and that a predetermined phase-shift amount be obtained by using β instead. Thus, the phase is able to be shifted while keeping the impedance matching. In contrast, if the used frequency band is narrow, α may be set to a large value, and β may be set to a small value instead. Thus, the change in the phase-shift amount depending on the frequency is able to be significantly reduced or prevented. For example, in the case of wide-band application, the coupling coefficient k of the transformer T is preferably in a range of about 0.5≤k<about 1.0; in the case of narrow-band application, the coupling coefficient k of the transformer T is preferably in a range of about 0.3≤k<about 1.0.

Third Preferred Embodiment

FIG. 12 is a circuit diagram of a transformer-type phase shifter 13 according to a third embodiment of the present invention. Although the first preferred embodiment has described a phase shifter including a transformer with an impedance conversion ratio of 1:1 in the examples illustrated in FIG. 1 and FIGS. 2A and 2B , the impedance conversion ratio may be 1:n (where n is a value other than 1). For example, if n<1 is satisfied, an antenna having impedance lower than the impedance of the feeder circuit is able to be matched with the impedance of the feeder circuit. Therefore, according to this preferred embodiment, it is possible to perform a predetermined phase shift and impedance matching.

Fourth Preferred Embodiment

FIG. 13 is a circuit diagram of a transformer-type phase shifter 14 according to a fourth preferred embodiment of the present invention. The transformer-type phase shifter 14 according to this preferred embodiment includes an autotransformer including the first coil L 1 and the second coil L 2 that are magnetically coupled to each other. The first capacitance element C 1 is connected between the first port P 1 and the ground, and the second capacitance element C 2 is connected between the second port P 2 and the ground. In addition, the third capacitance element C 3 is connected between the first port P 1 and the second port P 2 .

In the autotransformer such as the transformer in this preferred embodiment, since the coupling coefficient between the first coil L 1 and the second coil L 2 is less than 1, a parallel inductance component and a series inductance component are generated. In addition, the impedance matching is performed by using the capacitance elements C 1 , C 2 , and C 3 .

Fifth Preferred Embodiment

A fifth preferred embodiment of the present invention will describe a communication terminal apparatus. FIG. 14 is a block diagram of a communication terminal apparatus 200 according to the fifth preferred embodiment. The communication terminal apparatus 200 according to this preferred embodiment includes the antenna 1 , an antenna matching circuit 40 , a phase-shift circuit 30 , a communication circuit 41 , a baseband circuit 42 , an application processor 43 , and an input/output circuit 44 . The communication circuit 41 includes a transmission circuit and a reception circuit for a low band (for example, about 700 MHz to about 1.0 GHz) and a high band (for example, about 1.4 GHz to about 2.7 GHz) and an antenna duplexer. The antenna 1 is a monopole antenna or an inverse F antenna compatible with the low band and the high band.

The above-described components are stored preferably in a single housing. For example, the antenna matching circuit 40 , the phase-shift circuit 30 , the communication circuit 41 , the baseband circuit 42 , and the application processor 43 are mounted on a printed wiring board, and the printed wiring board is stored in the housing. The input/output circuit 44 is incorporated in the housing as a display and touch panel. The antenna 1 is mounted on the printed wiring board or arranged on an inner wall of the housing or inside the housing.

The communication terminal apparatus having the above-described configuration and including an antenna that provides matching in a wide band is obtained.

Note that, instead of being inserted in a multi-band communication signal path, the phase-shift circuit 30 may, for example, be used in one of a low-band (for example, about 700 MHz to about 1.0 GHz) line and a high-band (for example, about 1.4 GHz to about 2.7 GHz) line as illustrated in FIG. 14 .

Note that the above-described preferred embodiments have illustrated examples in which the impedance adjustment circuit that adjusts the impedance of the transformer preferably includes three capacitance elements C 1 , C 2 , and C 3 . The impedance adjustment circuit is a circuit that corrects or actively modifies the displacement in the impedance due to the parallel inductance component and the series inductance component, which are parasitic components of the transformer. Accordingly, the impedance adjustment circuit does not necessarily include or require three capacitance elements. The impedance of the transformer T may be finely adjusted by connecting the transformer to a predetermined reactance element in parallel or in series.

Lastly, the above-described preferred embodiments are illustrative in all points and are not limiting. A person skilled in the art may make changes or modifications as appropriate. For example, elements or features of the configurations described in different preferred embodiments may be replaced or combined with each other. The scope of the present invention is to be defined by the scope of the claims, not the above-described preferred embodiments. In addition, the scope of the present invention is to include equivalents to the scope of the claims and all changes within the scope of the claims.

While preferred embodiments of the present invention have been described above, it is to be understood that variations and modifications will be apparent to those skilled in the art without departing from the scope and spirit of the present invention. The scope of the present invention, therefore, is to be determined solely by the following claims.