DC/DC converter, driving method thereof, and power supply using the same

This application claims priority under 35 U.S.C. § 119 to Korean Patent Application No. 10-2015-0058268, filed on Apr. 24, 2015, the entire contents of which are incorporated herein by reference in their entirety.

1. Technical Field

Inventive concepts relate to a DC/DC converter and a driving method thereof, and more particularly, to a DC/DC converter, a driving method thereof, and a power supply using the same.

2. Description of Related Art

Generally, switching regulators are apparatuses that convert power, which is input from an external source, to direct current (DC) power necessary for a system by using a plurality of power switches. Examples of switching regulators include DC/DC converters and/or the like for boosting or dropping input DC power. A plurality of power switches configuring a DC/DC converter may stably operate for enhancing performance of a system.

The inventive concepts provide a DC/DC converter which efficiently manages power of a driver driving a plurality of power switches, thereby reducing damage to the power switches.

The inventive concepts provide a power supply which efficiently manages power of a driver driving a plurality of power switches included in a DC/DC converter, thereby reducing damage to the power switches.

The inventive concepts provide a method of driving a DC/DC converter for stably driving a plurality of power switches included in the DC/DC converter.

According to an aspect of the inventive concepts, there is provided a DC/DC converter including a plurality of power switches connected as a serial string between an input terminal and a ground terminal, a first capacitor connected to at least two power switches of the serial string in parallel, an inductor connected between an intermediate node of the serial string and an output terminal, a second capacitor connected between the output terminal and the ground terminal, and a plurality of drivers configured to generate a switching control signal for each of the plurality of power switches, wherein a voltage having no correlation with a voltage of the input terminal is supplied to a power terminal of each of the plurality of drivers.

The first capacitor may be connected between a first node and a second node of the serial string, the first node may be located between the intermediate node of the serial string and the input terminal, and the second node may be located between the intermediate node of the serial string and the ground terminal.

The plurality of power switches may include a plurality of high-side p-type metal-oxide-semiconductor (PMOS) transistors connected between the input terminal and the intermediate node of the serial string, and a plurality of low-side n-type metal-oxide-semiconductor (NMOS) transistors connected between the intermediate node of the serial string and the ground terminal.

The plurality of drivers may include a plurality of high-side drivers configured to respectively generate switching control signals for the plurality of high-side PMOS switches, and a plurality of low-side drivers configured to respectively generate switching control signals for the plurality of low-side NMOS transistors, and a negative source voltage supplied to at least one driver of the plurality of high-side drivers and a positive source voltage supplied to at least one driver of the plurality of low-side drivers may be constant voltages that are different from each other.

The negative source voltage supplied to at least one driver of the plurality of high-side drivers may be determined as a constant voltage that is lower than a voltage obtained by subtracting a source-gate voltage, which is used to turn on at least one of the plurality of high-side PMOS transistors, from a voltage of the input terminal.

The positive source voltage supplied to at least one driver of the plurality of low-side drivers may be determined as a constant voltage which is higher than a voltage obtained by adding a gate-source voltage, which is used to turn on at least one of the plurality of low-side NMOS transistors, to a voltage of the ground terminal.

The plurality of power switches may include a first PMOS transistor, a second PMOS transistor, a first NMOS transistor, and a second NMOS transistor, and a source terminal of the first PMOS transistor may be connected to the input terminal, a drain terminal of the first PMOS transistor is connected to a first node, a source terminal of the second PMOS transistor is connected to the first node, a drain terminal of the second PMOS transistor is connected to the intermediate node, a drain terminal of the first NMOS transistor is connected to the intermediate node, a source terminal of the first NMOS transistor is connected to a second node, a drain terminal of the second NMOS transistor is connected to the second node, and a source terminal of the second NMOS transistor is connected to the ground terminal. The first capacitor may be connected between the first node and the second node, and the plurality of drivers may include first to fourth drivers configured to respectively generate gate control signals for the first and second PMOS transistors and the first and second NMOS transistors.

A positive source voltage terminal of the first driver may be connected to the input terminal, a first constant voltage may be supplied to a negative source voltage terminal of the first driver, and the first constant voltage may be determined as a constant voltage that is lower than a voltage obtained by subtracting a source-gate voltage, which is used to turn on the first PMOS transistor, from a voltage of the input terminal.

A positive source voltage terminal of the second driver may be connected to the first node, a third capacitor may be connected between the positive source voltage terminal of the second driver and the first node, and a first diode may be connected between the positive source voltage terminal of the second driver and a negative source voltage terminal of the first driver.

A fourth capacitor may be connected between a positive source voltage terminal of the third driver and the second node, a second diode may be connected between the positive source voltage terminal of the third driver and a positive source voltage terminal of the fourth driver, and a negative source voltage terminal of the third driver may be connected to the second node.

A second constant voltage may be supplied to a positive source voltage terminal of the fourth driver, and the second constant voltage may be determined as a constant voltage that is higher than a voltage obtained by adding a gate-source voltage, which is used to turn on the second NMOS transistor, from a voltage of the ground terminal.

According to another aspect of the inventive concepts, there is provided a power supply including a feedback circuit configured to generate a first error signal and a second error signal, based on a difference between at least two feedback signals and at least one reference source, a controller configured to generate a plurality of control signals for making one of an output voltage, an output current, and an input current converge on a target value, based on the first error signal and the second error signal, and a DC/DC converter configured to change a path of a current, flowing to at least one inductor and capacitor which are included in a circuit between an input terminal and an output terminal, to generate the output voltage and the at least two feedback signals, based on the plurality of control signals, the DC/DC converter including a plurality of power switches connected as a serial string between the input terminal and a ground terminal and a plurality of drivers configured to generate a switching control signal for each of the plurality of power switches, wherein a voltage having no correlation with a voltage of the input terminal is supplied to a power terminal of each of the plurality of drivers.

The feedback circuit may include a 1Ath amplifier configured to generate the first error signal corresponding to a difference between the reference source and the output voltage detected from the output terminal, a second amplifier configured to generate a first signal corresponding to a voltage difference between a first node and a second node included in the serial string, and a 3Ath amplifier configured to generate the second error signal corresponding to a voltage difference between the first signal and the input voltage applied to the input terminal.

The feedback circuit may include a 1Bth amplifier configured to generate the first error signal corresponding to a difference between the reference source and the output current detected between the output terminal and a load circuit, a second amplifier configured to generate a first signal corresponding to a voltage difference between a first node and a second node included in the serial string, and a third amplifier configured to generate the second error signal corresponding to a voltage difference between the first signal and the input voltage applied to the input terminal.

The feedback circuit may include a 1Cth amplifier configured to generate the first error signal corresponding to a difference between the reference source and the input current detected from the input terminal, a second amplifier configured to generate a first signal corresponding to a voltage difference between a first node and a second node included in the serial string, and a third amplifier configured to generate the second error signal corresponding to a voltage difference between the first signal and the input voltage applied to the input terminal.

The controller may include an adder configured to generate a first signal corresponding to a sum of the first error signal and the second error signal, a subtractor configured to generate a second signal corresponding to a difference between the first error signal and the second error signal, a first comparator configured to compare the first signal with a first triangular wave signal to generate a first control signal and a fourth control signal, and a second comparator configured to compare the second signal with a second triangular wave signal to generate a second control signal and a third control signal, the first triangular wave signal having a 180-degree phase difference with the second triangular wave signal.

The power supply may further include a plurality of level shifters configured to respectively shift levels of the plurality of control signals, wherein output signals of the plurality of level shifters may be respectively supplied to input terminals of the plurality of drivers, and a source voltage of each of the plurality of level shifters may be applied as a source voltage that is the same as a source voltage of a corresponding driver.

The power supply may further include a regulator circuit configured to generate a source voltage supplied to at least one driver of the plurality of drivers, wherein the regulator circuit may include a first voltage regulator configured to generate a first source voltage through a first terminal a second voltage regulator configured to generate a second source voltage through a second terminal, a third capacitor connected between the first terminal and the input terminal, a first diode connected between the first terminal and a third node, a fourth capacitor connected between the third node and a first node included in the serial string, a fifth capacitor connected between the second terminal and the ground terminal, a second diode connected between the second terminal and a fourth node, and a sixth capacitor connected between the fourth node and a second node included in the serial string. The first terminal may be connected to a negative source voltage terminal of a first driver, the second terminal may be connected to a positive source voltage terminal of a fourth driver, the third node may be connected to a negative source voltage terminal of a second driver, and the fourth node may be connected to a positive source voltage terminal of a third driver.

The plurality of power switches may include a first PMOS transistor, a second PMOS transistor, a first NMOS transistor, and a second NMOS transistor, and a source terminal of the first PMOS transistor may be connected to the input terminal, a drain terminal of the first PMOS transistor is connected to the first node, a source terminal of the second PMOS transistor is connected to the first node, a drain terminal of the second PMOS transistor is connected to an intermediate node, a drain terminal of the first NMOS transistor is connected to the intermediate node, a source terminal of the first NMOS transistor is connected to a second node, a drain terminal of the second NMOS transistor is connected to the second node, and a source terminal of the second NMOS transistor is connected to the ground terminal. A positive source voltage terminal of the first driver may be connected to the input terminal, a positive source voltage terminal of the second driver is connected to the first node, a negative source voltage terminal of the third driver is connected to the second node, and a negative source voltage terminal of the fourth driver is connected to the ground terminal, and output terminals of the first to fourth drivers may be respectively connected to gate terminals of the first and second PMOS transistors and the first and second NMOS transistors.

According to another aspect of the inventive concepts, there is provided a method of driving a DC/DC converter, including a plurality of power switches that are connected as a serial string between an input terminal and a ground terminal and include a first PMOS transistor, a second PMOS transistor, a first NMOS transistor, and a second NMOS transistor, including: supplying a positive source voltage of a first driver, which is used to drive the first PMOS transistor, from a source terminal of the first PMOS transistor and supplying a first constant voltage as a negative source voltage of the first driver, supplying a positive source voltage of a second driver, which is used to drive the second PMOS transistor, from a source terminal of the second PMOS transistor and supplying a negative source voltage of the second driver from a node located between a first diode and a first capacitor in a circuit where the first diode and the first capacitor are serially connected between a negative source voltage terminal of the first driver and a source terminal of the second PMOS transistor, supplying a second constant voltage as a positive source voltage of a fourth driver for driving the second NMOS transistor, and supplying a ground voltage as a negative source voltage of the fourth driver, and supplying a positive source voltage of a third driver, which is used to drive the first NMOS transistor, from a node located between a second diode and a second capacitor in a circuit where the second diode and the second capacitor are serially connected between a positive source voltage terminal of the fourth driver and a drain terminal of the first NMOS transistor, wherein output terminals of the first to fourth drivers are respectively connected to gate terminals of the first PMOS transistor, the second PMOS transistor, the first NMOS transistor, and the second NMOS transistor, and a signal for controlling switching of the first PMOS transistor, the second PMOS transistor, the first NMOS transistor, and the second NMOS transistor is applied to an input terminal of each of the first to fourth drivers.

Exemplary embodiments of the inventive concepts will be more clearly understood from the following detailed description taken in conjunction with the accompanying drawings in which:

FIG. 1 is a diagram illustrating a configuration of a DC/DC converter according to an exemplary embodiment;

FIG. 2 is a table showing switching states of a plurality of power switches configuring the DC/DC converter illustrated in FIG. 1 and a voltage state of a node ND 3 ;

FIG. 3 is a diagram illustrating a configuration of a DC/DC converter according to another exemplary embodiment;

FIG. 4 is a diagram showing waveforms of main signals of the DC/DC converter illustrated in FIG. 3 ;

FIG. 5 is a diagram illustrating a configuration of a DC/DC converter according to another exemplary embodiment;

FIG. 6 exemplarily illustrates a diagram of a configuration in which power switches of FIG. 5 are implemented using transistors;

FIG. 7 is a diagram illustrating a connection relationship between a plurality of level shifters added into the DC/DC converter of FIG. 5 ;

FIG. 8 is a diagram illustrating input/output terminals of a high-side voltage regulator circuit which generates source voltages of high-side drivers illustrated in FIG. 6 ;

FIG. 9 is a diagram illustrating input/output terminals of a low-side voltage regulator circuit which generates source voltages of low-side drivers illustrated in FIG. 6 ;

FIG. 10 is a diagram illustrating a detailed configuration of the high-side voltage regulator circuit illustrated in FIG. 8 ;

FIG. 11 is a diagram illustrating a detailed configuration of the low-side voltage regulator circuit illustrated in FIG. 9 ;

FIG. 12 is a block diagram illustrating a configuration of a power supply according to an exemplary embodiment;

FIG. 13 is a diagram illustrating an example of a detailed circuit configuration of the power supply illustrated in FIG. 12 ;

FIG. 14 is a diagram illustrating another example of a detailed circuit configuration of the power supply illustrated in FIG. 12 ;

FIG. 15 is a diagram illustrating another example of a detailed circuit configuration of the power supply illustrated in FIG. 12 ;

FIG. 16 is a diagram illustrating another example of a detailed circuit configuration of the power supply illustrated in FIG. 12 ;

FIG. 17 shows an example of waveforms of main signals of a DC/DC converter and a power supply according to an exemplary embodiment;

FIG. 18 shows another example of waveforms of main signals of a DC/DC converter and a power supply according to an exemplary embodiment;

FIG. 19 illustrates an example of a configuration of an electronic device to which a power supply according to an exemplary embodiment is applied; and

FIG. 20 illustrates another example of a configuration of an electronic device to which a power supply according to an exemplary embodiment is applied.

The inventive concepts will now be described more fully hereinafter with reference to the accompanying drawings, in which embodiments of the invention are shown. The embodiments may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the inventive concepts to those skilled in the art. In the drawings, the size and relative sizes of layers and regions may be exaggerated for clarity. Like numbers refer to like elements throughout.

It will be understood that when an element is referred to as being “connected” or “coupled” to another element, it can be directly connected or coupled to the other element or intervening elements may be present. In contrast, when an element is referred to as being “directly connected” or “directly coupled” to another element, there are no intervening elements present. As used herein, the term “and/or” includes any and all combinations of one or more of the associated listed items and may be abbreviated as “/”.

It will be understood that, although the terms first, second, etc, may be used herein to describe various elements, these elements should not be limited by these terms. These terms are only used to distinguish one element from another. For example, a first signal could be termed a second signal, and, similarly, a second signal could be termed a first signal without departing from the teachings of the disclosure.

The terms used in this application, only certain embodiments have been used to describe, is not intended to limit the present embodiments. In the following description, the technical terms are used only to explain specific exemplary embodiments while not limiting the exemplary embodiments. The terms of a singular form may include plural forms unless referred to the contrary. The meaning of “include,” “comprise,” “including,” or “comprising,” specifies a property, a region, a fixed number, a step, a process, an element and/or a component but does not exclude other properties, regions, fixed numbers, steps, processes, elements and/or components.

Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. It will be further understood that terms, such as those defined in commonly used dictionaries, should be interpreted as having a meaning that is consistent with their meaning in the context of the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein.

FIG. 1 is a diagram illustrating a configuration of a DC/DC converter 100 according to an exemplary embodiment.

As illustrated in FIG. 1 , the DC/DC converter 100 may include four n-type metal-oxide-semiconductor (NMOS) transistors M 11 to M 14 that operate as switches, two capacitors C 11 and C 12 , an inductor L 11 , and a driver circuit 10 that drives the four NMOS transistors M 11 to M 14 .

A circuit configuration of the DC/DC converter 100 is described as follows.

The four NMOS transistors M 11 to M 14 may be connected as a serial string between an input terminal IN 1 and a ground terminal GND. The capacitor C 11 may be connected between a node ND 1 and a node ND 2 of the serial string. A source terminal of the NMOS transistor M 11 and a drain terminal of the NMOS transistor M 12 may be connected to the node ND 1 , and a source terminal of the NMOS transistor M 13 and a drain terminal of the NMOS transistor M 14 may be connected to the node ND 2 . The inductor L 11 may be connected between an output terminal OUT 1 and a node ND 3 corresponding to an intermediate node of the serial string, and the capacitor C 12 may be connected between the output terminal OUT 1 and the ground terminal GND. An input voltage VIN may be applied to the input terminal IN 1 .

A gate driver may be connected to a gate terminal of each of the four NMOS transistors M 11 to M 14 . That is, a gate driver GD 11 may be connected to the gate terminal of the NMOS transistor M 11 , a gate driver GD 12 may be connected to the gate terminal of the NMOS transistor M 12 , a gate driver GD 13 may be connected to the gate terminal of the NMOS transistor M 13 , and a gate driver GD 14 may be connected to the gate terminal of the NMOS transistor M 14 .

Logic states of control signals T 1 , T 2 , B 1 and B 2 respectively applied to the gate drivers GD 11 to GD 14 may be determined in order for states S 1 to S 4 of FIG. 2 to be sequentially provided.

FIG. 2 is a table showing switching states of a plurality of power switches configuring the DC/DC converter 100 illustrated in FIG. 1 and a voltage state of the node ND 3 .

Referring to FIG. 2 , in the state S 1 , the NMOS transistors M 11 and M 12 may be turned on, and the NMOS transistors M 13 and M 14 may be turned off. Therefore, a voltage Vx of the node ND 3 may be VIN.

In the state S 2 , the NMOS transistors M 12 and M 14 may be turned on, and the NMOS transistors M 11 and M 13 may be turned off. In the state S 2 , the voltage Vx of the node ND 3 may be VIN/2, based on a voltage “VIN/2” charged into the capacitor C 11 that operates as a flying capacitor.

In the state S 3 , the NMOS transistors M 11 and M 13 may be turned on, and the NMOS transistors M 12 and M 14 may be turned off. In the state S 3 , the voltage Vx of the node ND 3 may be VIN/2, based on a voltage “VIN/2” charged into the capacitor C 11 that operates as the flying capacitor.

In the state S 4 , the NMOS transistors M 13 and M 14 may be turned on, and the NMOS transistors M 11 and M 12 may be turned off. Therefore, a voltage Vx of the node ND 3 may be 0 V.

A voltage VOUT of the output terminal OUT 1 may be controlled by switching operations of the four NMOS transistors M 11 to M 14 .

In the driver circuit 10 , the gate driver GD 11 may output a higher voltage than the input voltage VIN in the states S 1 and S 3 where the NMOS transistor M 11 is turned on. Therefore, a higher voltage than the input voltage VIN may be supplied to the driver circuit 10 . Therefore, circuit elements of the driver circuit 10 may be implemented to have a standard of a withstand voltage that is higher than the input voltage VIN.

FIG. 3 is a diagram illustrating a configuration of a DC/DC converter 200 according to another exemplary embodiment.

As illustrated in FIG. 3 , the DC/DC converter 200 may include four transistors M 21 to M 24 , two capacitors C 21 and C 22 , an inductor L 21 , and a plurality of gate drivers GD 21 to GD 23 .

Two p-type metal-oxide-semiconductor (PMOS) transistors M 21 and M 22 and two NMOS transistors M 23 and M 24 may be connected as a serial string between an input terminal IN 2 and a ground terminal GND. The capacitor C 21 may be connected between a node ND 21 and a node ND 22 of the serial string. A drain terminal of the PMOS transistor M 21 and a source terminal of the PMOS transistor M 22 may be connected to the node ND 21 , and a source terminal of the NMOS transistor M 23 and a drain terminal of the NMOS transistor M 24 may be connected to the node ND 22 . The inductor L 21 may be connected between an output terminal OUT 2 and a node ND 23 corresponding to an intermediate node of the serial string, and the capacitor C 22 may be connected between the output terminal OUT 2 and the ground terminal GND. An input voltage VIN may be applied to the input terminal IN 2 .

The gate driver GD 21 is a driver for driving the PMOS transistor M 21 and may be connected to a gate terminal of the PMOS transistors M 21 . The gate driver GD 22 is a driver for driving the PMOS transistor M 22 and the NMOS transistor M 23 and may be connected to a gate terminal of the PMOS transistors M 22 and a gate terminal of the NMOS transistor M 23 . The gate driver GD 23 is a driver for driving the NMOS transistor M 24 and may be connected to a gate terminal of the NMOS transistors M 24 .

Waveforms of control signals T 21 , MID 21 and B 21 respectively applied to the gate drivers GD 21 to GD 23 and a voltage V L of the node ND 23 are shown in FIG. 4 .

The gate driver GD 21 may operate at voltages of a range from VIN/2 to VIN, and the gate driver GD 23 may operate at voltages of a range from 0 to VIN/2. Also, the gate driver GD 22 may operate with a voltage charged into the capacitor C 21 that operates as a flying capacitor. That is, the gate driver GD 22 may operate at 0 to VIN/2 or VIN/2 to VIN.

Therefore, the voltage charged into the capacitor C 21 may be always maintained as VIN/2 in order for the gate driver GD 22 to stably operate. Also, a voltage regulator for generating VIN/2 may be used.

When an input voltage VIN that is two or more times higher than a gate-source rated voltage of each of the PMOS transistors M 21 and M 22 which are power switches is applied, the PMOS transistors M 21 and M 22 are damaged. For example, when a gate-source voltage “VIN−VIN/2” in a section where the PMOS transistor M 21 is turned on is applied as two or more times higher than a rated voltage, the PMOS transistor M 21 is damaged.

As described above, in another exemplary embodiment, a new DC/DC converter 300 for overcoming limitations of the DC/DC converter 100 of FIG. 1 and the DC/DC converter 200 of FIG. 3 is proposed.

Hereinafter, the new DC/DC converter 300 proposed according to another exemplary embodiment will be described in detail.

FIG. 5 is a diagram illustrating a configuration of the DC/DC converter 300 according to another exemplary embodiment.

As illustrated in FIG. 5 , the DC/DC converter 300 may include a plurality of power switches SW 1 to SW 4 , a plurality of drivers DR 1 to DR 4 , a plurality of capacitors C 1 and C 2 , and an inductor L 1 .

The plurality of power switches SW 1 to SW 4 may be connected as a serial string between an input terminal IN and a ground terminal GND. The capacitor C 1 may be connected between a node Nd 1 and a node Nd 2 of the serial string. In the serial string, the node Nd 1 may be disposed between the power switch SW 1 and the power switch SW 2 , and the node Nd 2 may be disposed between the power switch SW 3 and the power switch SW 4 . The inductor L 1 may be connected between an output terminal OUT and a node Nd 3 corresponding to an intermediate node of the serial string, and the capacitor C 2 may be connected between the output terminal OUT and the ground terminal GND. The input voltage VIN may be applied to the input terminal TN. In the serial string, the node Nd 3 may be disposed between the power switch SW 2 and the power switch SW 3 .

The plurality of drivers DR 1 to DR 4 may be respectively connected to the plurality of power switches SW 1 to SW 4 . In detail, an output signal of the driver DR 1 may be applied to the power switch SW 1 , an output signal of the driver DR 2 may be applied to the power switch SW 2 , an output signal of the driver DR 3 may be applied to the power switch SW 3 , and an output signal of the driver DR 4 may be applied to the power switch SW 4 . The plurality of power switches SW 1 to SW 4 may be turned on or off based on voltage levels of respective signals applied from the plurality of drivers DR 1 to DR 4 .

The drivers DR 1 and DR 2 may correspond to high-side drivers that respectively generate switching control signals for high-side power switches SW 1 and SW 2 connected between the input terminal IN and the node Nd 3 of the serial string. Also, the drivers DR 3 and DR 4 may correspond to low-side drivers that respectively generate switching control signals for low-side power switches SW 3 and SW 4 connected between the node Nd 3 of the serial string and the ground terminal GND.

A negative source voltage supplied to at least one of the high-side drivers DR 1 and DR 2 and a positive source voltage supplied to at least one of the low-side drivers DR 3 and DR 4 may be set as different constant voltages.

For example, a negative source voltage of the high-side driver DR 1 may be set as a constant voltage VSSH, and a positive source voltage supplied to the low-side driver DR 4 may be set as a constant voltage PVDD. The constant voltages VSSH and PVDD may correspond to different constant voltages. The constant voltage VSSH may be determined as having a voltage level that satisfies a condition for turning on the power switch SW 3 , and the constant voltage PVDD may be determined as having a voltage level that satisfies a condition for turning on the power switch SW 4 .

For example, the constant voltages VSSH and PVDD may be generated by different voltage regulators. Also, a positive source voltage of the high-side driver DR 1 may be set as the input voltage VIN, and a negative source voltage supplied to the low-side driver DR 4 may be set as a ground voltage.

A positive source voltage terminal of the high-side driver DR 2 may be connected to the node Nd 1 , and a negative source voltage VSS_IM may be generated from the constant voltage VSSH and a voltage VDDM of the node Nd 1 . A negative source voltage terminal of the low-side driver DR 3 may be connected to the node Nd 2 , and a positive source voltage VDD_IM may be generated from the constant voltage PVDD and a voltage VSSM of the node Nd 2 . A detailed circuit for generating the negative source voltage VSS_IM of the high-side driver DR 2 and the positive source voltage VDD_IM of the low-side driver DR 3 will be described below in detail.

A plurality of control signals P 1 ′ to P 4 ′ necessary to switching-control the plurality of power switches SW 1 to SW 4 may be respectively applied to the plurality of drivers DR 1 to DR 4 .

FIG. 6 exemplarily illustrates a diagram of a configuration in which the power switches SW 1 to SW 4 of FIG. 5 are implemented using transistors.

Referring to FIG. 6 , a DC/DC converter 300 ′ may include a plurality of transistors M 1 to M 4 that are connected as a serial string between an input terminal IN and a ground terminal GND. For example, high-side transistors M 1 and M 2 may be implemented as PMOS transistors between the input terminal IN and an intermediate node Nd 3 of the serial string, and low-side transistors M 3 and M 4 which are connected between the intermediate node Nd 3 of the serial string and the ground terminal GND may be implemented as NMOS transistors.

The drivers DR 1 and DR 2 may respectively generate gate control signals for the high-side PMOS transistors M 1 and M 2 . The drivers DR 3 and DR 4 may respectively generate gate control signals for the low-side NMOS transistors M 3 and M 4 .

FIG. 7 is a diagram illustrating an interconnection between a plurality of level shifters, control signals and voltage shifted control signals added into the DC/DC converter of FIG. 5 .

Referring to FIG. 7 , the DC/DC converter 300 of FIG. 5 may further include a plurality of level shifters LS 1 to LS 4 . The level shifters LS 1 to LS 4 may respectively adjust voltage levels of control signals P 1 to P 4 output from a controller (see descriptions of FIGS. 12 to 16 ). For example, operating voltage ranges of the level shifters LS 1 to LS 4 may respectively match those of drivers DR 1 to DR 4 corresponding thereto. For example, a positive source voltage and a negative source voltage of the level shifter LS 1 may match those of the driver DR 1 . In this way, positive source voltages and negative source voltages of the level shifters LS 2 to LS 4 may respectively match those of the drivers DR 2 to DR 4 . The level shifters generate voltage shifted control signals P 1 ′ to P 4 ′ by shifting voltage levels of the control signal P 1 to P 4 .

FIG. 8 is a diagram illustrating input/output terminals of a high-side voltage regulator circuit 310 which generates source voltages of the high-side drivers DR 1 and DR 2 illustrated in FIG. 6 .

The high-side voltage regulator circuit 310 may generate the constant voltage VSSH, based on the input voltage VIN and a first reference voltage VREF 1 . Also, the high-side voltage regulator circuit 310 may generate the negative source voltage VSS_IM, based on the constant voltage VSSH and the voltage VDDM of the node Nd 1 . For example, the negative source voltage VSS_IM may be generated from the constant voltage VSSH and the voltage VDDM of the node Nd 1 by using a diode and a capacitor circuit. For example, referring again to FIG. 6 , the high-side voltage regulator circuit 310 may be implemented so that the negative source voltage VSS_IM is discharged to a voltage “VSSH+VD” when the transistor M 1 is turned on. Here, VD is a diode forward voltage.

FIG. 9 is a diagram illustrating input/output terminals of a low-side voltage regulator circuit 320 which generates source voltages of low-side drivers DR 3 and DR 4 illustrated in FIG. 6 .

The low-side voltage regulator circuit 320 may generate the constant voltage PVDD, based on the input voltage VIN and a second reference voltage VREF 2 . Also, the low-side voltage regulator circuit 320 may generate the positive source voltage VDD_IM, based on the constant voltage PVDD and the voltage VSSM of the node Nd 2 . For example, the positive source voltage VDD_IM may be generated from the constant voltage PVDD and the voltage VSSM of the node Nd 2 by using a diode and a capacitor circuit. For example, referring again to FIG. 6 , the low-side voltage regulator circuit 320 may be implemented so that the positive source voltage VDD_IM is discharged to a voltage “PVDD−VD” when the transistor M 4 is turned on. Here, VD is a diode forward voltage.

FIG. 10 is a diagram illustrating a detailed configuration of the high-side voltage regulator circuit 310 illustrated in FIG. 8 .

Referring to FIG. 10 , the high-side voltage regulator circuit 310 may include a first voltage regulator “Vreg_H” 310 - 1 , a plurality of capacitors C 3 and C 4 , and a diode D 1 .

The first voltage regulator 310 - 1 may generate the constant voltage VSSH from the input voltage VIN, based on the first reference voltage VREF 1 . Various types of constant voltage generation circuits may be applied to the first voltage regulator 310 - 1 . For example, a low-dropout (LDO) regulator may be applied. The constant voltage VSSH may be determined as having a voltage value that is lower by a certain margin voltage than a voltage value obtained by subtracting a source-gate voltage, which is necessary to turn on the PMOS transistor M 1 , from the input voltage VIN.

The capacitor C 3 may be connected between an input voltage VIN terminal and a node Nd 4 from which the constant voltage VSSH is output. A diode D 1 may be connected between the node Nd 4 and a node Nd 5 from which the negative source voltage VSS_IM of the driver DR 2 is output. An anode of the diode D 1 may be connected to the node Nd 5 , and a cathode of the diode D 1 may be connected to the node Nd 4 . Also, the capacitor C 4 may be connected between the node Nd 5 and the node Nd 1 of FIG. 6 from which the voltage VDDM is generated.

Referring to FIGS. 6 and 10 , when the transistor M 1 is turned on by using the negative source voltage VSS_IM of the driver DR 2 , the voltage VDDM of the node Nd 1 may become equal to the input voltage VIN. Also, when the voltage VSS_IM is higher than the constant voltage VSSH, the diode D 1 may be turned on. Therefore, the voltage VSS_IM may be discharged to “VSSH+VD”. Here, VD is a diode forward voltage. The voltage VSS_IM may be used as the negative source voltage of the driver DR 2 .

FIG. 11 is a diagram illustrating a detailed configuration of the low-side voltage regulator circuit 320 illustrated in FIG. 9 .

Referring to FIG. 11 , the low-side voltage regulator circuit 320 may include a second voltage regulator “Vreg_L” 320 - 1 , a plurality of capacitors C 5 and C 6 , and a diode D 2 .

The second voltage regulator 320 - 1 may generate the constant voltage PVDD from the input voltage VIN, based on the second reference voltage VREF 2 . Various types of constant voltage generation circuits may be applied to the second voltage regulator 320 - 1 . For example, an LDO regulator may be applied. The constant voltage PVDD may be determined as having a voltage value that is higher by a certain margin voltage than a voltage value obtained by adding a source-gate voltage, which is necessary to turn on the NMOS transistor M 4 , from the ground voltage.

The capacitor C 5 may be connected between a ground terminal GND and a node Nd 6 from which the constant voltage PVDD is output. A diode D 2 may be connected between the node Nd 6 and a node Nd 7 from which the positive source voltage VDD_IM of the driver DR 3 is output. An anode of the diode D 2 may be connected to the node Nd 6 , and a cathode of the diode D 2 may be connected to the node Nd 7 . Also, the capacitor C 6 may be connected between the node Nd 7 and the node Nd 2 of FIG. 6 from which the voltage VSSM is generated.

Referring to FIGS. 6 and 11 , when the transistor M 4 is turned on by using the positive source voltage VDD_IM of the driver DR 3 , the voltage VSSM of the node Nd 2 may become equal to the ground voltage. Also, when the voltage VDD_IM is lower than the constant voltage PVDD, the diode D 2 may be turned on. Therefore, the voltage VDD_IM may be discharged to “PVDD−VD”. Here, VD is a diode forward voltage. The voltage VDD_IM may be used as the positive source voltage of the driver DR 3 .

FIG. 12 is a block diagram illustrating a configuration of a power supply 1000 according to an exemplary embodiment.

Referring to FIG. 12 , the power supply 1000 may include a DC/DC converter 1100 , a regulator circuit 1200 , a feedback circuit 1300 , and a controller 1400 .

The DC/DC converter 1100 may generate an output voltage VOUT, which is lower by a certain voltage than an input voltage VIN, from a plurality of control signals P 1 to P 4 supplied from the controller 1400 . The DC/DC converter 1100 may change a path of a current, flowing to at least one inductor and capacitor which are included in a circuit between an input terminal IN and an output terminal OUT, to generate an output voltage and at least two feedback signals fb 1 to fbi (where I is an integer equal to or more than two), based on the control signals P 1 to P 4 .

For example, the DC/DC converter 1100 may use the DC/DC converter 300 of FIG. 5 or the DC/DC converter 300 ′ of FIG. 6 .

Some source voltages necessary for a plurality of drivers included in the DC/DC converter 1100 may be supplied from the regulator circuit 1200 . For example, the regulator circuit 1200 may include the high-side voltage regulator circuit 310 of FIG. 10 and the low-side voltage regulator circuit 320 of FIG. 11 .

The feedback circuit 1300 may receive at least two feedback signals fb 1 to fbi from the DC/DC converter 1100 and generate a first error signal Verr 1 and a second error signal Verr 2 , based on a difference between the at least two feedback signals fb 1 to fbi and a reference source. The first error signal Verr 1 and the second error signal Verr 2 generated by the feedback circuit 1300 may be output to the controller 1400 .

The controller 1400 may generate the control signals P 1 to P 4 for allowing one of an output voltage, an output current, and an input current to converge on a target value, based on the first error signal Verr 1 and the second error signal Verr 2 . The control signals P 1 to P 4 may be output to the DC/DC converter 1100 .

FIG. 13 is a diagram illustrating an example 1000 A of a detailed circuit configuration of the power supply 1000 illustrated in FIG. 12 .

Referring to FIG. 13 , a power supply 1000 A may include a DC/DC converter 1100 A, a feedback circuit 1300 A, a controller 1400 A, a high-side voltage regulator circuit 310 , and a low-side voltage regulator circuit 320 . For reference, a load circuit 20 which is connected between an output terminal and a ground terminal GND of the DC/DC converter 1100 A is illustrated. FIG. 13 illustrates a circuit that controls an output voltage VOUT to maintain a target voltage level.

The DC/DC converter 1100 A may include a circuit configuration where the level shifters LS 1 to LS 4 of FIG. 7 are added into the DC/DC converter 300 ′ illustrated in FIG. 6 . In the DC/DC converter 1100 A, a negative source voltage VSSH of a driver DR 1 and a negative source voltage VSS_IM of a driver DR 2 may be applied from the high-side voltage regulator circuit 310 . Also, a positive source voltage PVDD of a driver DR 4 and a positive source voltage VDD_IM of a driver DR 3 may be applied from the low-side voltage regulator circuit 320 .

The feedback circuit 1300 A may include first to third amplifier circuits 1300 A- 1 to 1300 A- 3 . The output voltage VOUT, a voltage VDDM, and a voltage VSSM may be input from the DC/DC converter 1100 A to the feedback circuit 1300 A. The voltage VDDM may be a voltage that is detected from a node Nd 1 of the DC/DC converter 1100 A, and the voltage VSSM may be a voltage that is detected from a node Nd 2 of the DC/DC converter 1100 A.

The first amplifier circuit 1300 A- 1 may be configured with an operational (OP) amplifier Ala and two impedances Z 1 a and Z 2 a . The first amplifier circuit 1300 A- 1 may amplify a difference between the output voltage VOUT of the DC/DC converter 1100 A and a reference source REF 1 to output a first error signal Verr 1 , based on a gain value of the first amplifier circuit 1300 A- 1 . A gain value of the first amplifier circuit 1300 A- 1 may be adjusted based on a ratio of the two impedances Z 1 a and Z 2 a.

The second amplifier circuit 1300 A- 2 may be configured with an OP amplifier A 2 a . The second amplifier circuit 1300 A- 2 may amplify a difference between the voltage VDDM and the voltage VSSM input from the DC/DC converter 1100 A to output a voltage Vcf. For example, a gain value of the second amplifier circuit 1300 A- 2 may be set to 1.

The third amplifier circuit 1300 A- 3 may be configured with an OP amplifier A 3 a and two impedances Z 3 a and Z 4 a . The third amplifier circuit 1300 A- 3 may amplify a difference between an input voltage VIN of the DC/DC converter 1100 A and the voltage Vcf output from the second amplifier circuit 1300 A- 2 to output a second error signal Verr 2 , based on a gain value of the third amplifier circuit 1300 A- 3 . A gain value of the third amplifier circuit 1300 A- 3 may be adjusted based on a ratio of the two impedances Z 3 a and Z 4 a.

The first error signal Verr 1 and the second error signal Verr 2 generated by the feedback circuit 1300 A may be output to the controller 1400 A.

The controller 1400 A may include an adder 1400 A- 1 , a subtractor 1400 A- 2 , and first and second comparators CMP 1 a and CMP 2 a.

The controller 1400 A may compare an operational combination of the first and second error signals Verr 1 and Verr 2 with a triangular wave having a 180-degree phase difference with the operational combination to generate a plurality of control signals P 1 to P 4 as follows.

The adder 1400 A- 1 may summate a voltage of the first error signal Verr 1 and a voltage of the second error signal Verr 2 to generate a first signal SUM 1 . The subtractor 1400 A- 2 may generate a second signal SUB 1 corresponding to a difference between the first error signal Verr 1 and the second error signal Verr 2 . In detail, the subtractor 1400 A- 2 may subtract the voltage of the second error signal Verr 2 from the voltage of the first error signal Verr 1 to generate the second signal SUB 1 .

The first comparator CMP 1 a may compare a voltage level of the first signal SUM 1 with a voltage level of a first triangular wave signal S 1 and generate the control signals P 1 and P 4 , based on a result of the comparison. Therefore, the control signals P 1 and P 4 may be square waves having the same phase.

The second comparator CMP 2 a may compare a voltage level of the second signal SUB 1 with a voltage level of a second triangular wave signal S 2 and generate the control signals P 2 and P 3 , based on a result of the comparison. Here, the second triangular wave signal S 2 may be implemented to have a 180-degree phase difference with the first triangular wave signal S 1 . Therefore, the control signals P 2 and P 3 may be square waves having the same phase.

Switching duties of a plurality of transistors M 1 to M 4 included in the DC/DC converter 1100 A may be adjusted by the generated control signals P 1 to P 4 , and thus, the voltage Vcf corresponding to the difference between the voltage VDDM and the voltage VSSM may be controlled to VIN/2, thereby obtaining a desired output voltage VOUT.

For example, circuits implemented as one semiconductor chip among the circuit elements configuring the power supply 1000 A of FIG. 13 are referred to by CH 1 .

FIG. 14 is a diagram illustrating another example 1000 B of a detailed circuit configuration of the power supply 1000 illustrated in FIG. 12 .

Referring to FIG. 14 , a power supply 1000 B may include a DC/DC converter 1100 B, a feedback circuit 1300 B, a controller 1400 B, a high-side voltage regulator circuit 310 , and a low-side voltage regulator circuit 320 . For reference, a load circuit 20 which is connected between an output terminal and a ground terminal GND of the DC/DC converter 1100 B is illustrated. FIG. 14 illustrates a circuit that controls an output current IOUT to maintain a target current level.

The DC/DC converter 1100 B may include a circuit configuration where the level shifters LS 1 to LS 4 of FIG. 7 are added into the DC/DC converter 300 ′ illustrated in FIG. 6 . In the DC/DC converter 1100 B, a negative source voltage VSSH of a driver DR 1 and a negative source voltage VSS_IM of a driver DR 2 may be applied from the high-side voltage regulator circuit 310 . Also, a positive source voltage PVDD of a driver DR 4 and a positive source voltage VDD_IM of a driver DR 3 may be applied from the low-side voltage regulator circuit 320 .

The DC/DC converter 1100 B may include a current sensor 30 that senses the output current IOUT flowing from an output terminal OUT to the load circuit 20 .

The feedback circuit 1300 B may include first to third amplifier circuits 1300 B- 1 to 1300 B- 3 . The output current IOUT, a voltage VDDM, and a voltage VSSM may be input from the DC/DC converter 1100 B to the feedback circuit 1300 B. The voltage VDDM may be a voltage that is detected from a node Nd 1 of the DC/DC converter 1100 B, and the voltage VSSM may be a voltage that is detected from a node Nd 2 of the DC/DC converter 1100 B.

The first amplifier circuit 1300 B- 1 may be configured with an OP amplifier A 1 b and two impedances Z 1 b and Z 2 b . The first amplifier circuit 1300 B- 1 may amplify a difference between the output current IOUT and a reference source REF 2 to output a first error signal Verr 1 , based on a gain value of the first amplifier circuit 1300 B- 1 . A gain value of the first amplifier circuit 1300 B- 1 may be adjusted based on a ratio of the two impedances Z 1 b and Z 2 b.

The second amplifier circuit 1300 B- 2 may be configured with an OP amplifier A 2 b . The second amplifier circuit 1300 B- 2 may amplify the difference between the voltage VDDM and the voltage VSSM input from the DC/DC converter 1100 B to output a voltage Vcf. For example, a gain value of the second amplifier circuit 1300 B- 2 may be set to 1.

The third amplifier circuit 1300 B- 3 may be configured with an OP amplifier A 3 b and two impedances Z 3 b and Z 4 b . The third amplifier circuit 1300 B- 3 may amplify a difference between an input voltage VIN of the DC/DC converter 1100 B and the voltage Vcf output from the second amplifier circuit 1300 B- 2 to output a second error signal Verr 2 , based on a gain value of the third amplifier circuit 1300 B- 3 . A gain value of the third amplifier circuit 1300 B- 3 may be adjusted based on a ratio of the two impedances Z 3 b and Z 4 b.

The first error signal Verr 1 and the second error signal Verr 2 generated by the feedback circuit 1300 B may be output to the controller 1400 B.

The controller 1400 B may include an adder 1400 B- 1 , a subtractor 1400 B- 2 , and first and second comparators CMP 1 b and CMP 2 b.

The controller 1400 B may compare an operational combination of the first and second error signals Verr 1 and Verr 2 with a triangular wave having a 180-degree phase difference with the operational combination to generate a plurality of control signals P 1 to P 4 as follows.

The adder 1400 B- 1 may summate a voltage of the first error signal Verr 1 and a voltage of the second error signal Verr 2 to generate a first signal SUM 1 . The subtractor 1400 B- 2 may generate a second signal SUB 1 corresponding to a difference between the first error signal Verr 1 and the second error signal Verr 2 . In detail, the subtractor 1400 B- 2 may subtract the voltage of the second error signal Verr 2 from the voltage of the first error signal Verr 1 to generate the second signal SUB 1 .

The first comparator CMP 1 b may compare a voltage level of the first signal SUM 1 with a voltage level of a first triangular wave signal S 1 and generate the control signals P 1 and P 4 , based on a result of the comparison. Therefore, the control signals P 1 and P 4 may be square waves having the same phase.

The second comparator CMP 2 b may compare a voltage level of the second signal SUB 1 with a voltage level of a second triangular wave signal S 2 and generate the control signals P 2 and P 3 , based on a result of the comparison. Here, the second triangular wave signal S 2 may be implemented to have a 180-degree phase difference with the first triangular wave signal S 1 . Therefore, the control signals P 2 and P 3 may be square waves having the same phase.

Switching duties of a plurality of transistors M 1 to M 4 included in the DC/DC converter 1100 B may be adjusted by the generated control signals P 1 to P 4 , and thus, the voltage Vcf corresponding to the difference between the voltage VDDM and the voltage VSSM may be controlled to VIN/2, thereby obtaining a desired output current TOUT. Accordingly, a constant current circuit for controlling an output current not to be changed may be implemented.

For example, circuits implemented as one semiconductor chip among the circuit elements configuring the power supply 1000 B of FIG. 14 are referred to by CH 2 .

FIG. 15 is a diagram illustrating another example 1000 C of a detailed circuit configuration of the power supply 1000 illustrated in FIG. 12 .

Referring to FIG. 15 , a power supply 1000 C may include a DC/DC converter 1100 C, a feedback circuit 1300 C, a controller 1400 C, a high-side voltage regulator circuit 310 , and a low-side voltage regulator circuit 320 . For reference, a load circuit 20 which is connected between an output terminal and a ground terminal GND of the DC/DC converter 1100 C is illustrated. FIG. 15 illustrates a circuit that controls an input current IIN to maintain a target current level.

The DC/DC converter 1100 C may include a circuit configuration where the level shifters LS 1 to LS 4 of FIG. 7 are added into the DC/DC converter 300 ′ illustrated in FIG. 6 . In the DC/DC converter 1100 C, a negative source voltage VSSH of a driver DR 1 and a negative source voltage VSS_IM of a driver DR 2 may be applied from the high-side voltage regulator circuit 310 . Also, a positive source voltage PVDD of a driver DR 4 and a positive source voltage VDD_IM of a driver DR 3 may be applied from the low-side voltage regulator circuit 320 .

The DC/DC converter 1100 C may include a current sensor 40 that senses the input current IIN flowing from an input terminal IN to a transistor M 1 of the DC/DC converter 1100 C.

The feedback circuit 1300 C may include first to third amplifier circuits 1300 C- 1 to 1300 C- 3 . The input current IIN, a voltage VDDM, and a voltage VSSM may be input from the DC/DC converter 1100 C to the feedback circuit 1300 C. The voltage VDDM may be a voltage that is detected from a node Nd 1 of the DC/DC converter 1100 C, and the voltage VSSM may be a voltage that is detected from a node Nd 2 of the DC/DC converter 1100 C.

The first amplifier circuit 1300 C- 1 may be configured with an OP amplifier A 1 c and two impedances Z 1 c and Z 2 c . The first amplifier circuit 1300 C- 1 may amplify a difference between the input current RN and a reference source REF 3 to output a first error signal Verr 1 , based on a gain value of the first amplifier circuit 1300 C- 1 . A gain value of the first amplifier circuit 1300 C- 1 may be adjusted based on a ratio of the two impedances Z 1 c and Z 2 c.

The second amplifier circuit 1300 C- 2 may be configured with an OP amplifier A 2 c . The second amplifier circuit 1300 C- 2 may amplify the difference between the voltage VDDM and the voltage VSSM input from the DC/DC converter 1100 C to output a voltage Vcf. For example, a gain value of the second amplifier circuit 1300 C- 2 may be set to 1.

The third amplifier circuit 1300 C- 3 may be configured with an OP amplifier A 3 c and two impedances Z 3 c and Z 4 c . The third amplifier circuit 1300 C- 3 may amplify a difference between an input voltage VIN of the DC/DC converter 1100 C and the voltage Vcf output from the second amplifier circuit 1300 C- 2 to output a second error signal Verr 2 , based on a gain value of the third amplifier circuit 1300 C- 3 . A gain value of the third amplifier circuit 1300 C- 3 may be adjusted based on a ratio of the two impedances Z 3 c and Z 4 c.

The first error signal Verr 1 and the second error signal Verr 2 generated by the feedback circuit 1300 C may be output to the controller 1400 C.

The controller 1400 C may include an adder 1400 C- 1 , a subtractor 1400 C- 2 , and first and second comparators CMP 1 c and CMP 2 c.

The controller 1400 C may compare an operational combination of the first and second error signals Verr 1 and Verr 2 with a triangular wave having a 180-degree phase difference with the operational combination to generate a plurality of control signals P 1 to P 4 as follows.

The adder 1400 C- 1 may summate a voltage of the first error signal Verr 1 and a voltage of the second error signal Verr 2 to generate a first signal SUM 1 . The subtractor 1400 C- 2 may generate a second signal SUB 1 corresponding to a difference between the first error signal Veal and the second error signal Verr 2 . In detail, the subtractor 1400 C- 2 may subtract the voltage of the second error signal Verr 2 from the voltage of the first error signal Verr 1 to generate the second signal SUB 1 .

The first comparator CMP 1 c may compare a voltage level of the first signal SUM 1 with a voltage level of a first triangular wave signal S 1 and generate the control signals P 1 and P 4 , based on a result of the comparison. Therefore, the control signals P 1 and P 4 may be square waves having the same phase.

The second comparator CMP 2 c may compare a voltage level of the second signal SUB 1 with a voltage level of a second triangular wave signal S 2 and generate the control signals P 2 and P 3 , based on a result of the comparison. Here, the second triangular wave signal S 2 may be implemented to have a 180-degree phase difference with the first triangular wave signal S 1 . Therefore, the control signals P 2 and P 3 may be square waves having the same phase.

Switching duties of a plurality of transistors M 1 to M 4 included in the DC/DC converter 1100 C may be adjusted by the generated control signals P 1 to P 4 , and thus, the voltage Vcf corresponding to the difference between the voltage VDDM and the voltage VSSM may be controlled to VIN/2, thereby obtaining a desired input current IIN. Accordingly, a constant power circuit for controlling input power not to be changed may be implemented.

For example, circuits implemented as one semiconductor chip among the circuit elements configuring the power supply 1000 C of FIG. 15 are referred to by CH 3 .

FIG. 16 is a diagram illustrating another example 1000 D of a detailed circuit configuration of the power supply 1000 illustrated in FIG. 12 .

Referring to FIG. 16 , a power supply 1000 D may include a DC/DC converter 1100 D, a feedback circuit 1300 D, a controller 1400 D, a high-side voltage regulator circuit 310 , and a low-side voltage regulator circuit 320 . For reference, a load circuit 20 which is connected between an output terminal and a ground terminal GND of the DC/DC converter 1100 D is illustrated. FIG. 16 illustrates a circuit that controls an input current IIN to maintain a target level.

The DC/DC converter 1100 D may include a circuit configuration where the level shifters LS 1 to LS 4 of FIG. 7 are added into the DC/DC converter 300 ′ illustrated in FIG. 6 . In the DC/DC converter 1100 D, a negative source voltage VSSH of a driver DR 1 and a negative source voltage VSS_IM of a driver DR 2 may be applied from the high-side voltage regulator circuit 310 . Also, a positive source voltage PVDD of a driver DR 4 and a positive source voltage VDD_IM of a driver DR 3 may be applied from the low-side voltage regulator circuit 320 .

The DC/DC converter 1100 D may include a current sensor 30 that senses an output current IOUT flowing from an output terminal OUT to the load circuit 20 . The DC/DC converter 1100 D may include a current sensor 40 that senses the input current IIN flowing from an input terminal IN to a transistor M 1 of the DC/DC converter 1100 D.

The feedback circuit 1300 D may include a multiplexer 50 and first to third amplifier circuits 1300 D- 1 to 1300 D- 3 . The input current IIN, the output current IOUT, an output voltage VOUT, a voltage VDDM, and a voltage VSSM may be input from the DC/DC converter 1100 D to the feedback circuit 1300 D. The voltage VDDM may be a voltage that is detected from a node Nd 1 of the DC/DC converter 1100 D, and the voltage VSSM may be a voltage that is detected from a node Nd 2 of the DC/DC converter 1100 D.

The multiplexer 50 may select a signal of one input terminal from among the input current TIN, the output current IOUT, and the output voltage VOUT input to an input terminal according to a selection signal SEL and output the selected signal to an output terminal.

The first amplifier circuit 1300 D- 1 may be configured with an OP amplifier A 1 d and two impedances Z 1 d and Z 2 d . The first amplifier circuit 1300 D- 1 may amplify a difference between one signal output from the multiplexer 50 and a reference source REF 4 to output a first error signal Verr 1 , based on a gain value of the first amplifier circuit 1300 D- 1 . When the input current IN or the output current IOUT is selected by the multiplexer 50 , the selected current may be a reference current, and when the output voltage VOUT is selected, the selected voltage may be a reference voltage.

A gain value of the first amplifier circuit 1300 D- 1 may be adjusted based on a ratio of the two impedances Z 1 d and Z 2 d.

The second amplifier circuit 1300 D- 2 may be configured with an OP amplifier A 2 d . The second amplifier circuit 1300 D- 2 may amplify the difference between the voltage VDDM and the voltage VSSM input from the DC/DC converter 1100 D to output a voltage Vcf. For example, a gain value of the second amplifier circuit 1300 D- 2 may be set to 1.

The third amplifier circuit 1300 D- 3 may be configured with an OP amplifier A 3 d and two impedances Z 3 d and Z 4 d . The third amplifier circuit 1300 D- 3 may amplify a difference between an input voltage VIN of the DC/DC converter 1100 D and the voltage Vcf output from the second amplifier circuit 1300 D- 2 to output a second error signal Verr 2 , based on a gain value of the third amplifier circuit 1300 D- 3 . A gain value of the third amplifier circuit 1300 D- 3 may be adjusted based on a ratio of the two impedances Z 3 d and Z 4 d.

The first error signal Verr 1 and the second error signal Verr 2 generated by the feedback circuit 1300 D may be output to the controller 1400 D.

The controller 1400 D may include an adder 1400 D- 1 , a subtractor 1400 D- 2 , and first and second comparators CMP 1 d and CMP 2 d.

The controller 1400 D may compare an operational combination of the first and second error signals Verr 1 and Verr 2 with a triangular wave having a 180-degree phase difference with the operational combination to generate a plurality of control signals P 1 to P 4 as follows.

The adder 1400 D- 1 may summate a voltage of the first error signal Verr 1 and a voltage of the second error signal Verr 2 to generate a first signal SUM 1 . The subtractor 1400 D- 2 may generate a second signal SUB 1 corresponding to a difference between the first error signal Verr 1 and the second error signal Verr 2 . In detail, the subtractor 1400 D- 2 may subtract the voltage of the second error signal Verr 2 from the voltage of the first error signal Verr 1 to generate the second signal SUB 1 .

The first comparator CMP 1 d may compare a voltage level of the first signal SUM 1 with a voltage level of a first triangular wave signal S 1 and generate the control signals P 1 and P 4 , based on a result of the comparison. Therefore, the control signals P 1 and P 4 may be square waves having the same phase.

The second comparator CMP 2 d may compare a voltage level of the second signal SUB 1 with a voltage level of a second triangular wave signal S 2 and generate the control signals P 2 and P 3 , based on a result of the comparison. Here, the second triangular wave signal S 2 may be implemented to have a 180-degree phase difference with the first triangular wave signal S 1 . Therefore, the control signals P 2 and P 3 may be square waves having the same phase.

Switching duties of a plurality of transistors M 1 to M 4 included in the DC/DC converter 1100 C may be adjusted by the generated control signals P 1 to P 4 , and thus, the voltage Vcf corresponding to the difference between the voltage VDDM and the voltage VSSM may be controlled to VIN/2, thereby obtaining a desired input current IIN, a desired output current IOUT, or a desired output voltage VOUT. Accordingly, a constant current circuit that controls the input current IIN not to be changed according to a signal selected by the multiplexer 50 may be implemented, or a constant voltage circuit that controls the output voltage VOUT not to be changed according to a signal selected by the multiplexer 50 may be implemented.

FIG. 17 shows an example of waveforms of main signals of the DC/DC converter 300 ( 300 ′) and the power supply 1000 A ( 1000 B to 1000 D) according to an exemplary embodiment. FIG. 17 shows waveforms of main signals when duties of the control signals P 2 and P 3 are equal to or lower than 0.5.

In T 1 and T 3 sections, since the control signals P 1 and P 4 are in a high state and the control signals P 2 and P 3 are in a high state, the transistors M 1 and M 2 may be turned off, and the transistors M 3 and M 4 may be turned on. Therefore, the voltage VDDM may become VIN/2, and the voltage VSSM may become 0 V. Accordingly, a current may flow from the output terminal OUT to the ground terminal GND through the inductor L 1 and the transistors M 3 and M 4 .

In a T 2 section, since the control signals P 1 and P 4 are in a low state and the control signals P 2 and P 3 are in a high state, the transistors M 1 and M 3 may be turned on, and the transistors M 2 and M 4 may be turned off. Therefore, the voltage VDDM may become VIN, and the voltage VSSM may become VIN/2.

In a T 4 section, since the control signals P 1 and P 4 are in a high state and the control signals P 2 and P 3 are in a low state, the transistors M 1 and M 3 may be turned off, and the transistors M 2 and M 4 may be turned on. Therefore, the voltage VDDM may become VIN, and the voltage VSSM may become VIN/2.

Referring to FIG. 10 , in the T 2 and T 4 sections, when the voltage VSSM becomes 0 V and the voltage VDD_IM is lower than the voltage PVDD, the diode D 2 may be turned on, and thus, the voltage VDD_IM may be charged to “PVDD−VD”. Here, VD is a diode forward voltage.

FIG. 18 shows another example of waveforms of main signals of the DC/DC converter 300 ( 300 ′) and the power supply 1000 A ( 1000 B to 1000 D) according to an exemplary embodiment. FIG. 18 shows waveforms of main signals when duties of the control signals P 2 and P 3 are equal to or higher than 0.5.

In T 1 ′ and T 3 ′ sections, since the control signals P 1 and P 4 are in a low state and the control signals P 2 and P 3 are in a low state, the transistors M 1 and M 2 may be turned on, and the transistors M 3 and M 4 may be turned off. Therefore, the voltage VDDM may become VIN, and the voltage VSSM may become VIN/2. Accordingly, a current may flow from the input terminal IN to the output terminal OUT through the transistors M 1 and M 2 and the inductor L 1 .

In a T 2 ′ section, since the control signals P 1 and P 4 are in a high state and the control signals P 2 and P 3 are in a low state, the transistors M 1 and M 3 may be turned off, and the transistors M 2 and M 4 may be turned on. Therefore, the voltage VDDM may become VIN/2, and the voltage VSSM may become 0 V.

In a T 4 ′ section, since the control signals P 1 and P 4 are in a low state and the control signals P 2 and P 3 are in a high state, the transistors M 1 and M 3 may be turned on, and the transistors M 2 and M 4 may be turned off. Therefore, the voltage VDDM may become VIN/2, and the voltage VSSM may become 0 V.

Referring to FIG. 10 , in the T 1 ′ and T 3 ′ sections, when the voltage VSSM becomes VIN and the voltage VSS_IM is higher than the voltage VSSH, the diode D 1 may be turned on, and thus, the voltage VSS_IM may be discharged to “VSSH+VD”. Here, VD is a diode forward voltage.

Referring to FIG. 11 , in the T 2 ′ and T 4 ′ sections, when the voltage VSSM becomes 0 V and the voltage VDD_IM is lower than the voltage PVDD, the diode D 2 may be turned on, and thus, the voltage VDD_IM may be charged to “PVDD−VD”. Here, VD is a diode forward voltage.

Since the DC/DC converter 300 ( 300 ′) is driven with the source voltages of the drivers generated by the high-side voltage regulator circuit 310 and the low-side voltage regulator circuit 320 as illustrated in FIGS. 10 and 11 , a correlation between the driving voltage and the input voltage of the drivers is eliminated. Therefore, even when an input voltage applied to an input terminal is high, a plurality of power switches (for example, the transistors M 1 to M 4 ) are turned on with a constant voltage. Therefore, the power switches have a constant conduction resistance irrespective of the input voltage, and thus, stable switching operations of the power switches are performed. That is, even when the input voltage is low, the power switches operate by using a low conduction resistance.

FIG. 19 illustrates an example of a configuration of an electronic device 2000 to which the power supply according to an exemplary embodiment is applied.

Referring to FIG. 19 , the electronic device 2000 may include a battery 2100 , a power management circuit 2200 , and first to Nth processors 2300 - 1 to 2300 -N (where N is an integer equal to or more than two).

The electronic device 2000 may include, for example, a personal computer (PC), a mobile terminal, a personal digital assistant (PDA), a portable multimedia player (PMP), an MP3 player, a camera, a camcorder, a television (TV), a display apparatus, and/or the like.

The battery 2100 may supply DC power to the electronic device 2000 . For example, the battery 2100 may generate a DC voltage “V BAT ” and output the DC voltage to the power management circuit 2200 .

The power management circuit 2200 may include a plurality of power supplies 2200 - 1 to 2200 -N. Each of the plurality of power supplies 2200 - 1 to 2200 -N may use the power supply 1000 illustrated in FIG. 12 . For example, each of the plurality of power supplies 2200 - 1 to 2200 -N may use one or more of the power supplies 1000 A to 1000 D illustrated in FIGS. 13 to 16 . The plurality of power supplies 2200 - 1 to 2200 -N may generate different voltages V 1 to VN.

The first to Nth processors 2300 - 1 to 2300 -N may operate with respective voltages generated by the plurality of power supplies 2200 - 1 to 2200 -N. For example, the first to Nth processors 2300 - 1 to 2300 -N may be a baseband processor, an application processor, and/or the like.

As illustrated in FIG. 20 , an electronic device 3000 may include a power supply 3100 , a central processing unit (CPU) 3200 , a signal processor 3300 , a user interface 3400 , a storage 3500 , a device interface 3600 , and a bus 3700 .

The electronic device 3000 may include, for example, a PC, a mobile terminal, a PDA, a PMP, an MP3 player, a camera, a camcorder, a TV, a display apparatus, and/or the like.

The power supply 3100 may generate a source voltage necessary for the electronic device 3000 and supply the source voltage to a plurality of elements. The power supply 3100 may use the power supply 1000 according to an exemplary embodiment. For example, the power supply 3100 may use one or more of the power supplies 1000 A to 1000 D illustrated in FIGS. 13 to 16 .

The CPU 3200 may overall control the electronic device 3000 . For example, the CPU 3200 may control the elements of the electronic device 3000 , based on information input through the user interface 3400 .

The signal processor 3300 may process a signal received through the device interface 3600 or a signal read from the storage 3500 according to the defined standard. For example, the signal processor 3300 may perform video signal processing, audio signal processing, and/or the like.

The user interface 3400 may be an input unit that is used for a user to set information necessary for a function setting and an operation of the electronic device 3000 .

The storage 3500 may store various pieces of information necessary for an operation of the electronic device 3000 . Also, the storage 3500 may store data received through the device interface 3600 or pieces of data obtained through processing by the electronic device 3000 .

The device interface 3600 may perform data communication with an external device which is connected to the electronic device 300 by wire or wirelessly.

The bus 3700 may transmit information between the elements of the electronic device 3000 .

While the inventive concepts have been particularly shown and described with reference to exemplary embodiments thereof, it will be understood that various changes in form and details may be made therein without departing from the spirit and scope of the following claims.