Method and apparatus to reconfigure a filter

This present disclosure claims the benefit of U.S. Provisional Application No. 62/008,847, “Reconfigurable Low Pass Filter Structure with High Linearity and Low Gain Loss” filed on Jun. 6, 2014, which is incorporated herein by reference in its entirety.

The background description provided herein is for the purpose of generally presenting the context of the disclosure. Work of the presently named inventors, to the extent the work is described in this background section, as well as aspects of the description that may not otherwise qualify as prior art at the time of filing, are neither expressly nor impliedly admitted as prior art against the present disclosure.

Filters are widely used in communication devices. In an example, operational transconductance amplifier (OTA) based filters are often used in transmitter circuit and receiver circuit in a communication device. The OTA based filters are required to satisfy, for example, noise and linearity requirements in order for the communication devices to satisfy performance requirements.

Aspects of the disclosure provide a circuit having a filter circuit and a controller. The filter circuit drives a load circuit having different input impedances under different operation conditions. The filter circuit is configured to have a first output circuit coupled with a first resistor and a second output circuit coupled with a second resistor. The controller is configured to generate control signals to select one of the first output circuit and the second output circuit based on an operation condition of the load circuit.

According to an aspect of the disclosure, the load circuit is a mixer circuit configured to operate in a plurality of frequency bands, and the controller is configured to generate the control signals to select one of the first output circuit and the second output circuit based on an operation frequency of the mixer circuit. In an example, the load circuit includes a voltage mode passive mixer having an input impedance that varies with a frequency of a carrier signal.

In an embodiment, the first resistor has a higher resistance than the second resistor, and the controller is configured to select the first output circuit when the load circuit operates at a first frequency, and select the second output circuit when the load circuit operates at a second frequency that is higher than the first frequency.

In an example, the first output circuit includes a first transistor in a common gate arrangement cascaded with a second transistor in a common source arrangement to form a first driving path to drive an output node coupled with the first resistor, and a third transistor in the common gate arrangement cascaded with a fourth transistor in the common source arrangement to form a second driving path to drive the output node coupled with the first resistor. Further, the first output circuit includes a plurality of switches at gate terminals of the first transistor and the third transistor to be switched on/off to select or dis-select the first output circuit.

In an embodiment, the filter circuit includes an input stage configured to receive an input voltage and amplify the input voltage to generate an intermediate signal that is provided to the first driving path and a level shift circuit configured to voltage-shift the intermediate signal to generate a shifted intermediate signal that is provided to the second driving path.

Aspects of the disclosure provide a method. The method includes determining an operation condition of a load circuit having different input impedance under different operation conditions, generating control signals based the an operation condition, and providing the control signals to select one of a first output circuit and a second output circuit in a filter circuit coupled with the load circuit. The first output circuit is coupled with a first resistor and the second output circuit is coupled with a second resistor.

Aspects of the disclosure provide a transmitter circuit that includes a mixer circuit, a filter circuit, and a controller. The mixer circuit is configured to have different impedances under different operation frequency bands. The filter circuit is configured to have a first output circuit coupled with a first resistor to drive the mixer circuit and a second output circuit coupled with a second resistor to drive the mixer circuit. The controller is configured to generate control signals to select one of the first output circuit and the second output circuit based on an operation frequency of the mixer circuit.

Various embodiments of this disclosure that are proposed as examples will be described in detail with reference to the following figures, wherein like numerals reference like elements, and wherein:

FIG. 1 shows a block diagram of an electronic device 100 according to an embodiment of the disclosure;

FIG. 2 shows a schematic diagram of an operational transconductance amplifier 240 according to an embodiment of the disclosure; and

FIG. 3 shows a flow chart outlining a process example 300 according to an embodiment of the disclosure.

FIG. 1 shows a diagram of an electronic device 100 according to an embodiment of the disclosure. The electronic device 100 includes a filter 120 having an output stage with multiple switchable outputs (MOs), and a configuration controller, such as a low pass filter (LPF) configuration controller 119 . The configuration controller 119 is configured to generate control signals to switch the outputs of the filter 120 to configure the filter 120 to be used in different operation conditions, such as different frequency bands, and the like.

The electronic device 100 can be any suitable device, such as a desktop computer, a laptop computer, a tablet computer, a smart phone, a network switch, an access point, a router, a set-top box, a television, and the like, that includes a filter which may need to operate in different operation conditions.

In an example, the electronic device 100 is a communication terminal device, such as a smart phone and the like, configured according to a standard for wireless communication, such as a Long-Term Evolution (LTE) standard for wireless communication of high-speed data. The electronic device 100 includes a transmitter 110 configured according to the LTE standard to transmit high speed data. The transmitter 110 has a relatively high requirement for filter linearity and gain loss.

Specifically, in the FIG. 1 example, the transmitter 110 includes various circuit components, such as a digital to analog converter (DAC) module 111 , a first processing path (I-path) 112 , a second processing path (Q-path) 113 , a first mixer 114 , a second mixer 115 and the like coupled together as shown in FIG. 1 .

In an example, the DAC module 111 respectively converts an in-phase component and a quadrature component of data for transmission from a digital form to an analog form. The first processing path 112 filters the in-phase component to remove/reduce high frequency components that are the noise introduced during processing. The first mixer 114 receives and modulates an in-phase component (LOI) of a carrier signal to carry the in-phase component of the data and generates the in-phase component of the signal for transmission. Similarly, the second processing path 113 filters the quadrature component to remove/reduce high frequency components that are the noise introduced during processing. The second mixer 115 receives and modulates the quadrature component (LOQ) of the carrier signal to carry the quadrature component of the data and generates the quadrature component of the signal for transmission. The in-phase component and quadrature component of the signal for transmission are suitably combined as mixer output.

Specifically, in the FIG. 1 example, the first processing path 112 is a third-order low pass filter that includes an OTA based low pass filter 120 followed by passive low pass filters formed of resistors 121 - 124 and capacitors 125 - 126 . The OTA based low pass filter 120 is in a differential biquad (second-order) filter topology. In addition, the OTA based low pass filter 120 has multiple pairs of differential outputs, and can selectively drive one pair of differential outputs based on control signals from the LPF configuration controller 119 . For example, the low pass filter 120 generates a first pair of differential outputs OUTP 1 and OUTM 1 , and a second pair of differential outputs OUTP 2 and OUTM 2 . Based on the control signals from the LPF configuration controller 119 , the OTA based low pass filter 120 can drive the first pair of differential outputs OUTP 1 and OUTM 1 or can drive the second pair of differential outputs OUTP 2 and OUTM 2 .

The resistors 121 - 124 and the capacitors 125 - 126 form passive low pass filters that are coupled to the outputs of the OTA based low pass filter 120 as shown in FIG. 1 . Specifically, the resistor 121 is coupled to the output OUTM 1 and the capacitor 125 , the resistor 122 is coupled to the output OUTM 2 and the capacitor 125 , the resistor 123 is coupled to the output OUTP 2 and the capacitor 126 , and the resistor 124 is coupled to the output OUTP 1 and the capacitor 126 . The resistors 121 and 124 have a same resistance of R 5 _ 1 , and the resistors 122 and 123 have a same resistance of R 5 _ 2 . The capacitors 125 and 126 have a same capacitance C 3 .

In an embodiment, the first mixer 114 is a voltage mode passive mixer with input impedance varying with frequency. For example, the input impedance is high when the carrier signal is in a low frequency band, and the input impedance is low when the carrier signal is in a medium frequency band. According to an aspect of the disclosure, in order to keep relatively constant gain by the third-order low pass filter in the processing path 112 across multiple frequency bands, the OTA based low pass filter 120 selectively drives a pair of differential outputs according to the frequency of the carrier signal.

In an example, R 5 _ 1 is about 100 ohm and R 5 _ 2 is about 50 ohm. When the carrier signal is in a first frequency band, the OTA based low pass filter 120 drives the outputs OUTP 1 and OUTM 1 ; and when the carrier signal is in a second frequency band with higher frequencies than the first frequency band, the OTA based low pass filter 120 drives the outputs OUTP 2 and OUTM 2 .

In a related example, an OTA based low pass filter drives a single pair of outputs. Further, switches are used to couple the outputs to different resistors. In the related example, the switches can cause gain loss. To reduce gain loss, the switches are implemented using large size transistors. The large size transistors introduce excessive parasitic capacitance, and affect OTA stability.

According to an aspect of the disclosure, the OTA based low pass filter 120 is implemented using OTAs with high linearity to improve the filter linearity. Specifically, the OTA based low pass filter 120 includes a first OTA 130 , a second OTA 140 , resistors 131 - 132 of resistance R 3 , resistors 135 - 136 of resistance R 1 , resistors 141 - 142 of R 2 , resistors 147 - 148 of resistance R 4 _ 1 , resistors 149 - 150 of resistance R 4 _ 2 , capacitors 133 - 134 of capacitance C 1 , capacitors 143 - 144 of capacitance C 2 _ 1 , capacitors 145 - 146 of capacitance C 2 _ 2 . Further, the OTA based low pass filter 120 includes a plurality of switches S 1 -S 8 . The switches S 1 -S 8 are switched on/off based on control signals provided by the LPF configuration controller 119 . These elements are coupled together as shown in FIG. 1 .

In the FIG. 1 example, the first OTA 130 , the resistors 131 - 132 and 135 - 136 and the capacitors 133 - 134 form a leaky integrator. The switches S 5 -S 8 selectively couple the capacitors 143 - 144 or the capacitors 145 - 146 with the second OTA 140 and the resistors 141 - 142 to form an integrator. Further, the switches S 1 -S 4 selectively couple the resistors 147 - 148 or the resistors 149 - 150 with the leaky integrator and the integrator to form the OTA based low pass filter 120 which is a second-order (biquad) low pass filter.

According to an aspect of the disclosure, the second OTA 140 has an output stage having multiple output circuits that respectively drive the multiple pair of differential outputs. In an embodiment, the second OTA 140 can be controlled to select one of the output circuits to drive one pair of differential the multiple outputs, and disconnect the other outputs circuits. In the FIG. 1 example, the second OTA 140 can be controlled to drive the first pair of differential outputs OUTP 1 and OUTM 1 or can be controlled to drive the second pair of differential outputs OUTP 2 and OUTM 2 . The first OTA 130 can be implemented using any suitable topology.

In an embodiment, the LPF configuration controller 119 is configured to generate control signals based on the frequency of a local oscillator (LO) signal, and provide the control signals to control the switches S 1 -S 8 and the second OTA 140 . In an example, the carrier signal is generated based on the local oscillator signal. For example, when the frequency of the LO signal is in a first frequency band, the LPF configuration controller 119 is configured to generate the control signals to close the switches S 2 , S 3 , S 6 and S 7 , open the switches S 1 , S 4 , S 5 and S 8 , and control the second OTA 140 to drive the first pair of differential outputs OUTP 1 and OUTM 1 ; and when the frequency of the LO signal is in a second frequency band with frequencies higher than the first frequency band, the LPF configuration controller 119 is configured to generate the control signals to open the switches S 2 , S 3 , S 6 and S 7 , close the switches S 1 , S 4 , S 5 and S 8 , and control the second OTA 140 to drive the second pair of differential outputs OUTP 2 and OUTM 2 .

The LPF configuration controller 119 can be implemented using various techniques. In an example, the LPF configuration controller 119 is implemented using circuits. In another example, the LPF configuration controller 119 is implemented as a processing circuitry executing software instructions.

It is noted that, in an example, the transmitter 110 includes other components that are not shown, such as power amplifier, encoder, and the like. In an embodiment, the transmitter 110 is integrated with other circuit on an integrated circuit (IC) chip.

It is also noted that the OTA based low pass filter 120 can be suitably modified to have more than two pair of differential outputs.

It is also noted that the second processing path 113 can be similarly configured as the first processing path 112 .

FIG. 2 shows a schematic diagram of an operational transconductance amplifier (OTA) 240 according to an embodiment of the disclosure. The OTA 240 includes an input stage 250 , a first level shift circuit 270 , a second level shift circuit 280 , and an output stage having a first output circuit 260 to drive a first pair of differential outputs OUTP 1 and OUTM 1 , and a second output circuit 290 to drive a second pair of differential outputs OUTP 2 and OUTM 2 . The first output circuit 260 and the second output circuit 290 can be selected to drive the first pair of differential outputs OUTP 1 and OUTM 1 or the second pair of differential outputs OUTP 2 and OUTM 2 . In an example, the OTA 240 can be used in the FIG. 1 example as the second OTA 140 .

In the FIG. 2 example, the input stage 250 receives a pair of differential voltage inputs INP and INM, amplifies, and outputs a pair of differential intermediate outputs OUT 1 P and OUT 1 M. The input stage 250 includes N-type metal-oxide-semiconductor field effect transistor (MOSFET) transistors N 1 -N 4 and P-type MOSFET transistors P 1 -P 6 coupled together as shown in FIG. 2 .

Specifically, the P-type MOSFET transistors P 3 and P 1 are cascaded to form a first cascode amplifier with the P-type MOSFET transistor P 3 in a common-gate arrangement and the P-type MOSFET transistor P 1 in a common-source arrangement. Similarly, the P-type MOSFET transistor P 4 and P 2 are cascaded to form a second cascode amplifier. The first cascode amplifier and the second cascode amplifier are coupled to form a differential pair to receive the differential voltage inputs INP and INM. The P-type MOSFET transistor P 5 and P 6 form PMOS cascode current source with both transistors in the common-gate arrangement. The N-type MOSFET transistors N 1 and N 3 are cascaded to form a first NMOS cascode current source and the N-type MOSFET transistors N 2 and N 4 are cascaded to form a second NMOS cascode current source. The first NMOS cascode current source and the second NMOS cascode current source are coupled with the cascode differential amplifier to act as NMOS cascode loads.

During operation, in an example, when the differential voltage inputs INP and INM are of the same voltage level, the first cascode amplifier and the second cascode amplifier are balanced, and the current provided by the PMOS cascode current source is equally divided between the first and second cascode amplifier. In an example, the first NMOS cascode current source and the second NMOS cascode current source are configured to each conduct one half of the current as the PMOS cascode current source, then the differential intermediate outputs OUT 1 M and OUT 1 P are of the same voltage level.

When the differential voltage inputs INP and INM are of different voltage levels, the current provided by the PMOS cascode current source is imbalanced between the first cascode amplifier and the second cascode amplifier. In an example, the first NMOS cascode current source and the second NMOS cascode current source are configured to each conduct one half of the current as the PMOS cascode current source, then the differential intermediates outputs OUT 1 M and OUT 1 P are of different voltage levels and have output current.

The first level shift circuit 270 and the second level shift circuit 280 suitably shift the intermediate outputs OUT 1 P and OUT 1 M to generate the shifted intermediate output OUT 1 P_S and OUT 1 M_S to enable a class AB arrangement at the first output circuit 260 and the second output circuit 290 .

The first output circuit 260 is configured to operate in class AB arrangement to drive the first pair of differential current outputs OUTP 1 and OUTM 1 . According to the class AB arrangement, an output node is driven by two switching paths, and each of the switching paths has a conduction angle larger than 180° in response to an input. In an example, when the input is a sinusoidal signal. One of the switching paths drives the output node when the input has an angle in the range of [0°, 180°], and other switching path drives the output node when the input has an angle in the range of [180°, 360°]. The conduction angles of the two switching path overlap at about 0° and about 180°.

In the FIG. 2 example, the first output circuit 260 includes N-type MOSFET transistors N 12 -N 15 , and P-type MOSFET transistors P 14 -P 17 coupled together as shown in FIG. 2 .

In the example, the N-type MOSFET transistors N 12 and N 13 form a first NMOS cascode amplifier to drive the current output OUTP 1 in response to the intermediate output OUT 1 M, the N-type MOSFET transistors N 14 and N 15 form a second NMOS cascode amplifier to drive the current output OUTM 1 in response to the intermediate output OUT 1 P.

Similarly, the P-type MOSFET transistors P 14 and P 15 form a first PMOS cascode amplifier to drive the current output OUTP 1 in response to the shifted intermediate output OUT 1 M_S, and the P-type MOSFET transistors P 16 and P 17 form a second PMOS cascode amplifier to drive the current output OUTM 1 in response to the shifted intermediate output OUT 1 P-S.

According to an aspect of the disclosure, the first output circuit 260 is suitably biased and controlled to operate in class AB arrangement, such that the first cascode PMOS amplifier and the first cascode NMOS amplifier are not simultaneously off, and the second cascode PMOS amplifier and the second cascode PMOS amplifier are not simultaneously off.

The first level shifter 270 and the second level shifter 280 can use any suitable technique to shift the intermediate outputs OUT and OUT to enable the class AB arrangement.

The second output circuit 290 is similarly configured as the first output circuit 260 to operate in class AB arrangement to drive the second pair of differential current outputs OUTP 2 and OUTM 2 . In the FIG. 2 example, the second output circuit 290 includes N-type MOSFET transistors N 16 -N 19 , and P-type MOSFET transistors P 18 -P 21 coupled together as shown in FIG. 2 .

In the example, the N-type MOSFET transistors N 16 and N 17 form a first NMOS cascode amplifier to drive the current output OUTP 2 in response to the intermediate output OUT 1 M, the N-type MOSFET transistors N 18 and N 19 form a second NMOS cascode amplifier to drive the current output OUTM 2 in response to the intermediate output OUT 1 P.

Further, the P-type MOSFET transistors P 18 and P 19 form a first PMOS cascode amplifier to drive the current output OUTP 2 in response to the shifted intermediate output OUT 1 M_S, and the P-type MOSFET transistors P 20 and P 21 form a second PMOS cascode amplifier to drive the current output OUTM 2 in response to the shifted intermediate output OUT 1 P-S.

Similarly configured as the first output circuit 260 , the second output circuit 290 is suitably biased and controlled to operate in class AB arrangement, such that the first cascode PMOS amplifier and the first cascode NMOS amplifier are not simultaneously off, and the second cascode PMOS amplifier and the second cascode PMOS amplifier are not simultaneously off.

The OTA 240 includes switches SN 1 -SN 4 and SP 1 -SP 4 that are controlled to select the first output circuit 260 or the second output circuit 290 . In an embodiment, the switches SN 1 -SN 4 and SP 1 -SP 4 are implemented using transistors.

In an example, to select the first output circuit 260 , the switches SP 1 , SN 1 , SP 4 and SN 4 are closed, and the switches SP 2 , SN 2 , SP 3 , and SN 3 are open; and to select the second output circuit 290 , the switches SP 1 , SN 1 , SP 4 and SN 4 are open, and the switches SP 2 , SN 2 , SP 3 , and SN 3 are closed. In an embodiment, the switches SN 1 -SN 4 and SP 1 -SP 4 are controlled according to signals generated by a configuration controller, such as the LPF configuration controller 119 .

According to an aspect of the disclosure, the switches SN 1 -SN 4 and SP 1 -SP 4 are coupled to gate terminals of the transistors N 12 , N 14 , N 16 , N 18 , P 15 , P 16 , P 18 and P 20 having the common-gate arrangement to select the first output circuit 260 or the second output circuit 290 . The switches SN 1 -SN 4 and SP 1 -SP 4 are not on the signal amplification path. Thus, the switches SN 1 -SN 4 and SP 1 -SP 4 do not cause gain loss and thus can be implemented using relatively small size transistors to increase LPF stability. In addition, the resistor adjustment for the passive low pass filter does not affect the LPF linearity.

It is noted that the OTA 240 can include other suitable circuits that are not shown. In an example, the OTA 240 includes a bias circuit (not shown) to generate suitable bias voltages, such as cs_bp, cas_bp, cas_bp1, cas_bp2, cs_bn, cas_bn, cas_bn1 and the like.

In a simulation example, the OTA 240 is used in the processing path 112 . The processing path 112 achieves 37.5 dBc improvement in the third-order harmonic distortion, more than 10° phase margin improvement, and about 1 dB voltage swing improvement.

FIG. 3 shows a flow chart outlining a process example 300 according to an embodiment of the disclosure. In an example, the process 300 is executed by the LPF configuration controller 119 to generate control signals to configure the first processing path 112 and the second processing path 113 . The process starts at 5301 and proceeds to S 310 .

At S 310 , a frequency band for transmission is determined. In an example, signals indicative of a frequency of a local oscillator (LO) are provided to the LPF configuration controller 119 . The LO generates a periodic signal of the frequency, and the periodic signal is used as a carrier signal for transmission in an example. The LPF configuration controller 119 determines the frequency band of the frequency for transmission.

At S 320 , resistors and capacitors are determined. In an example, the LPF configuration controller 119 determines suitable resistors and capacitors to use in the first processing path 112 and the second processing path 113 to achieve a relatively constant low pass filter gain across frequency bands.

At 5330 , control signals to control the switches to select OTA outputs, resistors and capacitors are generated and provided to the suitable switches. For example, the LPF configuration controller 119 generates suitable control signals to control the switches SN 1 -SN 4 and SP 1 -SP 4 to select an output circuit coupled to the determined resistors for the passive low pass filters. Further, the LPF configuration controller 119 generates and provides suitable control signals to control the switches S 1 -S 8 . Then, the process proceeds to S 399 and terminates.

When implemented in hardware, the hardware may comprise one or more of discrete components, an integrated circuit, an application-specific integrated circuit (ASIC), etc.

While aspects of the present disclosure have been described in conjunction with the specific embodiments thereof that are proposed as examples, alternatives, modifications, and variations to the examples may be made. Accordingly, embodiments as set forth herein are intended to be illustrative and not limiting. There are changes that may be made without departing from the scope of the claims set forth below.