Switching power supply device

The present invention relates to a switching power supply device designed to improve power conversion efficiency.

As a switching power supply device for various kinds of electronic instruments, a resonant converter is known. The resonant converter is configured by connecting a primary winding of an isolation transformer to a direct current voltage source via a capacitor. A series resonant circuit is formed of the leakage inductance of the isolation transformer and the capacitor. The resonant converter controls a current flowing through the series resonant circuit using first and second switching elements which are complementarily on/off driven, and obtains a stepped-up/down direct current voltage from the side of secondary windings of the isolation transformer.

A soft switching technology in this kind of switching power supply device is proposed in, for example, U.S. Pat. Nos. 5,886,884 and 7,391,194. The soft switching technology is such as to significantly reduce loss in the switching elements by turning on the switching elements when a voltage applied to the switching elements is zero (0) or when a current flowing through the inductance is zero (0). Also, as the switching power supply device, a multi-oscillated current resonant converter which causes the series resonant circuit to perform a current resonance operation is also proposed. The multi-oscillated current resonant converter, as well as causing the first switching element to perform a separately excited oscillation operation with a drive signal pulse-width controlled in accordance with an output voltage, causes the second switching element to perform a self-oscillation operation utilizing a voltage generated in an auxiliary winding of the isolation transformer.

This kind of resonance type switching power supply device includes a series resonant circuit which, by connecting a primary winding P 1 of an isolation transformer T to a direct current voltage source B via a capacitor C, as shown in, for example, FIG. 5 , is formed of the leakage inductance of the isolation transformer T and the capacitor C. A first switching element Q 1 connected in series to the primary winding P 1 of the isolation transformer T, by being driven by a drive control circuit A which performs a separately excited oscillation operation, applies a direct current input voltage Vin from the direct current voltage source B to the series resonant circuit.

The drive control circuit A is formed of, for example, a power supply IC (Integrated Circuit). Also, a second switching element Q 2 connected in parallel to the series resonant circuit, by being on-driven by the drive control circuit A when the first switching element Q 1 is turned off, forms a resonant current path of the series resonant circuit. The first and second switching elements Q 1 and Q 2 are each formed of, for example, a high-voltage n-type MOS-FET (Metal-Oxide-Semiconductor Field-Effect Transistor).

Power generated in secondary windings S 1 and S 2 of the isolation transformer T is rectified and smoothed via an output circuit formed of diodes D 1 and D 2 and an output capacitor Cout, and supplied to an unshown load as an output voltage Vout. A resonance type power conversion device main body is constructed by these circuit portions. Also, the output voltage Vout obtained in the output circuit, specifically, the deviation between the output voltage Vout and an output voltage set value, is detected by an output voltage detection circuit VS and fed back to the drive control circuit A as an FB voltage via a photocoupler PC. The FB voltage fed back to the drive control circuit A serves to pulse-width modulate an output control signal which on/off drives the first and second switching elements Q 1 and Q 2 , thereby stabilizing the output voltage Vout. Direct current power supplied from the direct current voltage source B, after being filtered via an input capacitor Cin, is supplied to the switching power supply device as the input voltage Vin.

Herein, the drive control circuit A, as an outline configuration thereof is shown in FIG. 6 , is configured mainly of an output control circuit 2 , a winding detection circuit 3 , and a drive signal generation circuit 4 . The output control circuit 2 is formed of a PWM (Pulse-Width Modulation) control circuit which generates, as a PWM signal, an output control signal with a pulse width corresponding to a feedback signal FB fed back from the output voltage detection circuit VS. Also, the winding detection circuit 3 determines the polarity of a voltage generated in a tertiary winding P 3 of the isolation transformer T, and outputs a winding detection signal VW. The drive signal generation circuit 4 , in accordance with the winding detection signal VW and output control signal, generates a pulse-width controlled drive signal of the first switching element Q 1 .

Reference numeral 5 in FIG. 6 is a drive amplifier acting as a drive circuit which drives the first switching element Q 1 upon receiving the drive signal output by the drive signal generation circuit 4 . Also, reference numeral 6 is an internal power supply circuit which, upon receiving a drive voltage VCC applied to the drive control circuit A, generates an internal voltage VDD necessary for the output control circuit 2 , winding detection circuit 3 , and drive signal generation circuit 4 to operate.

A brief description will be given of an operation of the multi-oscillated current resonant converter which is the switching power supply device of the heretofore described configuration. In the multi-oscillated current resonant converter, when the second switching element Q 2 is in an off-state, a current flows through the series resonant circuit by turning on the first switching element Q 1 . In this condition, when the first switching element Q 1 is turned off, an unshown parasitic capacitance of the first switching element Q 1 is charged by a current flowing through the inductance of the series resonant circuit. At the same time, an unshown parasitic capacitance of the second switching element Q 2 is discharged by the current.

Further, when a voltage charged in the parasitic capacitance of the first switching element Q 1 reaches the direct current input voltage Vin, the second switching element Q 2 is turned on, thereby realizing the zero-voltage switching of the second switching element Q 2 . As a result of turning on the second switching element Q 2 , power energy accumulated in the capacitor C flows via the second switching element Q 2 . Consequently, the current flowing through the inductance of the series resonant circuit is inverted.

Subsequently, when the second switching element Q 2 is turned off, the parasitic capacitance of the second switching element Q 2 is charged by the current inverted in the way heretofore described. At the same time, the parasitic capacitance of the first switching element Q 1 is discharged by the inverted current. Further, it is detected, from an inversion of the polarity of the voltage generated in the tertiary winding P 3 , that the voltage charged in the parasitic capacitance of the second switching element Q 2 reaches a zero (0) voltage. The first switching element Q 1 is turned on at this detection timing, thereby realizing the zero-voltage switching of the first switching element Q 1 . By the first switching element Q 1 being turned on, the current of the series resonant circuit is inverted and flows again via the first switching element Q 1 .

Technical Problem

Herein, the winding detection circuit 3 includes two voltage dividing resistors R 1 and R 2 connected in series and a comparator 7 , as shown in, for example, FIG. 7 . The voltage dividing resistors R 1 and R 2 divide the internal voltage VDD or drive voltage VCC and sets a winding threshold voltage VWth. The comparator 7 compares the winding threshold voltage VWth and a winding voltage generated in the tertiary winding P 3 of the isolation transformer T. Further, when the winding voltage exceeds the winding threshold voltage VWth, the comparator 7 , determining from the excess that the polarity of the winding voltage is inverted, outputs the winding detection signal VW.

Herein, the winding threshold voltage VWth is preset based on the specifications of the switching power supply device. Because of this, for example, when the direct current input voltage Vin changes due to the aging, characteristic variation, or the like of parts configuring the switching power supply device, it is undeniable that a gap occurs in the timing of generating the winding detection signal VW. Specifically, at a low input voltage when the direct current input voltage Vin is lower than a voltage set in the specifications, the winding voltage generated in the tertiary winding P 3 decreases. Because of this, the timing of generating the winding detection signal VW is delayed, as shown in, for example, FIG. 8 . Then, a delay occurs in the timing of turning on the first switching element Q 1 , and an ineffective time for power conversion increases. As a result, a current peak lengthens out, loss in the first switching element Q 1 increases, and power conversion efficiency decreases.

In contrast, at a high input voltage when the direct current input voltage Vin is higher than the voltage set in the specifications, the winding voltage generated in the tertiary winding P 3 increases. Thus, the timing of generating the winding detection signal VW is accelerated, as shown in, for example, FIG. 9 . Then, the first switching element Q 1 is turned on before the voltage applied to the first switching element Q 1 reaches zero (0). As a result, the zero-voltage switching cannot be performed, loss in the first switching element Q 1 increases, and power conversion efficiency decreases.

The invention, having been contrived with these kinds of circumstances in mind, can provide a switching power supply device of a simple configuration. It is thereby possible to realize the zero-voltage switching by appropriately setting the timing of turning on the first and second switching elements regardless of a change in the direct current input voltage Vin, and thus possible to prevent a decrease in conversion efficiency.

Solution to Problem

A switching power supply device according to an embodiment of the invention basically includes a series resonant circuit which, by connecting a primary winding P 1 of an isolation transformer T to a direct current voltage source via a capacitor C, is formed of the leakage inductance of the isolation transformer and the capacitor C; a first switching element Q 1 which is driven by a drive control circuit which performs a separately excited oscillation operation, and when turned on, applies a direct current input voltage Vin from the direct current voltage source to the series resonant circuit; a second switching element Q 2 , connected in parallel to the series resonant circuit, which is on-driven by a voltage generated in an auxiliary winding P 2 of the isolation transformer when the first switching element is turned off, thus forming a current path of the series resonant circuit; and an output circuit which rectifies, smooths, and outputs power generated on the side of secondary windings S 1 and S 2 of the isolation transformer.

Furthermore, the switching power supply device according to the invention is such that the drive control circuit is configured of

an output control circuit which generates an output control signal with a pulse width corresponding to an output voltage of the output circuit; a winding detection circuit which compares a voltage generated in a tertiary winding P 3 of the isolation transformer with a winding threshold voltage VWth and outputs a winding detection signal VW; and

a drive signal generation circuit which, in accordance with the winding detection signal and output control signal, generates a pulse-width controlled drive signal which drives the first switching element. Further, in particular, the switching power supply device is characterized in that the drive control circuit is provided with a threshold setting circuit which, by changing the winding threshold voltage VWth set in the winding detection circuit in accordance with the direct current input voltage Vin applied to the series resonant circuit, adjusts the timing of turning on the first switching element Q 1 .

The drive circuit is formed of, for example, one which generates a drive signal with a pulse width with a rise of the winding detection signal VW, which indicates an inversion of the polarity of the voltage generated in the tertiary winding P 3 of the isolation transformer, as an on-trigger and a rise of the output control signal as an off-trigger. Further, the threshold setting circuit is configured so as to set the winding threshold voltage VWth to be high when the direct current input voltage Vin is higher than a preset threshold voltage Vth, and set the winding threshold voltage VWth to be low when the direct current input voltage Vin is lower than the threshold voltage Vth. Preferably, the threshold setting circuit is configured so as to, by switching between analog switches in accordance with a result of the comparison of the direct current input voltage Vin and preset threshold voltage Vth, select one of a plurality of preset winding threshold voltages VWth 1 and VWth 2 to VWthn and set the one in the winding detection circuit.

Advantageous Effects of Invention

According to the switching power supply device of the heretofore described configuration, as the winding threshold voltage VWth is changed in accordance with a change in the direct current input voltage Vin, it is possible to appropriately generate the winding detection signal VW at the timing at which the voltage applied to the first switching element Q 1 reaches zero (0). As a result of this, it is possible to zero-voltage switch the first switching element Q 1 regardless of a change in the direct current input voltage Vin. Therefore, according to the invention, it is possible to effectively suppress an increase in loss in the first switching element Q 1 .

Also, according to the switching power supply device of the previously described configuration, it is possible to prevent a gap in the timing of turning on the first switching element Q 1 resulting from a change in the direct current input voltage Vin, with a simple configuration such as to change the winding threshold voltage VWth in accordance with a change in the direct current input voltage Vin. Consequently, according to the invention, the advantage of it being possible to relax circuit design conditions set with the aging, characteristic variation, or the like of parts configuring the switching power supply device taken into account, or the like, is produced.

FIG. 1 is an outline configuration diagram of a multi-oscillated current resonant converter acting as a switching power supply device according to an embodiment of the invention.

FIG. 2 is a diagram showing a configuration example of a drive control circuit shown in FIG. 1 .

FIG. 3 is a diagram showing a configuration example of a winding detection circuit shown in FIG. 2 .

FIG. 4 is a signal waveform diagram for illustrating an operation of the switching power supply device shown in FIG. 1 .

FIG. 5 is an outline configuration diagram of a multi-oscillated current resonant converter.

FIG. 6 is a diagram showing a configuration example of a drive control circuit shown in FIG. 5 .

FIG. 7 is a diagram showing a configuration example of a winding detection circuit shown in FIG. 6 .

FIG. 8 is a signal waveform diagram for illustrating a problem when an input voltage Vin is low in a switching power supply device shown in FIG. 5 .

FIG. 9 is a signal waveform diagram for illustrating a problem when the input voltage Vin is high in the switching power supply device shown in FIG. 5 .

Hereafter, a description will be given, referring to the drawings, of a switching power supply device according to an embodiment of the invention.

FIG. 1 is an outline configuration diagram of a multi-oscillated current resonant converter acting as the switching power supply device according to the embodiment. The multi-oscillated current resonant converter is basically configured in the same way as a heretofore known multi-oscillated current resonant converter shown in FIG. 5 . Consequently, the same signs are given to portions equivalent to those of the multi-oscillated current resonant converter shown in FIGS. 5 and 6 , and a redundant description thereof will be omitted.

A switching power supply device 1 formed of the multi-oscillated current resonant converter includes voltage dividing resistors Ra and Rb connected in series, as shown in FIG. 1 . The voltage dividing resistors Ra and Rb connected in series, by being connected in parallel to the input capacitor Cin, divide and detect the direct current input voltage Vin. Further, a voltage BO proportional to the direct current input voltage Vin detected via the voltage dividing resistors Ra and Rb is given to a winding detection circuit 3 in the drive control circuit (power supply IC) A which drives the first switching element Q 1 , as shown in FIG. 2 . Further, the switching power supply device 1 is characterized in that the winding detection circuit 3 is configured so as to variably set the previously described winding threshold voltage VWth in accordance with a change in the direct current input voltage Vin.

The winding detection circuit 3 which variably sets the winding threshold voltage VWth in accordance with a change in the direct current input voltage Vin, specifically, includes a first comparator 11 which compares the voltage BO, which is proportional to the direct current input voltage Vin, with a preset threshold voltage Vth, as shown in FIG. 3 . The first comparator 11 assumes the role of complementarily switching between, for example, two analog switches 12 and 13 . Specifically, the first comparator 11 selects one of two kinds of different winding threshold voltages VWth 1 and VWth 2 (VWth1>VWth2), which are set by three voltage dividing resistors R 1 , R 2 , and R 3 connected in series dividing the internal voltage VDD or drive voltage VCC, and gives the one to a second comparator 14 , via the analog switches 12 and 13 . Consequently, the second comparator 14 compares one of the winding threshold voltages VWth 1 and VWth 2 and a winding voltage generated in the tertiary winding P 3 of the isolation transformer T. The second comparator 14 corresponds to the previously described comparator 7 shown in FIG. 7 .

Specifically, when the direct current input voltage Vin is high and the voltage BO proportional to the direct current input voltage Vin is higher than the threshold voltage Vth, the first winding threshold voltage VWth 1 set by the voltage dividing resistors R 1 , R 2 , and R 3 is set in the second comparator 14 . Consequently, the second comparator 14 detects an inversion of the polarity of the winding voltage with the first winding threshold voltage VWth 1 as a reference. Conversely, when the direct current input voltage Vin is low and the voltage BO proportional to the direct current input voltage Vin is lower than the threshold voltage Vth, the second winding threshold voltage VWth 2 (<VWth1) set by the voltage dividing resistors R 1 , R 2 , and R 3 is set in the second comparator 14 . Consequently, the second comparator 14 detects an inversion of the polarity of the winding voltage with the second winding threshold voltage VWth 2 as a reference.

Herein, it is also possible to adopt a configuration such that one of three or more kinds of winding threshold voltages VWth 1 and VWth 2 to VWthn is selected in accordance with the direct current input voltage Vin and set in the second comparator 14 . In this case, a plurality (n−1) of first comparators 11 having different threshold voltages Vth are used to compare the direct current input voltage Vin in parallel. Further, a configuration only has to be such that the respective outputs of the plurality of first comparators 11 are logically processed to alternatively drive the analog switches, thereby selecting one of the plurality of winding threshold voltages VWth 1 and VWth 2 to VWthn and setting the one in the second comparator 14 .

According to the winding detection circuit 3 which compares the winding threshold voltage VWth set in accordance with the direct current input voltage Vin and the winding voltage generated in the tertiary winding P 3 of the isolation transformer T, as heretofore described, the winding threshold voltage VWth is selected and set in accordance with a change in the direct current input voltage Vin, as shown in FIG. 4 . Consequently, even when the winding voltage changes as a result of a change in the direct current input voltage Vin, it is possible to reliably detect a change in the polarity of the winding voltage. Therefore, it is possible to generate a winding detection signal VW at an appropriate timing regardless of a change in the direct current input voltage Vin.

As a result of this, it is possible to turn on the first switching element Q 1 in accordance with the timing of an inversion of the polarity of the winding voltage generated in the tertiary winding P 3 of the isolation transformer T, and thus in accordance with the timing at which a voltage applied to the first switching element Q 1 reaches zero (0). Consequently, it is possible to minimize the switching loss in the first switching element Q 1 and improve the conversion efficiency thereof.

Moreover, as the winding threshold voltage VWth is changed in accordance with a change in the direct current input voltage Vin, it is possible to significantly relax circuit design conditions set with the aging, characteristic variation, or the like of parts configuring the switching power supply device taken into account. Furthermore, as it is possible to easily configure the circuit itself which variably sets the winding threshold voltage VWth in accordance with the direct current input voltage Vin, the advantage of it also being easy to incorporate the circuit into the drive control circuit A formed of the power supply IC, or the like, is produced.

The invention is not limited to the heretofore described embodiment. For example, it goes without saying that the winding threshold voltage VWth set in the second comparator 14 in accordance with a change in the direct current input voltage Vin can be set in three or more stages, as previously described. Also, in the embodiment, a pulse-width modulated PWM signal is shown as a drive control signal, the pulse width of which is varied in accordance with an output voltage Vout, but the drive control signal may be a pulse-frequency modulated PFM signal.

Furthermore, herein, a second switching element Q 2 which self-oscillates is set on a high voltage side, while the first switching element Q 1 which is driven by the drive control circuit which performs a separately excited oscillation operation is set on a low voltage side, but it is also possible to set the two switching elements the other way around. In this case, it is sufficient to use p-type MOS-FETs as the first and second switching elements Q 1 and Q 2 , and set the previously described operation logic in the drive control circuit A in an opposite manner. Apart from this, the invention can be implemented modified in various ways without departing from the scope thereof.