Sensor device

This application is a continuation of PCT/JP2013/051704 filed on Jan. 28, 2013, which claims priority to Japanese Application No. 2012-066004 filed on Mar. 22, 2012. The entire contents of these applications are incorporated herein by reference.

The present invention relates to a sensor device for detecting a physical quantity and converting the detected physical quantity, such as magnetic field intensity, into an electric signal.

A magnetic sensor device is used as a sensor for detecting an open/close state in a flip mobile phone, a notebook computer, or the like and as a sensor for detecting a rotary position of a motor or the like (see, for example, Patent Literature 1). FIG. 15 illustrates a circuit diagram of the magnetic sensor device. In the magnetic sensor device, a magnetoelectric conversion element (such as Hall element) outputs a voltage corresponding to (typically, substantially proportional to) magnetic field intensity (or magnetic flux density), the output voltage is amplified by an amplifier, and a comparator is used to determine whether the amplified output voltage is larger or smaller than a predetermined magnetic field intensity or magnetic flux density (the result is output as a binary value, H signal or L signal).

In general, the output voltage of the magnetoelectric conversion element is minute, and hence there is a problem in that an error may occur due to an offset voltage of the magnetoelectric conversion element (element offset voltage), an offset voltage of the amplifier or the comparator (input offset voltage), and noise of those components, and the accuracy may be lower. The element offset voltage is generated mainly by stress on the magnetoelectric conversion element received from a package. The input offset voltage is generated mainly by characteristic fluctuations in elements of an input circuit of the amplifier or the comparator. The noise is generated mainly by flicker noise of a single transistor of the circuit or thermal noise of a single transistor or a resistive element.

The magnetic sensor device illustrated in FIG. 15 is configured as follows in order to reduce the influence of the offset voltages of the magnetoelectric conversion element and the amplifier. The magnetic sensor device includes a Hall element 1501 , a switch circuit 1502 for switching the Hall element 1501 between a first detection state and a second detection state, a differential amplifier circuit 1503 for amplifying a voltage difference (V 1 −V 2 ) of two output terminals of the switch circuit 1502 , a capacitor 1504 having one end connected to one output terminal of the differential amplifier circuit 1503 , a switch 1506 connected between the other output terminal of the differential amplifier circuit 1503 and the other end of the capacitor 1504 , and a comparator 1505 .

In this case, in the first detection state, a power supply voltage is input from terminals A and C, and a detection voltage is output from terminals B and D. In the second detection state, the power supply voltage is input from the terminals B and D, and a detection voltage is output from the terminals A and C. A differential signal voltage (hereinafter referred to as “element signal voltage”) of the Hall element 1501 corresponding to the magnetic field intensity is represented by Vh, an offset voltage (hereinafter referred to as “element offset voltage”) of the Hall element 1501 is represented by Voh, the gain of the differential amplifier circuit 1503 is represented by G, and an input offset voltage of the differential amplifier circuit 1503 is represented by Voa. The element signal voltages Vh in the first detection state and the second detection state are represented by Vh1 and Vh2, respectively, and the element offset voltages Voh in the first detection state and the second detection state are represented by Voh1 and Voh2, respectively.

The element offset voltages of the Hall element 1501 may be canceled out by a known method, typically called “spinning current”. Specifically, the switch circuit is switched so as to obtain an element offset component that is reverse in phase to a common-mode signal component (or an element offset component that is in-phase to a normal-mode signal component), thereby cancelling out the offset components. Due to the spinning current, Vh2 becomes substantially equal to Vh1, and Voh2 becomes substantially equal to Voh1. V 1 −V 2 =Vh1+Voh1 is established in the first detection state, and V 1 −V 2 =−Vh2+Voh2 is established in the second detection state.

In the first detection state, the switch 1506 is turned ON, and Vc1=(V 3 −V 4 )=G×(V 1 −V 2 )=G×(Vh1+Voh1+Voa) is charged in the capacitor 1504 . Subsequently, in the second detection state, the switch 1506 is turned OFF, and V 3 −V 4 =G×(V 1 −V 2 )=G×(−Vh2+Voh2+Voa) is output from the differential amplifier circuit 1503 .

From the above, a differential input voltage of the comparator 1505 in the second detection state is determined as follows.

V ⁢ ⁢ 5 - V ⁢ ⁢ 6 = ⁢ ( V ⁢ ⁢ 3 - Vc ⁢ ⁢ 1 ) - V ⁢ ⁢ 4 = ⁢ ( V ⁢ ⁢ 3 - V ⁢ ⁢ 4 ) - Vc ⁢ ⁢ 1 = ⁢ G × ( - Vh ⁢ ⁢ 2 + Voh ⁢ ⁢ 2 + Voa ) - G × ⁢ ( Vh ⁢ ⁢ 1 + Voh ⁢ ⁢ 1 + Voa ) = ⁢ G × ( - Vh ⁢ ⁢ 1 - Vh ⁢ ⁢ 2 ) + G × ( Voh ⁢ ⁢ 2 - Voh ⁢ ⁢ 1 )

The influence of the input offset voltage is canceled out, and because Voh2 is substantially equal to Voh1, the influence of the element offset voltage is also canceled out.

In this manner, a compact and inexpensive magnetic sensor can be realized, which is capable of obtaining an output with small fluctuations without being affected by the input offset voltage.

Patent Literature

[PTL 1] JP 2001-337147 A

In the related-art magnetic sensor device as described above, however, the input offset voltage of the comparator 1505 cannot be removed, and hence there is a problem in that a fluctuation or an error occurs in detection magnetic field intensity due to a fluctuation in input offset voltage or a change in input offset voltage caused by stress or the like.

The present invention has been made in view of the above-mentioned circumstances, and it is an object thereof to provide a magnetic sensor device capable of removing the influence of element offset voltages and input offset voltages of an amplifier and a comparator by a simple circuit configuration and detecting magnetic field intensity with high precision.

In order to solve the related-art problems, a sensor device according to one embodiment of the present invention has the following configuration.

According to one embodiment of the present invention, there is provided a sensor device for generating a logic output in accordance with intensity of a physical quantity applied to a sensor element, the sensor device including: a switch circuit connected to a first terminal pair and a second terminal pair of the sensor element, for controlling switching between a terminal pair supplied with power and a terminal pair for outputting a detection voltage corresponding to the intensity of the physical quantity, the switch circuit including a first output terminal and a second output terminal for outputting the detection voltage; a differential amplifier including a first input terminal and a second input terminal that are connected to the first output terminal and the second output terminal of the switch circuit, respectively, and including a first output terminal and a second output terminal for outputting a result of differentially amplifying the detection voltage; an amplifier including a first input terminal, a second input terminal, a third input terminal, a fourth input terminal, and an output terminal, the first input terminal being connected to the output terminal via a first switch, the first input terminal being connected to the first output terminal of the differential amplifier via a first capacitor, the second input terminal being connected to the second output terminal of the differential amplifier; and a detection voltage setting circuit for outputting different voltages to the third input terminal and the fourth input terminal of the amplifier.

The sensor device according to one embodiment of the present invention effectively utilizes the switch and the capacitor, thereby being capable of removing offset components that are generated in the sensor element and the differential amplifier included in the sensor device with a simpler circuit configuration. Consequently, the detection voltage level of the intensity of the physical quantity can be accurately set, and hence a highly-precise sensor device can be provided.

FIG. 1 is a circuit diagram of a magnetic sensor device according to a first embodiment of the present invention.

FIG. 2 is an exemplary circuit diagram of a differential amplifier used in the magnetic sensor device of the present invention.

FIG. 3 is an exemplary circuit diagram of an amplifier used in the first embodiment.

FIG. 4 is an exemplary timing chart of a switch control signal in the magnetic sensor device of the present invention.

FIG. 5 is an exemplary circuit diagram of a detection voltage setting circuit used in the magnetic sensor device of the present invention.

FIG. 6 is an exemplary timing chart of switch control signals in the first embodiment.

FIG. 7 is another exemplary timing chart of the switch control signals in the first embodiment.

FIG. 8 is another exemplary timing chart of the switch control signals in the first embodiment.

FIG. 9 is another exemplary circuit diagram of the detection voltage setting circuit used in the magnetic sensor device of the present invention.

FIG. 10 is an exemplary circuit diagram of the amplifier used in the first embodiment.

FIG. 11 is another exemplary circuit diagram of the differential amplifier used in the magnetic sensor device of the present invention.

FIG. 12 is a circuit diagram of a magnetic sensor device according to a second embodiment of the present invention.

FIG. 13 is an exemplary circuit diagram of an amplifier used in the second embodiment.

FIG. 14 is an exemplary circuit diagram of the amplifier used in the second embodiment.

FIG. 15 is a circuit diagram of a related-art magnetic sensor device.

A sensor device of the present invention, such as a magnetic sensor device, is widely used as a sensor for detecting the state of magnetic field intensity, as represented by a sensor for detecting an open/close state in a flip mobile phone, a notebook computer, or the like or a sensor for detecting a rotary position of a motor or the like. Embodiments of the present invention are described below with reference to the accompanying drawings.

First Embodiment

FIG. 1 is a circuit diagram of a magnetic sensor device according to a first embodiment of the present invention. The magnetic sensor device in the first embodiment includes a Hall element 1 serving as a magnetoelectric conversion element, a switch circuit 2 , a differential amplifier 3 , an amplifier 4 , a capacitor C 1 , a switch S 1 , and a detection voltage setting circuit 5 . The detection voltage setting circuit 5 includes a reference voltage setting circuit Vref 1 and a reference voltage setting circuit Vref 2 .

The Hall element 1 includes a first terminal pair A-C and a second terminal pair B-D.

The switch circuit 2 includes four input terminals connected to the terminals A, B, C, and D of the Hall element 1 , respectively, a first output terminal, and a second output terminal.

The differential amplifier 3 has a first input terminal V 1 , a second input terminal V 2 , a first output terminal V 3 , and a second output terminal V 4 . The first input terminal V 1 and the second input terminal V 2 are connected to the first output terminal and the second output terminal of the switch circuit 2 , respectively.

The capacitor C 1 has two terminals. One terminal is connected to the first output terminal V 3 of the differential amplifier 3 . The other terminal is connected to a first input terminal V 5 of a first differential input pair of the amplifier 4 .

The amplifier 4 has four input terminals and one output terminal. Specifically, the amplifier 4 has the first input terminal V 5 and a second input terminal V 6 of the first differential input pair, a first input terminal V 7 and a second input terminal V 8 of a second differential input pair, and an output terminal VO. The second input terminal V 6 of the first differential input pair of the amplifier 4 is connected to the second output terminal V 4 of the differential amplifier 3 . The first input terminal V 7 of the second differential input pair is connected to a positive terminal of the reference voltage setting circuit Vref 1 . The second input terminal V 8 of the second differential input pair is connected to a positive terminal of the reference voltage setting circuit Vref 2 .

The switch S 1 has two terminals. One terminal is connected to the first input terminal V 5 of the first differential input pair of the amplifier 4 . The other terminal is connected to the output terminal VO of the amplifier 4 . The switch S 1 is controlled to be turned ON or OFF in accordance with a switch control signal (not shown in the circuit diagram).

Next, the operation of the magnetic sensor device in the first embodiment is described below.

The switch circuit 2 has a function of switching between a first detection state and a second detection state. In the first detection state, a power supply voltage is input to the first terminal pair A-C of the Hall element 1 , and a detection voltage is output from the second terminal pair B-D. In the second detection state, the power supply voltage is input to the second terminal pair B-D, and a detection voltage is output from the first terminal pair A-C.

The Hall element 1 outputs a signal voltage corresponding to magnetic field intensity (or magnetic flux density), and also outputs an offset voltage as an error component.

The differential amplifier 3 has a function of amplifying a difference between two input voltages and outputting the amplified difference as a difference between two output voltages. This amplifier function is represented as follows:

V 3− V 4= G ×( V 1− V 2)  (1)

where G is a gain, and V 1 to V 4 are voltages at the respective terminals V 1 to V 4 . This function of the differential amplifier 3 is realized by a circuit configuration illustrated in FIG. 3 , for example.

FIG. 2 is a circuit diagram illustrating an example of the differential amplifier 3 .

The differential amplifier 3 of FIG. 2 includes differential amplifiers 31 and 32 and resistors R 11 , R 12 , and R 13 , which are connected as follows. In the differential amplifier 3 , the first input terminal V 1 is connected to a non-inverting input terminal of the differential amplifier 31 , the second input terminal V 2 is connected to a non-inverting input terminal of the differential amplifier 32 , the first output terminal V 3 is connected to an output terminal of the differential amplifier 31 , and the second output terminal V 4 is connected to an output terminal of the differential amplifier 32 . The resistors R 11 , R 12 , and R 13 are connected in series between the first output terminal V 3 and the second output terminal V 4 . A connection point V 1 ′ of the resistors R 11 and R 12 is connected to an inverting input terminal of the differential amplifier 31 . A connection point V 2 ′ of the resistors R 12 and R 13 is connected to an inverting input terminal of the differential amplifier 32 .

The differential amplifier 3 has the above-mentioned connections and operates as follows.

The differential amplifier 31 operates as a non-inverting amplifier so that the connection point V 1 ′ connected to the inverting input terminal may have a voltage substantially equal to the voltage V 1 connected to the non-inverting input terminal. The differential amplifier 32 operates as a non-inverting amplifier so that the connection point V 2 ′ connected to the inverting input terminal may have a voltage substantially equal to the voltage V 2 connected to the non-inverting input terminal. Equal currents flow through the resistors R 11 , R 12 , and R 13 , and hence the following expressions are obtained.

( V 3− V 1)/ R 11=( V 1− V 2)/ R 12  (2)

( V 2− V 4)/ R 13=( V 1− V 2)/ R 12  (3)

Based on Expression (2) and Expression (3), V 3 and V 4 are calculated as follows.

V 3=+( R 11/ R 12+1/2)×( V 1− V 2)+( V 1+ V 2)/2  (4)

V 4=−( R 13/ R 12+1/2)×( V 1− V 2)+( V 1+ V 2)/2  (5)

The terms in parentheses including the resistors on the right side of Expression (4) and Expression (5) are represented by “gains G1” and “gain G2”, respectively, as follows.

G 1= R 11/ R 12+1/2  (6)

G 2= R 13/ R 12+1/2  (7)

Then, Expression (4) and Expression (5) are modified as follows.

V 3=+ G 1×( V 1− V 2)+( V 1+ V 2)/2  (8)

V 4=− G 2×( V 1− V 2)+( V 1+ V 2)/2  (9)

Based on Expression (8) and Expression (9), V 3 −V 4 is calculated as follows.

V 3− V 4=( G 1+ G 2)×( V 1− V 2)  (10)

When the gain G is given as follows,

G=G 1+ G 2  (11)

Expression (11) is modified as follows,

V 3− V 4= G ×( V 1− V 2)  (12)

to be the same result as Expression (1). In other words, the circuit exemplified in FIG. 2 has the function of amplifying a difference between two input voltages and outputting the amplified difference as a difference between two output voltages. Further, the circuit exemplified in FIG. 2 is configured as such an instrument amplifier to suppress an influence of in-phase noise at input. Note that, Expression (13) is obtained based on Expressions (11), (6), and (7).

G =( R 11+ R 12+ R 13)/ R 12  (13)

Thus, the gain G may be set arbitrarily based on the resistors R 11 , R 12 , and R 13 .

The amplifier 4 has a function of outputting the sum of a value obtained by amplifying the difference of one pair of input voltages and a value obtained by amplifying the difference of another pair of input voltages. This amplifying function is conceptually illustrated in FIG. 3 .

FIG. 3 is a conceptual diagram illustrating the function of the amplifier 4 .

The amplifier 4 of FIG. 3 includes differential amplifiers 41 and 42 and an adder 44 , and is connected as follows. In the amplifier 4 , the first input terminal V 5 of the first differential input pair is connected to an inverting input terminal of the differential amplifier 41 , the second input terminal V 6 of the first differential input pair is connected to a non-inverting input terminal of the differential amplifier 41 , the first input terminal V 7 of the second differential input pair is connected to an inverting input terminal of the differential amplifier 42 , and the second input terminal V 8 of the second differential input pair is connected to a non-inverting input terminal of the differential amplifier 42 . An output of the differential amplifier 41 and an output of the differential amplifier 42 are connected to inputs of the adder 44 , respectively. An output of the adder 44 is connected to the output terminal VO of the amplifier 4 .

The amplifier 4 is connected as described above, and operates as follows.

The differential amplifier 41 amplifies a voltage difference between the two input terminals V 5 and V 6 and inputs the amplified voltage difference to the adder 44 . The differential amplifier 42 amplifies a voltage difference between the two input terminals V 7 and V 8 and inputs the amplified voltage difference to the adder 44 . The adder 44 outputs the sum of the outputs of the differential amplifier 41 and the differential amplifier 42 . This amplifying function is expressed by Expression (14).

VO=A 1×( V 6− V 5)+ A 2×( V 8− V 7)  (14)

where A1 and A2 represent the gains of the differential amplifiers 41 and 42 , and V 5 to V 8 and VO represent the voltages at the terminals V 5 to V 8 and VO, respectively.

In the magnetic sensor device of FIG. 1 , the first input terminal V 5 of the first differential input pair and the output terminal VO of the amplifier 4 illustrated in FIG. 3 are respectively connected to both terminals of the switch S 1 .

When the switch S 1 is turned ON, VO and V 5 are substantially equal voltages, and hence VO is expressed as follows based on Expression (15).

VO=A 1/(1+ A 1)× V 6+ A 2/(1+ A 1)×( V 8− V 7)  (15)

For the sake of description, the gains A1 and A2 are assumed to be sufficiently large. Then, the following expression is obtained.

VO=V 6+( A 2/ A 1)×( V 8− V 7)  (16)

Specifically, when the switch S 1 is turned ON, the output terminal VO of the amplifier 4 is electrically connected to the inverting input terminal of the differential amplifier 41 via the first input terminal V 5 of the first differential input pair, to thereby form a feedback loop. Thus, the output voltage VO follows the input voltage V 6 , and becomes the sum of the input voltage V 6 and a voltage obtained by amplifying the voltage difference between the inputs V 8 and V 7 by the ratio of the gains A2 and A1. In other words, the amplifier 4 operates like a voltage follower.

On the other hand, when the switch S 1 is turned OFF, the feedback loop is not formed in the amplifier 4 , and hence VO operates as a comparator. Based on Expression (14), Expression (17) is obtained.

VO=A 1×{( V 6− V 5)+( A 2/ A 1)×( V 8− V 7)}  (17)

Specifically, when the switch S 1 is turned OFF, the amplifier 4 performs a comparison operation of amplifying, by the sufficiently large gain A1, the sum of a difference voltage between V 6 and V 5 and a voltage obtained by amplifying the difference between V 8 and V 7 by the ratio of the gains A1 and A2, and outputting a High level signal (typically, positive power supply voltage level) or a Low level signal (typically, negative power supply voltage level or GND level) to the output terminal VO.

FIG. 4 illustrates a timing chart of a switch control signal.

One period T of the detection operation is made up of a first phase φ1 and a second phase φ2. The switch S 1 is controlled by the switch control signal of FIG. 4 to be turned ON in the first phase φ1 and OFF in the second phase φ2. The switch circuit 2 and the Hall element 1 are also controlled by the switch control signal of FIG. 4 to have a first detection state T1 in the first phase φ1 and a second detection state T2 in the second phase φ2. The outline of the operation of the magnetic sensor device of FIG. 1 in the respective phases is now described. The first phase φ1 is a phase in which the element signal voltage and the element offset voltage of the Hall element 1 and the offset voltages of the differential amplifier 3 and the amplifier 4 are stored in the capacitor C 1 . The second phase φ2 is a phase in which a voltage determined in accordance with magnetic field intensity indicated by the element signal voltage is compared to a detection voltage while cancelling out the offset components generated in the first phase φ 1 . The details are described below.

In the first phase φ1, the Hall element 1 is in the first detection state T1, and the switch S 1 is turned ON. When the switch S 1 is turned ON, the amplifier 4 operates like a voltage follower as described above. This is expressed as follows based on Expression (16).

V 5φ1= VOφ 1= V 6φ1+( A 2/ A 1)×( V 8φ1− V 7φ1)  (18)

“φ1” suffixed to each voltage means the voltage in the first phase φ1. In the following, the same notation applies to other voltages and the second phase φ2.

Further, a difference ΔVC1φ1 between the voltage V 3 and the voltage V 5 is charged in the capacitor C 1 .

Δ VC 1φ1= V 3φ1− V 5φ1  (19)

Expression (18) is substituted into the above expression to obtain the following expression.

Δ VC 1φ1= V 3φ1− V 6φ1−( A 2/ A 1)×( V 8φ1− V 7φ1)  (20)

In this case, the connection is made so that V 6 =V 4 is established, and the first input terminal V 7 of the second differential input pair of the amplifier 4 is connected to the positive terminal of the reference voltage setting circuit Vref 1 while the second input terminal V 8 of the second differential input pair is connected to the positive terminal of the reference voltage setting circuit Vref 2 . Thus, when the voltages at the positive terminals of the respective reference voltage setting circuits are represented by Vref 1 and Vref 2 , Expression (20) is expressed as the following expression.

Δ VC 1φ1=( V 3φ1− V 4φ1)−( A 2/ A 1)×( V ref2φ1− V ref1φ1)  (21)

Based on Expression (12), Expression (22) is obtained.

V 3φ1− V 4φ1= G ×( V 1φ1− V 2φ1)  (22)

Expression (22) is substituted into Expression (21) to obtain the following expression.

Δ VC 1φ1= G ×( V 1φ1− V 2φ1)−( A 2/ A 1)×( V ref2φ1− V ref1φ1)  (23)

In other words, a voltage difference between the voltage obtained by amplifying the difference between V 1 and V 2 by the gain G and the voltage obtained by amplifying the difference between Vref 2 and Vref 1 by the ratio of the gains A2 and A1 is charged in the capacitor C 1 .

On the other hand, in the second phase φ2, the Hall element 1 is in the second detection state T2, and the switch S 1 is turned OFF. The voltage ΔVC1φ1 is charged in the capacitor C 1 , and hence the voltage V 5 is expressed by the following expression.

V 5φ2= V 3φ2−Δ VC 1φ1  (24)

The output of the amplifier 4 is expressed by the following expression based on Expression (17).

VOφ 2= A 1×{( V 6φ2− V 5φ2)+( A 2/ A 1)×( V 8φ2− V 7φ2)}  (25)

As described above, the connection is made so that V 6 =V 4 is established, and the first input terminal V 7 of the second differential input pair of the amplifier 4 is connected to the positive terminal of the reference voltage setting circuit Vref 1 while the second input terminal V 8 of the second differential input pair is connected to the positive terminal of the reference voltage setting circuit Vref 2 . Thus, Expression (25) is modified as follows.

VOφ 2= A 1×{( V 4φ2− V 5φ2)+( A 2/ A 1)×( V ref2φ2− V ref1φ2)}  (26)

Expression (24) is substituted into this expression to obtain the following expression.

VOφ 2= A 1×{( V 4φ2− V 3φ2)+( A 2/ A 1)×( V ref2φ2− V ref1φ2)}+ A 1×Δ VC 1φ1  (27)

The voltage ΔVC1φ1 charged in the capacitor C 1 , which is expressed by Expression (23), is substituted into the above expression to rearrange Expression (27), thereby obtaining the following expression.

For easy understanding of Expression (28), when a voltage component that is supplied to the amplifier 4 via the differential amplifier 3 is represented by ΔVsig, and a voltage component that is supplied to the amplifier 4 from the reference voltage setting circuit is represented by ΔVref, Expression (28) is expressed as follows.

VOφ 2= A 1×[Δ V sig+( A 2/ A 1)×Δ V ref]  (29)

In this case, the following expressions are established.

Δ V sig=−( V 3φ2− V 4φ2)−( V 3φ1− V 4φ1)  (30)

Δ V ref=( V ref2φ2− V ref1φ2)−( V ref2φ1− V ref1  (31)

Specifically, the result of comparing the voltage component ΔVsig supplied from the differential amplifier 3 to the voltage obtained by amplifying the voltage component ΔVref supplied from the detection voltage setting circuit 5 by the ratio of the gains A1 and A2 is finally output as a High level signal or a Low level signal from the output terminal VO of the amplifier 4 .

Based on Expression (12), Expression (32) is established.

V 3φ2− V 4φ2= G ×( V 1φ2− V 2φ2)  (32)

Thus, Expression (32) and Expression (22) are substituted into Expression (30) to obtain the following expression.

Δ V sig= G ×{( V 1φ2− V 2φ2)−( V 1φ1− V 2φ1)}  (33)

Next, a description is given of the transmission of an effective signal component when a differential output voltage of the output terminal pair of the Hall element 1 is represented by Vh and a common-mode voltage thereof is represented by Vcm. The differential output voltage Vh is a signal voltage corresponding to magnetic field intensity.

In the first phase φ1, the Hall element 1 is in the detection state T1, and the terminals V 1 and V 2 have the following voltages.

V 1φ1= V cmφ1+ Vhφ 1/2  (34)

V 2φ1= V cmφ1− Vhφ 1/2  (35)

From the above expressions and Expression (23), the following expression is obtained.

Δ VC 1φ1= G×Vhφ 1−( A 2/ A 1)×( V ref2φ1− V ref1φ1)  (36)

In the second phase φ2, the Hall element 1 is in the detection state T2, and the terminals V 1 and V 2 have the following voltages.

V 1φ2= V cmφ2− Vhφ 2/2  (37)

V 2φ2= V cmφ2+ Vhφ 2/2  (38)

Expressions (34), (35), (37), and (38) are substituted into Expression (33) to obtain the following expression.

Δ V sig= −G ×( Vhφ 2+ Vhφ 1)  (39)

Further, ΔVref is determined similarly to Expression (31). Therefore, the voltage component ΔVsig supplied to the amplifier 4 via the differential amplifier 3 is a signal component obtained by amplifying the signal voltage corresponding to signal intensity. In the second phase, the amplifier 4 compares the signal component ΔVsig to the reference voltage component ΔVref determined by the reference voltage setting circuit by the expression based on Expression (29), and outputs the comparison result to the terminal VO.

Next, a description is given of the transmission of an element offset component by performing the same calculation when the element offset voltage of the Hall element 1 is represented by Voh. In the above-mentioned calculation of the effective signal component, the output voltage components of the Hall element 1 are reverse in phase between the first detection state T1 and the second detection state T2, and hence the element offset components are in phase therebetween.

In the first phase φ1, the Hall element 1 is in the detection state T1, and the terminals V 1 and V 2 have the following voltages.

V 1φ1= V cmφ1+ Vohφ 1/2  (40)

V 2φ1= V cmφ1− Vohφ 1/2  (41)

From the above expressions and Expression (23), the following expression is obtained.

Δ VC 1φ1= G×Vohφ 1−( A 2/ A 1)×( V ref2φ1− V ref1φ1)  (42)

In the second phase φ2, the Hall element 1 is in the detection state T2, and the terminals V 1 and V 2 have the following voltages.

V 1φ2= V cmφ2+ Vhφ 2/2  (43)

V 2φ2= V cmφ2− Vhφ 2/2  (44)

Expressions (40), (41), (43), and (44) are substituted into Expression (33) to obtain the following expression.

Δ V sig= G ×( Vohφ 2− Vohφ 1)  (45)

In general, the element offset voltage Voh of the Hall element 1 is substantially equal between the first detection state and the second detection state, and hence the value of Vohφ2−Vohφ1 is almost zero. In the comparison operation of the amplifier 4 in the second phase, the element offset component is removed to suppress the influence of the element offset voltage that appears in the output as the comparison result.

Next, a description is given of the transmission of an offset component of the amplifier when the input offset voltages of the differential amplifier 3 are represented by Voa1 for the first input terminal V 1 and Voa2 for the second input terminal V 2 , and the input offset voltages of the respective differential input pairs of the amplifier 4 are represented by Voa3 for the second input terminal V 6 of the first differential input pair and Voa4 for the second input terminal V 8 of the second differential input pair.

In the first phase φ1, the terminals V 1 and V 2 have the following voltages.

V 1φ1= V cmφ1  (46)

V 2φ1= V cmφ1  (47)

From Expression (22) and the above expressions, the following expression is obtained.

V ⁢ ⁢ 3 ⁢ φ1 - V ⁢ ⁢ 4 ⁢ φ1 = ⁢ G × ( V ⁢ ⁢ 1 ⁢ φ1 + Voa ⁢ ⁢ 1 ⁢ φ1 ) - ( V ⁢ ⁢ 2 ⁢ φ1 + Voa ⁢ ⁢ 2 ⁢ φ1 ) = ⁢ G × ( Voa ⁢ ⁢ 1 ⁢ φ1 - Voa ⁢ ⁢ 2 ⁢ φ1 ) ( 48 )

Further, from Expression (18), the following expression is obtained.

V 5φ1= V 6φ1+ Voa 3φ1+( A 2/ A 1)×( V 8φ1+ Voa 4φ1− V 7φ1)  (49)

As described above, V 6 =V 4 , V 7 =Vref 1 , and V 8 =Vref 2 are established, and hence Expression (49) can be expressed as follows.

V 5φ1= V 4φ1+ Voa 3φ1+( A 2/ A 1)×( V ref2φ1− V ref1φ1 +Voa 4φ1)  (50)

Expression (50) is substituted into Expression (19) to obtain the following expression.

Δ VC 1φ1=( V 3φ1− V 4φ1)− Voa 3φ1−( A 2/ A 1)×( V ref2φ1− V ref1φ1+ Voa 4φ1)  (51)

In the second phase φ2, the terminals V 1 and V 2 have the following voltages.

V 1φ2= V cmφ2  (52)

V 2φ2= V cmφ2  (53)

From Expression (33) and the above expressions, the following expression is obtained.

V ⁢ ⁢ 3 ⁢ φ2 - V ⁢ ⁢ 4 ⁢ φ2 = ⁢ G × ( V ⁢ ⁢ 1 ⁢ φ2 + Voa ⁢ ⁢ 1 ⁢ φ2 ) - ( V ⁢ ⁢ 2 ⁢ φ2 + Voa ⁢ ⁢ 2 ⁢ φ2 ) = ⁢ G × ( Voa ⁢ ⁢ 1 ⁢ φ2 - Voa ⁢ ⁢ 2 ⁢ φ2 ) ( 54 )

Further, from Expression (25), the following expression is obtained.

VOφ 2= A 1×{( V 6φ2+ Voa 3φ2− V 5φ2)+( A 2/ A 1)×( V 8φ2+ Voa 4φ2− V 7φ2)}  (55)

As described above, V 6 =V 4 , V 7 =Vref 1 , and V 8 =Vref 2 are established, and hence Expression (55) can be expressed as follows.

VOφ 2= A 1×{( V 4φ2− V 5φ2+ Voa 3φ2)+( A 2/ A 1)×( V ref2φ2− V ref1φ2+ Voa 4φ2)}  (56)

From Expression (24), V 5 φ2=V 3 φ2−ΔVC1φ1 is established, and hence Expression (56) can be expressed as follows.

VOφ 2= A 1×{( V 4φ2− V 3φ2+Δ VC 1φ1+ Voa 3φ2)+( A 2/ A 1)×( V ref2φ2− V ref1φ2+ Voa 4φ2)}  (57)

ΔVC1φ1 in Expression (57) is the voltage charged in the capacitor C 1 in the first phase φ1, and is expressed by Expression (51). ΔVC1φ1 is substituted into Expression (57) to obtain the following expression.

VOφ 2= A 1×[{−( V 3φ2− V 4φ2)+( V 3φ1− V 4φ1)+( Voa 3φ2− Voa 3φ1)}+( A 2/ A 1)×{( V ref2φ2− V ref1φ2)−( V ref2φ1− V ref1φ1)+( Voa 4φ2− Voa 4φ1)}]  (58)

Expressions (48) and (54) are substituted into Expression (58) to obtain the following expression.

VOφ 2= A 1×[ G ×{−( Voa 1φ2− Voa 1φ1)+ G ×( Voa 2φ2− Voa 2φ1)+( Voa 3φ2− Voa 3φ1)}+( A 2/ A 1)×{( V ref2φ2− V ref1φ2)−( V ref2φ1− V ref1φ1)+( Voa 4φ2− Voa 4φ1)}]  (59)

In order to compare the above expression to Expression (29), the voltage component ΔVsig supplied to the amplifier 4 via the differential amplifier 3 and the voltage component ΔVref supplied to the amplifier 4 from the detection voltage setting circuit 5 are introduced into Expression (59) to be modified as follows.

VOφ 2= A 1×[ G×ΔV sig+( A 2/ A 1)×Δ V ref]  (60)

Expression (60) is the same as Expression (29). In Expression (60), the following expressions are established.

Δ V sig={−( Voa 1φ2− Voa 1φ1)+ G ×( Voa 2φ2− Voa 2φ1)+( Voa 3φ2− Voa 3φ1)}  (61)

Δ V ref=( V ref2φ2− V ref1φ2)−( V ref2φ1− V ref1φ1)+( Voa 4φ2− Voa 4φ1)  (62)

In a strict sense, the offset voltages Voa1 to Voa4 of the differential amplifier 3 and the amplifier 4 have a temporal change or a temperature change (temperature drift). However, if the time periods of the first phase φ1 and the second phase φ2 are sufficiently short with respect to the temporal change or the temperature change of those offset voltages, the values of the offset voltages are regarded as values substantially equal between the first phase φ1 and the second phase φ2. Therefore, in Expression (59), the values of Voa1φ2−Voa1 φ1, Voa2φ2−Voa2φ1, Voa3φ2−Voa3φ1, and Voa4φ2−Voa4φ1 are almost zero, and hence the offset components of the differential amplifier 3 and the amplifier 4 are removed in the comparison operation of the amplifier 4 in the second phase.

In conclusion, as expressed by Expressions (39), (45), and (59), the magnetic sensor device of the present invention may compare the signal component of the Hall element to the reference voltage while removing all offset components generated in the Hall element 1 , the differential amplifier 3 , and the amplifier 4 , and hence highly-precise magnetic field intensity detection is realized. In an ideal Hall element, the common-mode voltages Vcmφ1 and Vcmφ2 in the first detection state and the second detection state are equal to each other. In an actual Hall element, however, the common-mode voltages Vcmφ1 and Vcmφ2 are not always equal to each other, which is also responsible for an error in highly-precise magnetic field intensity detection. In the magnetic sensor device of the present invention, as expressed by Expression (59), the terms of Vcmφ1 and Vcmφ2 are not included in the expression indicating the comparison result, and hence highly-precise magnetic field intensity detection without a non-ideal component of the common-mode voltage of the Hall element is realized.

As expressed by Expression (31), the reference voltage component ΔVref to be compared to the signal component of the Hall element may be arbitrarily set by the values of the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 in the first phase φ1 and the second phase φ2. In other words, in the magnetic sensor device of the present invention, the reference voltage may be arbitrarily set, and hence the magnetic field intensity to be detected may be arbitrarily set.

In general, the sensitivity of the Hall element depends on temperature, and hence the signal voltage corresponding to magnetic field intensity output from the Hall element 1 also depends on temperature. In order to correct the dependence, for example, temperature dependence is given to the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 , to thereby suppress the temperature dependence of the magnetic field intensity to be detected.

Now, a description is given of an exemplary circuit configuration of the detection voltage setting circuit 5 , which is a component of the magnetic sensor device of FIG. 1 .

FIG. 5 is an example of the detection voltage setting circuit 5 .

The detection voltage setting circuit 5 of FIG. 5 includes resistors R 51 , R 52 , and R 53 and switches S 51 , S 51 x , S 52 , and S 52 x , which are connected as follows. The resistors R 51 , R 52 , and R 53 are connected in series between a positive power supply voltage terminal (hereinafter referred to as power supply voltage terminal) VDD and a negative power supply voltage terminal (hereinafter referred to as ground terminal) VSS. A connection point of the resistors R 51 and R 52 is represented by “Vn 1 ”, and a connection point of the resistors R 52 and R 53 is represented by “Vn 2 ”. The switches S 51 , SS 1 x , S 52 , and S 52 x each have two terminals, and are controlled to be turned ON or OFF in accordance with switch control signals (not shown). One terminal of the switch S 51 is connected to the connection point Vn 1 , and the other terminal is connected to a positive terminal of the reference voltage setting circuit Vref 1 . One terminal of the switch S 51 x is connected to the connection point Vn 2 , and the other terminal is connected to the positive terminal of the reference voltage setting circuit Vref 1 . One terminal of the switch S 52 is connected to the connection point Vn 1 , and the other terminal is connected to a positive terminal of the reference voltage setting circuit Vref 2 . One terminal of the switch S 52 x is connected to the connection point Vn 2 , and the other terminal is connected to the positive terminal of the reference voltage setting circuit Vref 2 . In the following description, voltages at the power supply voltage terminal VDD and the ground terminal VSS are represented by “VDD” and “VSS”, respectively; voltages at the connection points Vn 1 and Vn 2 , “Vn 1 ” and “Vn 2 ”, respectively; voltages at the positive terminal of the reference voltage setting circuit Vref 1 and the positive terminal of the reference voltage setting circuit Vref 2 , “reference voltage Vref 1 ” and “reference voltage Vref 2 ”, respectively.

The detection voltage setting circuit 5 has the above-mentioned connections, and operates as follows.

The voltages at the connection points Vn 1 and Vn 2 are the following voltages obtained by dividing VDD and VSS by the resistors R 51 , R 52 , and R 53 , respectively.

Vn 1= R 51/( R 51+ R 52+ R 53)×( VDD−VSS )  (63)

Vn 2=( R 51+ R 52)/( R 51+ R 52+ R 53)×( VDD−VSS )  (64)

The voltages Vn 1 and Vn 2 can be set arbitrarily based on the resistors R 51 , R 52 , and R 53 .

The switches S 51 and S 51 x are controlled so that one switch is turned ON and the other is turned OFF. Therefore, any one of the voltages Vn 1 and Vn 2 is output as the reference voltage Vref 1 . Similarly, the switches S 52 and S 52 x are controlled so that one switch is turned ON and the other is turned OFF. Therefore, any one of the voltages Vn 1 and Vn 2 is output as the reference voltage Vref 2 .

Now, it is assumed that the respective switches of the detection voltage setting circuit 5 illustrated in FIG. 5 are controlled by switch control signals of a timing chart illustrated in FIG. 6 .

FIG. 6 illustrates the timing chart of the switch control signals.

As described above, the switch S 1 is controlled by the switch control signal to be turned ON in the first phase φ1 and OFF in the second phase φ2. The switches S 51 and S 51 x are controlled by the switch control signals. The switch S 51 is turned ON both in φ1 and φ2, and the switch S 51 x is turned OFF both in φ1 and φ2. The switches S 52 and S 52 x are controlled by the switch control signals. The switch S 52 is turned ON in φ1 and OFF in φ2. The switch S 51 x is turned OFF both in φ1 and φ2. In this case, when the switch S 1 is turned OFF, in order to prevent an error in a voltage charged in the capacitor C 1 in the first phase φ1, it is necessary to delay the timing of switching the switches S 52 and S 52 x as compared to the timing of turning OFF the switch S 1 . For clear illustration, the timings are exaggerated in the timing chart of FIG. 6 . Note that, in the timing chart of FIG. 6 , also when the switch S 1 is turned ON, the timing of switching the switches S 52 and S 52 x is delayed. However, when the switch S 1 is turned ON, the timing of turning ON the switch S 1 and the timing of switching the switches S 52 and S 52 x may be the same. In contrast, the timing of switching the switches S 52 and S 52 x may be earlier.

Through the above-mentioned control of the switches, the reference voltages Vref 1 and Vref 2 in the respective phases are determined as follows.

V ref1φ1= Vn 1

V ref1φ2= Vn 1

V ref2φ1= Vn 1

V ref2φ2= Vn 2

Based on the expressions above and Expression (31), Expression (65) is obtained.

Δ V ref=( Vn 2− Vn 1)  (65)

Therefore, the voltage ΔVref to be compared to the signal component from the Hall element in the amplifier 4 is given as the difference between the voltages Vn 1 and Vn 2 that may be arbitrarily set. As described above, according to the magnetic sensor device of the present invention, the reference voltage may be arbitrarily set, in other words, the magnetic field intensity to be detected may be arbitrarily set.

Next, it is assumed that the respective switches of the detection voltage setting circuit 5 illustrated in FIG. 5 are controlled by switch control signals of a timing chart illustrated in FIG. 7 .

FIG. 7 illustrates an exemplary timing chart of the switch control signals. The timing chart of FIG. 7 is different from the timing chart of FIG. 6 in that the switch S 51 is turned OFF in φ1 and ON in φ2 and the switch S 51 x is turned ON in φ1 and OFF in φ2. Through the above-mentioned control of the switches, the reference voltages Vref 1 and Vref 2 in the respective phases are determined as follows.

V ref1φ1= Vn 2

V ref1φ2= Vn 1

V ref2φ1= Vn 1

V ref2φ2= Vn 2

Based on the above expressions and Expression (31), Expression (66) is obtained.

Δ V ref=2×( Vn 2− Vn 1)  (66)

Therefore, the voltage that is twice the voltage ΔVref obtained in the timing chart of FIG. 6 is obtained. Specifically, by switching the control of ON and OFF of the switches S 51 , S 51 x , S 52 , and S 52 x in the state where the values of the resistors R 51 , R 52 , and R 53 are uniquely determined and the voltages at the connection points Vn 1 and Vn 2 are uniquely determined, the magnetic field intensity to be detected may be switched.

Alternatively, it is assumed that the respective switches of the detection voltage setting circuit 5 illustrated in FIG. 5 are controlled by switch control signals of a timing chart illustrated in FIG. 8 .

FIG. 8 illustrates an exemplary timing chart of the switch control signals. The timing chart of FIG. 8 is different from the timing chart of FIG. 6 in that the switch S 51 is turned OFF in φ2, the switch S 51 x is turned ON in φ2, the switch S 52 is turned ON both in the phases φ1 and φ2, and the switch S 52 x is turned OFF both in the phases φ1 and φ2.

Through the above-mentioned control of the switches, the reference voltages Vref 1 and Vref 2 in the respective phases are determined as follows.

V ref1φ1= Vn 1

V ref1φ2= Vn 2

V ref2φ1= Vn 1

V ref2φ2= Vn 1

Based on the above expressions and Expression (31), Expression (67) is obtained.

Δ V ref=−( Vn 2− Vn 1)  (67)

Therefore, the voltage that is reverse in polarity to the voltage ΔVref obtained in the timing chart of FIG. 6 is obtained. In this case, the characteristics of the Hall element show that the polarity of the differential output voltage Vh to be output to the output terminal pair of the Hall element 1 is reversed between the S-pole and the N-pole. Specifically, the signal component input to the amplifier 4 is reversed in polarity between the detection of the S-pole and the detection of the N-pole. It is therefore necessary for the detection voltage setting circuit 5 to set detection voltages having reverse polarities. In the case of the timing chart of FIG. 8 , ΔVref with the polarity reverse to that in the case of the timing chart of FIG. 6 is obtained, which satisfies the above-mentioned requirement. Specifically, by switching the switches to be turned ON and OFF between the first phase φ1 and the second phase φ2 so as to reverse the polarity of ΔVref, the detection may be performed while discriminating between the S-pole and the N-pole.

FIG. 9 illustrates another example of the detection voltage setting circuit 5 .

FIG. 9 is an example of the detection voltage setting circuit 5 . FIG. 9 is different from FIG. 5 in that a resistor R 54 and switches S 51 x ′ and S 52 x ′ are added. The added components are configured and connected as follows. The resistor R 54 is connected in series between the power supply voltage terminal VDD and the resistor R 53 of FIG. 5 . A connection point between the resistors R 53 and R 54 is referred to as “Vn 2 ′”. Each of the switches S 51 x ′ and S 52 x ′ has two terminals, and is controlled to be turned ON or OFF in accordance with a switch control signal (not shown). One terminal of the switch S 51 x ′ is connected to the connection point Vn 2 ′, and the other terminal thereof is connected to the positive terminal of the reference voltage setting circuit Vref 1 . One terminal of the switch S 52 x ′ is connected to the connection point Vn 2 ′, and the other terminal thereof is connected to the positive terminal of the reference voltage setting circuit Vref 2 . The other components than the added ones are connected similarly to FIG. 5 .

The detection voltage setting circuit 5 is connected as described above, and operates as follows.

The voltages at the connection points Vn 1 , Vn 2 , and Vn 2 ′ are the following voltages obtained by dividing VDD and VSS by the resistors R 51 , R 52 , R 53 , and R 54 .

Vn 1= R 51/( R 51+ R 52+ R 53+ R 54)×( VDD−VSS )  (68)

Vn 2=( R 51+ R 52)/( R 51+ R 52+ R 53+ R 54)×( VDD−VSS )  (69)

Vn 2=( R 51+ R 52+ R 53)/( R 51+ R 52+ R 53+ R 54)×( VDD−VSS )  (70)

The voltages Vn 1 , Vn 2 , and Vn 2 ′ may be arbitrarily set by the resistors R 51 , R 52 , R 53 , and R 54 .

The switches S 51 , S 51 x , and S 51 x ′ are controlled so that the switches S 51 x and S 51 x ′ are both turned OFF when the switch S 51 is turned ON and that one of the switches S 51 x and S 51 x ′ is turned ON and the other is turned OFF when the switch S 51 is turned OFF. Similarly, the switches S 52 , S 52 x , and S 52 x ′ are controlled so that the switches S 52 x and S 52 x ′ are both turned OFF when the switch S 52 is turned ON and that one of the switches S 52 x and S 52 x ′ is turned ON and the other is turned OFF when the switch S 52 is turned OFF. The switches S 51 x ′ and S 52 x ′ are provided to provide hysteresis to the detection voltage. In the case where magnetic field intensity is detected by a detection voltage set by the switch S 51 x ′ or S 52 x ′, the switch to be turned ON in the next detection period T is changed from S 51 x ′ to S 51 x or from S 52 x ′ to S 52 x . Similarly, in the case where magnetic field intensity detection is cancelled, the switch to be turned ON in the next detection period T is changed from S 51 x to S 51 x ′ or from S 52 x to S 52 x ′. In this way, chattering generated at the time of magnetic field intensity detection and cancelling may be suppressed.

In conclusion, as described above with reference to FIGS. 5 to 9 , the magnetic sensor device of the present invention may realize arbitrary setting of the magnetic field intensity to be detected by the magnitude and polarity of the voltage component ΔVref supplied from the detection voltage setting circuit 5 to the amplifier 4 , easy discrimination between the S-pole and the N-pole, and easy setting of hysteresis between the detection and cancelling. Those various functions of the circuit configuration in this embodiment are achieved by not only the circuit configuration of the detection voltage setting circuit 5 illustrated in FIGS. 5 and 9 but also the circuit configuration in which, as expressed by Expression (31), the reference voltage component ΔVref may be set by the voltages of the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 in the first phase φ1 and the second phase φ2.

Now, a description is given of an exemplary circuit configuration of the amplifier 4 , which is a component of the magnetic sensor device of FIG. 1 . The function of the amplifier 4 illustrated in the conceptual diagram of FIG. 3 is more specifically realized by a circuit configuration illustrated in FIG. 10 , for example.

FIG. 10 is an exemplary circuit configuration of the amplifier 4 .

The amplifier 4 includes a constant current circuit I 1 , NMOS transistors M 43 , M 44 A, M 44 B, M 45 A, M 46 A, M 45 B, and M 46 B, and PMOS transistors M 41 and M 42 , which are connected as follows. One terminal of the constant current circuit I 1 is connected to the power supply voltage terminal VDD, and the other terminal thereof is connected to a drain and a gate of the NMOS transistor M 43 . This connection point is referred to as “VBN”. VBN is connected to a gate of the NMOS transistor M 44 a and a gate of the NMOS transistor M 44 b . A source of each of the NMOS transistors M 43 , M 44 A, and M 44 B is connected to the ground terminal VSS. A source of each of the NMOS transistors M 45 A and M 46 A is connected to a drain of the NMOS transistor M 44 A. A source of each of the NMOS transistors M 45 B and M 46 B is connected to a drain of the NMOS transistor M 44 B. A drain of each of the NMOS transistors M 45 A and M 45 B is connected to a drain of the PMOS transistor M 41 . This connection point is referred to as “VA”. A drain of each of the NMOS transistors M 46 A and M 46 B is connected to a drain of the PMOS transistor M 42 . This connection point is connected to the output terminal VO of the amplifier 4 . A gate of each of the PMOS transistors M 41 and M 42 is connected to the connection point VA, and a source thereof is connected to the power supply voltage terminal VDD. Gates of the NMOS transistors M 45 A and M 46 A are connected to the second input terminal V 6 and the first input terminal V 5 of the first differential input pair, respectively. Gates of the NMOS transistors M 45 B and M 46 B are connected to the second input terminal V 8 and the first input terminal V 7 of the second differential input pair, respectively.

The amplifier 4 is connected as described above, and operates as follows.

The constant current circuit I 1 generates a constant current to be supplied to the NMOS transistor M 43 . The NMOS transistors M 43 , M 44 A, and M 44 B form a current mirror circuit. A current based on a current flowing between the drain and source of the NMOS transistors M 43 flows between the drain and source of each of the NMOS transistors M 44 A and M 44 B. Five transistors of the NMOS transistors M 44 A, M 45 A, and M 46 A and the PMOS transistors M 41 and M 42 form a differential amplifier, which operates to amplify a gate voltage difference of the NMOS transistors M 45 A and M 46 A that form the first differential input pair, that is, a voltage difference between the second input terminal V 6 of the first differential input pair and the first input terminal V 5 of the first differential input pair, and output the amplified voltage difference to the output terminal VO. This gain is referred to as “A1”. The operations of the current mirror circuit configuration and the differential amplifier configuration are described in detail in the literature of CMOS analog circuits and the like, and the detailed description is herein omitted. Similarly, five transistors of the NMOS transistors M 44 B, M 45 B, and M 46 B and the PMOS transistors M 41 and M 42 form a differential amplifier, which operates to amplify a gate voltage difference between the NMOS transistors M 45 B and M 46 B that form the second differential input pair, that is, a voltage difference between the second input terminal V 8 of the second differential input pair and the first input terminal V 7 of the second differential input pair, and output the amplified voltage difference to the output terminal VO. This gain is referred to as “A2”. The drain of the NMOS transistor M 45 A forming the first differential input pair and the drain of the NMOS transistor M 45 B forming the second differential input pair are connected to the drain of the PMOS transistor M 41 at the connection point VA. The drain of the NMOS transistor M 46 A forming the first differential input pair and the drain of the NMOS transistor M 46 B forming the second differential input pair are connected to the drain of the PMOS transistor M 42 at the output terminal VO. The amplifier 4 operates so that the voltages obtained by amplification by the first differential input pair and the second differential input pair are added at the connection point VA and the output terminal VO. This operation is expressed as follows.

VO=A 1×( V 6− V 5)+ A 2×( V 8− V 7)  (71)

In other words, the operation similar to Expression (14) is performed.

The operation of the magnetic sensor device according to the first embodiment of the present invention has been described above to show that highly-precise magnetic detection may be realized. In the description above, the specific circuit configurations and timing chart for the differential amplifier 3 , the amplifier 4 , and the detection voltage setting circuit 5 according to the first embodiment have been described. However, this embodiment is not necessarily limited to this configuration as long as the operation described above may be performed. For example, the specific configuration of the differential amplifier 3 is not limited to FIG. 2 . As illustrated in FIG. 11 , the resistor R 13 may be removed for direct connection. The gain G of the differential amplifier 3 in this case is expressed by Expression (72) which is modified from the gain expression of Expression (13) with a minimum value of the resistor R 13 .

G =( R 11+ R 12)/ R 12  (72)

This does not deviate from the gist of the present invention, that is, realizing highly-precise magnetic detection.

The above description shows that the detection may be performed while discriminating between the S-pole and the N-pole, but the magnetic sensor device of the present invention is also applicable to bipolar detection (such as motor rotation detection). The bipolar detection is performed by a magnetic sensor device in a manner that the state of detecting only one pole (for example, S-pole) is switched to the state of detecting only the other pole (N-pole) after the one pole is detected.

Further, in the timing chart of FIG. 4 , 6 , 7 , or 8 , there may be employed a driving method in which a standby period of a given period of time is provided between one detection period T and another detection period T to suppress average current consumption of the magnetic sensor device. Also in this case, the effect of highly-precise magnetic detection may be obtained.

Note that, a description has been given above of the magnetic sensor device using a magnetoelectric conversion element. Alternatively, however, the device of the present invention may use, instead of using a magnetoelectric conversion element for outputting a voltage corresponding to magnetic field intensity, a conversion element for similarly outputting a voltage corresponding to an acceleration or a pressure.

Second Embodiment

FIG. 12 is a circuit diagram of a magnetic sensor device according to a second embodiment of the present invention. The differences from the first embodiment illustrated in FIG. 1 reside in that the amplifier 4 is changed to an amplifier 4 b and that the configuration of the detection voltage setting circuit 5 is changed. The differences of the amplifier 4 b from the amplifier 4 reside in that a third differential input pair is added and that a first input terminal V 9 of the third differential input pair and a second input terminal V 10 of the third differential input pair are added. The detection voltage setting circuit 5 is changed so that a reference voltage setting circuit Vref 3 and a reference voltage setting circuit Vref 4 are added. The first input terminal V 9 of the third differential input pair of the amplifier 4 b is connected to a positive terminal of the reference voltage setting circuit Vref 3 , and the second input terminal V 10 of the third differential input pair of the amplifier 4 b is connected to a positive terminal of the reference voltage setting circuit Vref 4 . The other connections and configurations are the same as in the first embodiment.

FIG. 13 is a conceptual diagram illustrating the function of the amplifier 4 b.

The difference of the amplifier 4 b illustrated in FIG. 13 from the amplifier 4 illustrated in FIG. 3 resides in that a differential amplifier 43 and its input terminals V 9 and V 10 are added. The first input terminal V 9 of the third differential input pair of the amplifier 4 b is connected to an inverting input terminal of the differential amplifier 43 , and the second input terminal V 10 of the third differential input pair is connected to a non-inverting input terminal of the differential amplifier 43 . An output of the differential amplifier 43 is connected to an input of an adder 44 . The other connections and configurations are the same as in the amplifier 4 illustrated in FIG. 3 . The differential amplifier 43 operates similarly to the differential amplifiers 41 and 42 . The differential amplifier 43 amplifies a difference of voltages of the two input terminals V 9 and V 10 and inputs the amplified difference to the adder 44 . The adder 44 outputs the sum of the outputs of the differential amplifier 41 and the differential amplifier 42 and the output of the differential amplifier 43 . This amplifier function is expressed by the expression as follows by adding the term of the differential amplifier 43 to Expression (14).

VO=A 1×( V 6− V 5)+ A 2×( V 8− V 7)+ A 3×( V 10− V 9)  (73)

where A3 is the gain of the differential amplifier 43 .

The operation of the magnetic sensor device according to the second embodiment of the present invention is described similarly to that of the operation of the magnetic sensor device according to the first embodiment, and is expressed by the following expression as a result.

The output of the amplifier 4 b in the second phase φ2 is expressed by the following expression similarly to Expression (29).

VOφ 2= A 1×[Δ V sig+( A 2/ A 1)×Δ V ref]  (74)

The signal component ΔVsig is expressed by the following expressions similarly to Expressions (30) and (33).

Δ V sig=−( V 3φ2− V 4φ2)+( V 3φ1− V 4φ1)  (75)

Δ V sig= G ×{( V 1φ2− V 2φ2)−( V 1φ1− V 2φ1)}  (76)

Further, the reference voltage component ΔVref is expressed by the following expression by adding the terms of the reference voltage setting circuit Vref 3 and the reference voltage setting circuit Vref 4 to Expression (31).

Δ V ref=( V ref2φ2− V ref1φ2)−( V ref2φ1− V ref1 φ1)+( V ref4φ2− V ref3φ2)−( V ref4φ1− V ref3φ1)  (77)

The above description of the magnetic sensor device in the first embodiment shows the multifunctionality, such as enabling arbitrary setting of magnetic field intensity to be detected, discrimination between the S-pole and the N-pole, and hysteresis setting between detection and cancellation, by appropriately setting the voltages of the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 in the first phase φ1 and the second phase φ2. However, in this circuit configuration, the voltages of the reference voltage setting circuit Vref 3 and the reference voltage setting circuit Vref 4 can further appropriately set in the first phase φ1 and the second phase φ2, which suggests that a wider magnetic field detection function can be realized. Further, the above description of the magnetic sensor device in the first embodiment shows that, by giving temperature dependence to the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 , temperature dependence of the sensitivity of the Hall element can be corrected to suppress temperature dependency of magnetic field intensity to be detected. However, in this configuration, for example, the reference voltage setting circuit Vref 1 and the reference voltage setting circuit Vref 2 are each configured to output a temperature-independent reference voltage, and the reference voltage setting circuit Vref 3 and the reference voltage setting circuit Vref 4 are each configured to output a temperature-dependent reference voltage. In this manner, the detection magnetic field intensity at the reference temperature can be set by Vref 1 and Vref 2 , and the temperature correction amount from the reference temperature can be set by Vref 3 and Vref 4 .

Now, a description is given of an exemplary circuit configuration of the amplifier 4 b serving as the element constructing the magnetic sensor device of FIG. 12 . More specifically, the function of the amplifier 4 b illustrated in the conceptual diagram of FIG. 13 can be realized by such a circuit configuration as illustrated in FIG. 14 , for example.

FIG. 14 is an exemplary circuit configuration of the amplifier 4 b.

The difference of the amplifier 4 b from the amplifier 4 resides in that NMOS transistors M 44 C, M 45 C, and M 46 C are added. The added elements are connected to the configuration of the amplifier 4 as follows. A source of the NMOS transistor M 44 C is connected to the ground terminal VSS. Sources of the NMOS transistors M 45 C and M 46 C are connected to a drain of the NMOS transistor M 44 C. A drain of the NMOS transistor M 45 C is connected to a connection point VA. A drain of the NMOS transistor M 46 C is connected to the output terminal VO. Gates of the NMOS transistors M 45 C and M 46 C are connected to the second input terminal V 10 and the first input terminal V 9 of the third differential input pair, respectively. The other connections and configurations are the same as in the amplifier 4 illustrated in FIG. 10 .

The amplifier 4 b is connected as describe above, and operates as follows. The additional configurations from the amplifier 4 are now described.

The NMOS transistors M 43 and M 44 C form a current mirror circuit. A current based on the current flowing between the drain and source of the NMOS transistor M 43 flows between the drain and source of the NMOS transistor M 44 C. Five transistors of the NMOS transistors M 44 C, M 45 C, and M 46 C and the PMOS transistors M 41 and M 42 form a differential amplifier, which operates to amplify a gate voltage difference of the NMOS transistors M 45 C and M 46 C of the third differential input pair, that is, a voltage difference between the second input terminal V 10 of the third differential input pair and the first input terminal V 9 of the first differential input pair, and output the amplified voltage difference to the output terminal VO. This gain is referred to as “A3”. The drain of the NMOS transistor M 45 A of the first differential input pair, the drain of the NMOS transistor M 45 B of the second differential input pair, and the drain of the NMOS transistor M 45 C of the third differential input pair are connected to the drain of the PMOS transistor M 41 at the node VA. The drain of the NMOS transistor M 46 A of the first differential input pair, the drain of the NMOS transistor M 46 B of the second differential input pair, and the drain of the NMOS transistor M 46 C of the third differential input pair are connected to the drain of the PMOS transistor M 42 at the output terminal VO. With those connections, the differential amplifier operates so that the voltages amplified by the first differential input pair, the second differential input pair, and the third differential input pair are added at the connection point VA and the output terminal VO. Those operations are expressed as follows.

VO=A 1×( V 6− V 5)+ A 2×( V 8− V 7)+ A 3×( V 10− V 9)  (78)

which shows that the operation similar to Expression (73) is performed.