Fast-settling capacitive-coupled amplifiers

This application is based upon and claims the benefit of priority from Chinese Patent Application No. 201310122194.8 filed Feb. 16, 2013.

1. Field of the Invention

The present invention relates to the field of fast settling amplifiers.

2. Prior Art

In current-sense applications for power management of cell phones, laptops, or electric cars, a small differential-mode (DM) voltage (in the order of mVolts) has to be measured across a current-sense resistor, in the presence of a large common-mode (CM) voltage on the sense resistor on the order of Volts or tens of Volts higher than the amplifier supply voltage. Instrumentation amplifiers (IAs) are frequently used for sensor interfacing. To solve the DM voltage measurement while the CM voltage is beyond the supply-rail, often a part of the input circuitry such as a resistor bridge or a voltage-to-current converter is allowed to draw its CM supply current from the sense resistor. This input CM supply current often has an undesired influence on the system to be measured. Therefore, IAs with capacitive-coupled input choppers have been developed that do not draw input CM supply current, while able to sense beyond their supply-rail voltage.

Capacitive-Coupled Amplifiers with Slow Settling

The prior art capacitive-coupled operational amplifiers and instrumentation amplifiers from the book “Operational Amplifiers, Theory and Design, Second Edition” (2011) of Johan Huijsing, one of the present inventors, do not have a straight frequency characteristic at the chopping frequency. These amplifiers have a slow-settling ripple at the chopping frequency after a step in the input signal. Therefore these amplifiers are not fit for fast-settling signal transfer. Their useful bandwidth lies below the chopper frequency.

The chopper-stabilized amplifiers of FIGS. 1 and 2 have an RC low-pass filter between the input chopper and the chopper behind, which acts as a notch filter in the signal transfer. This means that there is no feedback at the chopping frequency. Therefore an uncontrolled up-turn of the chopper ripple can occur. This causes a slow-settling ripple component after a step.

The chopper amplifiers of FIGS. 3 , 4 and 5 have a ripple reduction loop. This means by definition, there is a notch in the frequency response at the chopper frequency. That also means there is a slow-settling ripple at the clock frequency after a step.

FIG. 1 shows a prior art capacitive-coupled chopper-stabilized operational amplifier.

FIG. 2 shows a prior art capacitor-coupled chopper-stabilized instrumentation amplifier.

FIG. 3 shows a prior art capacitive-coupled chopper amplifier with ripple-reduction loop.

FIG. 4 shows a prior art capacitor-coupled chopper instrumentation amplifier with ripple-reduction loop.

FIG. 5 shows a prior art capacitor-coupled current-feedback chopper instrumentation amplifier with ripple-reduction loop with high-impedance input.

FIG. 6 shows a prior art chopper-stabilized operational amplifier.

FIG. 7 shows a prior art chopper-stabilized chopper operational amplifier.

FIG. 8 shows a capacitive-coupled chopper-stabilized operational amplifier with hybrid-nested Miller compensation and fast settling.

FIG. 9 shows a capacitive-coupled chopper-stabilized operational amplifier with hybrid-nested Miller compensation, fast settling and reduced chopper ripple.

FIG. 10 shows a capacitive-coupled chopper-stabilized instrumentation amplifier with hybrid-nested Miller compensation, fast settling and reduced chopper ripple.

FIG. 11 shows a capacitive-coupled chopper-stabilized chopper operational amplifier with hybrid-nested Miller compensation and fast settling.

FIG. 12 shows a prior art chopper-stabilized chopper operational amplifier with multipath hybrid-nested Miller compensation, auto-zero Gm 5 and Gm 4 .

FIG. 13 shows a capacitive-coupled chopper-stabilized chopper operational amplifier with hybrid-nested Miller compensation, fast settling and reduced chopper ripple.

FIG. 14 shows a capacitive-coupled chopper-stabilized chopper instrumentation amplifier with hybrid-nested Miller compensation, fast settling and reduced chopper ripple.

FIG. 15 illustrates a simplification of the embodiment of FIG. 8 .

FIG. 16 illustrates a simplification of the embodiment of FIG. 11 .

FIG. 17 illustrates a high differential input impedance embodiment of FIG. 14 with hybrid-nested Miller compensation.

FIG. 18 illustrates a simplified embodiment of FIG. 14 .

FIG. 19 illustrates an exemplary circuit for auto-zeroing Gm 5 as may be used in the embodiments of FIGS. 8-11 , 13 and 14 .

The present innovation allows measuring signals in Wide Band applications, meaning approaching the chopper frequency or beyond the chopper frequency without a slow-settling or non-settling chopper ripple response.

For a design of fast-settling capacitive-coupled amplifiers, reference is made to the chopper-stabilized amplifiers of Chapter 10.7 of “Operational Amplifiers, Theory and Design, Second Edition” (Johan Huijsing, 2011), such as the amplifier of FIG. 6 , and to the chopper-stabilized chopper amplifiers of Chapter 10.8, such as in FIG. 7 . These amplifiers have a straight frequency characteristic at the chopping frequency and basically no slow settling chopper ripple for a step function, meaning that their bandwidth is useful at much higher frequencies than the chopping frequency.

From the chopper-stabilized amplifier of FIG. 6 , the two-path capacitive-coupled amplifier of FIG. 8 is derived. It has a high frequency path through Gm 2 that includes the chopper frequency and a low frequency chopper path through Gm 5 . Without showing it to avoid cluttering the Figure, Gm 5 has still to be auto-zeroed or chopper stabilized or given a ripple reduction loop for a low output ripple. A complete Figure for Gm 5 with exemplary auto-zeroing circuit is shown in FIG. 19 . The hybrid-nested Miller compensation scheme with CM 31 and CM 32 takes care of a straight frequency characteristic (see section 6.2 of “Operational Amplifiers, Theory and Design, Second Edition” (Johan Huijsing, 2011) for further information on Compensation techniques, particularly Miller Compensation, Nested Miller Compensation and Hybrid-Nested Miller Compensation). For the same reason, the hybrid-nested Miller compensation is used in the embodiments of FIGS. 9-11 , 13 , 14 and 17 .

On the output side of the coupling capacitors C 21 ,C 22 and C 51 ,C 52 of the embodiment of FIG. 8 , as well as for the corresponding or equivalent coupling capacitors of the other embodiments to be described herein, a common mode reference V Ref is coupled to each side of the differential output of each capacitor pair through a resistor Rb and a pair of diodes connected in parallel with opposite conduction directions. As in the prior art, these diodes limit the magnitude of the voltage on each side of the respective differential signal path to one diode voltage drop above or below the voltage V Ref , and similarly limit the common mode voltage swing to two diode voltage drops. This protects the subsequent amplifiers from a common mode spike on the amplifier input from the sudden connection of the amplifier to a voltage source well out of the amplifier power supply voltage range, and from a high differential voltage caused by the momentary connection of only one line of the differential input to such a voltage. The resistors Rb cause the common mode of the differential voltage output of the respective capacitor pair to settle at the voltage V Ref .

The output ripple due to the offset of Gm 5 can be reduced by introducing a path from the output through the overall feedback to the input that rectifies the ripple and corrects the offset of Gm 5 . Such a path is created by inserting another chopper Ch 3 between the output of the coupling capacitors C 2 and the input of Gm 5 . Chopper Ch 3 rectifies the feedback triangle and square-wave ripple at the input and compensates the offset of Gm 5 . This is shown in FIG. 9 . Again without showing the complete auto-zeroing circuitry in this Figure, Gm 5 still has to be auto-zeroed (or chopper stabilized or given a ripple reduction loop) for a low output ripple, such as the auto-zeroing circuit of FIG. 19 . Without C 21 and C 22 the circuit falls back on FIG. 10.10.1 of the book mentioned earlier, which has a slow settling ripple after an input voltage step.

From FIG. 9 , a capacitive-coupled instrumentation amplifier can be created by adding two more capacitor pairs and an extra chopper Ch 4 as shown in FIG. 10 . Without the feed-forward capacitors C 21 and C 22 , C 31 and C 32 , the circuit falls back on FIG. 10.10.5 of the book with a slow settling ripple. The signal bandwidth must be smaller than the chopping frequency.

From the prior art chopper-stabilized amplifier of FIG. 7 , a two-path capacitive-coupled amplifier of FIG. 11 is derived. It has a low-and-high frequency path through the chopper amplifier stage Gm 2 and a chopper frequency path through Gm 5 . The hybrid-nested Miller compensation scheme with CM 3 takes care of a straight frequency characteristic. Without showing it, Gm 5 has still to be auto-zeroed or chopper stabilized or given a ripple reduction loop for a low output offset and ripple, the auto-zeroing being shown in FIG. 19 . Also the offset of Gm 4 has to be reduced, as shown by example in FIG. 12 . In particular, when switches S 51 and S 52 are set in the position to short the inputs to capacitors C 51 and C 52 together, switches S 53 and S 54 are set in the position to feed back the outputs of Gm 5 to in respective inputs. Since this creates two negative feedback loops, the differential input to Gm 5 will be changed to approximate the input offset of Gm 5 , schematically shown as Vos 5 , so that the differential output of Gm 5 is reduced to approximately the input offset of Gm 5 within an accuracy depending on the gain of Gm 5 . Since the inputs of capacitors C 51 and C 52 are shorted together at this time, the capacitors will be charged to the differential voltage correcting the input offset Vos 5 , which voltage the capacitors will hold as the switches are switched to their other position for normal operation of the signal path until the next auto-zeroing operation. The common mode voltage for the input and output of Gm 5 is set by the common mode voltage Vcm 5 input to Gm 5 . The auto-zeroing for Gm 4 works the same way.

The output ripple due to the offset of Gm 2 in FIG. 11 can be reduced by introducing a path from the output through the overall feedback to the input that rectifies the ripple and corrects the offset of Gm 2 . Such a path is created by inserting another chopper Ch 4 between the output of the coupling capacitors C 5 and the input of Gm 2 . This additional chopper CH 4 shown in FIG. 13 rectifies the feedback ripple at the input and partly compensates the offset of Gm 2 . Without showing it, Gm 5 still has to be auto-zeroed or chopper stabilized or given a ripple reduction loop for a low output offset and ripple, again as shown in FIG. 19 .

From FIG. 13 a capacitive coupled instrumentation amplifier can be created by adding two more capacitor pairs C 31 ,C 32 and C 41 ,C 42 and an extra chopper Ch 5 as shown in FIG. 14 .

FIG. 15 is an embodiment similar to FIG. 8 , but somewhat simplified to better illustrate the basic circuit. Similarly, FIG. 16 is a simplification of the embodiment of FIG. 11 . FIG. 17 illustrates a high differential input impedance embodiment of FIG. 14 , and FIG. 18 shows a simplified embodiment of FIG. 14 . The high differential input impedance of the embodiment of FIG. 17 results from the fact that each input has its own high input impedance amplifier coupled to the input.

It was previously mentioned that in the amplifier systems of FIGS. 8-11 , 13 and 14 , Gm 5 should be and is auto-zeroed. An exemplary circuit for such auto-zeroing of Gm 5 for each of these amplifier systems may be seen in FIG. 19 . In this circuit, when switches S 51 and S 52 are in the position shorting the two switches together, switches S 53 and S 54 are set at their lower positions, coupling the output of Gm 5 to capacitors C 61 and C 62 . With the switches S 51 and S 52 coupled together, the current output of Gm 5 will be Gm 5 *Vos 5 , the gain of Gm 5 times its input offset. Gm 6 will convert the voltage difference on capacitors C 61 and C 62 to provide a current offsetting the output current Gm 5 *Vos 5 of Gm 5 so that capacitors C 61 and C 62 will stop charging. Under this condition, neglecting the input offset of Gm 6 for the moment, the current output of Gm 6 will be equal and opposite to the output of Gm 5 so that the difference in voltage on capacitors C 61 and C 62 will be equal to Gm 5 /Gm 6 times the input offset voltage of Gm 5 . The influence of the input offset of Gm 6 will automatically be reduced by the voltage gain of Gm 6 . The auto-zeroing cycle is repeated periodically to update the charge on C 61 and C 62 . This auto zeroing circuit for Gm 5 may be used in the circuits of FIGS. 8-11 , 13 and 14 .

It was also pointed out that Gm 5 should be auto-zeroed (or chopper stabilized or given a ripple reduction loop), with FIG. 19 providing an exemplary auto-zeroing Gm 5 to avoid unnecessary cluttering of the other Figures. There are numerous auto-zeroing circuits that may be used, such as shown in the foregoing book in the left-hand part of FIG. 10.5.1, FIG. 10.5.2 a , and FIG. 10.5.2 b , with FIG. 19 being the middle one. The capacitors in the left-hand part of FIG. 10.5.1 have the memory function for the offset. Also with respect to Gm 4 , there are many ways the auto-zeroing and integration may co-exist. There is the serial way in which the auto-zeroing amplifier of FIG. 19 itself is actually the integrator amplifier Gm 4 (see FIG. 6 of “A Capacitively Coupled Chopper Instrumentation Amplifier with a ±30V Common-Mode Range, 160 dB CMRR and 5 μV Offset”, Q. Fan et al., 2012 ISSCC IEEE International Solid-State Circuits Conference, and also see FIG. 10.7 of the foregoing book). That is the easiest way. There is also a parallel way in which a separate integrator opamp Gm 4 is monitored by an auto-zeroing circuit. See for instance FIG. 10.8.2 of the foregoing book. Besides auto-zeroing, Gm 4 can instead be chopper stabilized using well known techniques.

In FIGS. 8-11 , 13 and 14 , at least one chopper is used to chop the input signal, which may have a common mode voltage which is well above (or conceivably well below) the amplifier system power supply voltage range. Consequently these choppers may be capacitively coupled to a clock operating within the power supply voltage operating range, if needed. The other choppers in these amplifiers could be capacitively coupled to the clock (see the foregoing article by Q. Fan et al.), but can also be directly driven by the clock as is commonly done.

When the capacitively coupled amplifier systems of the present invention are used as operational amplifiers in a resistive bridge feedback configuration, there will be two resistors, one from each leg of the voltage to be measured to a corresponding input to the amplifier system, and two additional resistors, one from each output of the amplifier system to a respective input (of the opposite polarity) of the amplifier system. In instrumentation amplifier systems in accordance with the present invention (FIGS. 10 and 14 ,), the outputs Vfb− and Vfb+ of the resistor divider R 1 and R 2 are connected back to respective feedback inputs of the opposite polarity for obtaining negative feedback. Note that in FIG. 10 , the clock is shown capacitively coupled to chopper Ch 4 , though such capacitive coupling may not be required, as the feedback voltages are well within the amplifier system power supply voltage range. Also in FIG. 13 , the clock is shown capacitively coupled to chopper Ch 5 , though such capacitive coupling also may not be required, as the feedback of this amplifier system is also within the amplifier system power supply voltage range.

When the operational amplifier versions of the amplifier system of the present invention are used in a resistive bridge feedback configuration, the resistors from each leg of the differential voltage to be measured to a corresponding input to the amplifier system create DC paths from the differential voltage to be measured. If the input CM voltage is large, these DC paths, if not very well balanced, will cause a DC offset on the amplifier system input which could be quite large, possibly exceeding the input signal sought to be measured. Accordingly in many applications, it may be better to use the capacitively coupled instrumentation amplifier system versions, as the feedback for the implementation amplifier systems has an amplifier system power supply voltage so the DC paths are of no significance. However if the operational amplifier systems of the present invention are used as integrators with capacitive feedback, the DC paths will remain as described.

In the foregoing description, reference has been made to amplifiers, with transconductance amplifiers being shown. These amplifiers are exemplary only, as voltage amplifiers may be used if desired. In that regard, the word amplifier as used in the claims to follow is used in the general sense to include amplifiers of both a single stage of amplification and amplifiers of multiple stages of amplification. Also the amplifiers may be realized in CMOS devices as well as bipolar devices or of a mixed transistor construction, though CMOS transistors in transconductance amplifiers are preferred.

Thus while certain preferred embodiments of the present invention have been disclosed and described herein for purposes of illustration and not for purposes of limitation, it will be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention.