Inductance based parallel amplifier phase compensation

This application claims the benefit of U.S. provisional patent application No. 61/551,596, filed Oct. 26, 2011, and U.S. provisional patent application No. 61/562,493, filed Nov. 22, 2011, the disclosures of which are incorporated herein by reference in their entireties.

The present disclosure relates to direct current (DC)-DC converters and circuits that use DC-DC converters.

DC-DC converters often include switching power supplies, which may be based on switching at least one end of an energy storage element, such as an inductor, between a source of DC voltage and a ground. As a result, an output voltage from a DC-DC converter may have a ripple voltage resulting from the switching associated with the energy storage element. Typically, the ripple voltage is undesirable and is minimized as much as sizes and costs permit. Thus, there is a need to minimize ripple voltage using techniques that minimize sizes and costs.

Embodiments of the present disclosure relate to a direct current (DC)-DC converter, which includes a parallel amplifier and a switching supply. The switching supply includes switching circuitry, a first inductive element, and a second inductive element. The parallel amplifier has a feedback input and a parallel amplifier output. The switching circuitry has a switching circuitry output. The first inductive element is coupled between the switching circuitry output and the feedback input. The second inductive element is coupled between the feedback input and the parallel amplifier output.

In one embodiment of the DC-DC converter, the parallel amplifier partially provides a first power supply output signal via the parallel amplifier output based on a voltage setpoint. The switching supply partially provides the first power supply output signal via the first inductive element and the second inductive element. The switching supply may provide power more efficiently than the parallel amplifier. However, the parallel amplifier may provide a voltage of the first power supply output signal more accurately than the switching supply. As such, in one embodiment of the DC-DC converter, the parallel amplifier regulates the voltage of the first power supply output signal based on the voltage setpoint of the first power supply output signal. Further, the switching supply regulates the first power supply output signal to minimize an output current from the parallel amplifier to maximize efficiency. In this regard, the parallel amplifier behaves like a voltage source and the switching supply behaves like a current source.

In one embodiment of the switching supply, a connection node is provided where the first inductive element and the second inductive element are connected to one another. The connection node provides a voltage to the feedback input. Further, in one embodiment of the parallel amplifier, the parallel amplifier has a limited open loop gain at high frequencies above a frequency threshold. At such frequencies, a group delay in the parallel amplifier may normally limit the ability of the parallel amplifier to accurately regulate the voltage of the first power supply output signal. However, by feeding back the voltage from the connection node to the feedback input instead of the voltage from the first power supply output signal, a phase-shift that is developed across the second inductive element at least partially compensates for the limited open loop gain of the parallel amplifier at frequencies above the frequency threshold, thereby improving the ability of the parallel amplifier to accurately regulate the voltage of the first power supply output signal.

Those skilled in the art will appreciate the scope of the disclosure and realize additional aspects thereof after reading the following detailed description in association with the accompanying drawings.

The accompanying drawings incorporated in and forming a part of this specification illustrate several aspects of the disclosure, and together with the description serve to explain the principles of the disclosure.

FIG. 1 shows a direct current (DC)-DC converter according to one embodiment of the present disclosure.

FIG. 2 shows the DC-DC converter according to an alternate embodiment of the DC-DC converter.

FIG. 3 shows a radio frequency (RF) communications system according to one embodiment of the present disclosure.

FIG. 4 shows the RF communications system according to an alternate embodiment of the RF communications system.

FIG. 5 shows the RF communications system according to an additional embodiment of the RF communications system.

FIG. 6 shows the RF communications system according to another embodiment of the RF communications system.

The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the disclosure and illustrate the best mode of practicing the disclosure. Upon reading the following description in light of the accompanying drawings, those skilled in the art will understand the concepts of the disclosure and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims.

FIG. 1 shows a direct current (DC)-DC converter 10 according to one embodiment of the present disclosure. The DC-DC converter 10 includes a switching supply 12 and a parallel amplifier 14 . The switching supply 12 includes switching circuitry 16 , a first inductive element L 1 , and a second inductive element L 2 . The parallel amplifier 14 has a feedback input FBI and a parallel amplifier output PAO. The switching circuitry 16 has a switching circuitry output SCO. The first inductive element L 1 is coupled between the switching circuitry output SCO and the feedback input FBI. The second inductive element L 2 is coupled between the feedback input FBI and the parallel amplifier output PAO.

In one embodiment of the DC-DC converter 10 , the parallel amplifier 14 partially provides a first power supply output signal PS 1 via the parallel amplifier output PAO based on a voltage setpoint. The switching supply 12 partially provides the first power supply output signal PS 1 via the first inductive element L 1 and the second inductive element L 2 . Specifically, the switching supply 12 partially provides the first power supply output signal PS 1 via a series combination of the first inductive element L 1 and the second inductive element L 2 . The switching supply 12 may provide power more efficiently than the parallel amplifier 14 . However, the parallel amplifier 14 may provide a voltage of the first power supply output signal PS 1 more accurately than the switching supply 12 . As such, in one embodiment of the DC-DC converter 10 , the parallel amplifier 14 regulates the voltage, called a first voltage V 1 , of the first power supply output signal PS 1 based on the voltage setpoint of the first power supply output signal PS 1 . Further, the switching supply 12 regulates the first power supply output signal PS 1 to minimize an output current, called a parallel amplifier output current IP, from the parallel amplifier 14 to maximize efficiency. In this regard, the parallel amplifier 14 behaves like a voltage source and the switching supply 12 behaves like a current source. Additionally, the switching circuitry 16 provides a switching output voltage VS and an inductor current IL to the first inductive element L 1 via the switching circuitry output SCO.

In one embodiment of the switching supply 12 , a connection node 18 is provided where the first inductive element L 1 and the second inductive element L 2 are connected to one another. The connection node 18 provides a second voltage V 2 to the parallel amplifier 14 via the feedback input FBI. Further, in one embodiment of the parallel amplifier 14 , the parallel amplifier 14 has a limited open loop gain at high frequencies that are above a frequency threshold. At such frequencies, a group delay in the parallel amplifier 14 may normally limit the ability of the parallel amplifier 14 to accurately regulate the first voltage V 1 of the first power supply output signal PS 1 . However, by feeding back the second voltage V 2 to the feedback input FBI instead of the first voltage V 1 , a phase-shift that is developed across the second inductive element L 2 at least partially compensates for the limited open loop gain of the parallel amplifier 14 at frequencies that are above the frequency threshold, thereby improving the ability of the parallel amplifier 14 to accurately regulate the first voltage V 1 . In this regard, in one embodiment of the DC-DC converter 10 , the parallel amplifier 14 partially provides the first power supply output signal PS 1 via the parallel amplifier output PAO based on the voltage setpoint and feeding back a voltage to the feedback input FBI from the connection node 18 between the first inductive element L 1 and the second inductive element L 2 .

In one embodiment of the DC-DC converter 10 , the DC-DC converter 10 receives a DC source signal VDC, such that the parallel amplifier 14 partially provides the first power supply output signal PS 1 using the DC source signal VDC and the switching supply 12 partially provides the first power supply output signal PS 1 using the DC source signal VDC.

FIG. 2 shows the DC-DC converter 10 according to an alternate embodiment of the DC-DC converter 10 . The DC-DC converter 10 illustrated in FIG. 2 is similar to the DC-DC converter 10 illustrated in FIG. 1 , except the DC-DC converter 10 illustrated in FIG. 2 further includes power supply control circuitry 20 and an offset capacitive element CO. Additionally, the switching supply 12 further includes a filter capacitive element CF. The power supply control circuitry 20 receives the DC source signal VDC and is coupled to the parallel amplifier 14 and the switching circuitry 16 . The first inductive element L 1 and the second inductive element L 2 provide a second power supply output signal PS 2 via the connection node 18 .

The offset capacitive element CO is coupled between the parallel amplifier output PAO and the second inductive element L 2 , such that the parallel amplifier 14 partially provides the first power supply output signal PS 1 via the parallel amplifier output PAO and the offset capacitive element CO based on the voltage setpoint. The offset capacitive element CO allows the first voltage V 1 to be higher than a voltage at the parallel amplifier output PAO. As a result, the parallel amplifier 14 may properly regulate the first voltage V 1 even if the first voltage V 1 is greater than a maximum output voltage from the parallel amplifier 14 at the parallel amplifier output PAO.

In general, the filter capacitive element CF is coupled between the parallel amplifier output PAO and a ground. In the embodiment of the DC-DC converter 10 illustrated in FIG. 2 , the filter capacitive element CF is coupled between the parallel amplifier output PAO and the ground through the offset capacitive element CO. In an alternate embodiment of the DC-DC converter 10 , the offset capacitive element CO is omitted, such that the filter capacitive element CF is directly coupled between the parallel amplifier output PAO and the ground. The first inductive element L 1 , the second inductive element L 2 , and the filter capacitive element CF form a first low-pass filter 22 having a first cutoff frequency. The second inductive element L 2 and the filter capacitive element CF form a second low-pass filter 24 having a second cutoff frequency. The second cutoff frequency may be significantly higher than the first cutoff frequency. As such, the first low-pass filter 22 may be used primarily to filter the switching output voltage VS, which is typically a square wave. However, the second low-pass filter 24 may be used to target specific high frequencies, such as certain harmonics of the switching output voltage VS.

In a first embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is at least 10 times greater than the first cutoff frequency. In a second embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is at least 100 times greater than the first cutoff frequency. In a third embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is at least 500 times greater than the first cutoff frequency. In a fourth embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is at least 1000 times greater than the first cutoff frequency. In a fifth embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is less than 1000 times greater than the first cutoff frequency. In a sixth embodiment of the first low-pass filter 22 and the second low-pass filter 24 , the second cutoff frequency is less than 5000 times greater than the first cutoff frequency.

The first inductive element L 1 has a first inductance and the second inductive element L 2 has a second inductance. In a first embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is at least 10 times greater than a magnitude of the second inductance. In a second embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is at least 100 times greater than a magnitude of the second inductance. In a third embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is at least 500 times greater than a magnitude of the second inductance. In a fourth embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is at least 1000 times greater than a magnitude of the second inductance. In a fifth embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is less than 1000 times greater than a magnitude of the second inductance. In a sixth embodiment of the first inductive element L 1 and the second inductive element L 2 , a magnitude of the first inductance is less than 5000 times greater than a magnitude of the second inductance.

FIG. 3 shows a radio frequency (RF) communications system 26 according to one embodiment of the present disclosure. The RF communications system 26 includes RF transmitter circuitry 28 , RF system control circuitry 30 , RF front-end circuitry 32 , an RF antenna 34 , and a DC power source 36 . The RF transmitter circuitry 28 includes transmitter control circuitry 38 , an RF power amplifier (PA) 40 , the DC-DC converter 10 , and PA bias circuitry 42 . The DC-DC converter 10 functions as an envelope tracking power supply. In an alternate embodiment of the RF communications system 26 , the DC power source 36 is external to the RF communications system 26 .

In one embodiment of the RF communications system 26 , the RF front-end circuitry 32 receives via the RF antenna 34 , processes, and forwards an RF receive signal RFR to the RF system control circuitry 30 . The RF system control circuitry 30 provides a power supply control signal VRMP and a transmitter configuration signal PACS to the transmitter control circuitry 38 . The RF system control circuitry 30 provides an RF input signal RFI to the RF PA 40 . The DC power source 36 provides a DC source signal VDC to the DC-DC converter 10 . In one embodiment of the DC power source 36 , the DC power source 36 is a battery. In one embodiment of the power supply control signal VRMP, the power supply control signal VRMP is an envelope power supply control signal.

The transmitter control circuitry 38 is coupled to the DC-DC converter 10 and to the PA bias circuitry 42 . The DC-DC converter 10 provides the first power supply output signal PS 1 to the RF PA 40 based on the power supply control signal VRMP. As such, the first power supply output signal PS 1 is a first envelope power supply signal. The DC source signal VDC provides power to the DC-DC converter 10 . As such, the first power supply output signal PS 1 is based on the DC source signal VDC. The power supply control signal VRMP is representative of the voltage setpoint of the first power supply output signal PS 1 . As such, the voltage setpoint is based on the power supply control signal VRMP. The RF PA 40 receives and amplifies the RF input signal RFI to provide an RF transmit signal RFT using the first envelope power supply signal, which is the first power supply output signal PS 1 . The first envelope power supply signal provides power for amplification to the RF PA 40 . The RF front-end circuitry 32 receives, processes, and transmits the RF transmit signal RFT via the RF antenna 34 . In one embodiment of the RF transmitter circuitry 28 , the transmitter control circuitry 38 configures the RF transmitter circuitry 28 based on the transmitter configuration signal PACS.

The PA bias circuitry 42 provides a PA bias signal PAB to the RF PA 40 . In this regard, the PA bias circuitry 42 biases the RF PA 40 via the PA bias signal PAB. In one embodiment of the PA bias circuitry 42 , the PA bias circuitry 42 biases the RF PA 40 based on the transmitter configuration signal PACS. In one embodiment of the RF front-end circuitry 32 , the RF front-end circuitry 32 includes at least one RF switch, at least one RF amplifier, at least one RF filter, at least one RF duplexer, at least one RF diplexer, at least one RF amplifier, the like, or any combination thereof. In one embodiment of the RF system control circuitry 30 , the RF system control circuitry 30 is RF transceiver circuitry, which may include an RF transceiver IC, baseband controller circuitry, the like, or any combination thereof. In one embodiment of the RF transmitter circuitry 28 , the first envelope power supply signal provides power for amplification and envelope tracks the RF transmit signal RFT.

FIG. 4 shows the RF communications system 26 according to an alternate embodiment of the RF communications system 26 . The RF communications system 26 illustrated in FIG. 4 is similar to the RF communications system 26 illustrated in FIG. 3 , except in the RF communications system 26 illustrated in FIG. 4 , the RF transmitter circuitry 28 further includes a digital communications interface 44 , which is coupled between the transmitter control circuitry 38 and a digital communications bus 46 . The digital communications bus 46 is also coupled to the RF system control circuitry 30 . As such, the RF system control circuitry 30 provides the power supply control signal VRMP ( FIG. 3 ) and the transmitter configuration signal PACS ( FIG. 3 ) to the transmitter control circuitry 38 via the digital communications bus 46 and the digital communications interface 44 .

FIG. 5 shows details of the DC-DC converter 10 illustrated in FIG. 3 according to one embodiment of the DC-DC converter 10 . The DC-DC converter 10 includes the power supply control circuitry 20 , the parallel amplifier 14 , and the switching supply 12 . The power supply control circuitry 20 controls the parallel amplifier 14 and the switching supply 12 . The parallel amplifier 14 and the switching supply 12 provide the first power supply output signal PS 1 , such that the parallel amplifier 14 partially provides the first power supply output signal PS 1 and the switching supply 12 partially provides the first power supply output signal PS 1 .

FIG. 6 shows the RF communications system 26 according to another embodiment of the RF communications system 26 . The RF communications system 26 illustrated in FIG. 6 is similar to the RF communications system 26 illustrated in FIG. 3 , except in the RF communications system 26 illustrated in FIG. 6 , the PA bias circuitry 42 ( FIG. 3 ) is omitted and the RF PA 40 includes a driver stage 48 and a final stage 50 , which is coupled to the driver stage 48 . The DC-DC converter 10 provides the second power supply output signal PS 2 , which is a second envelope power supply signal, to the driver stage 48 based on the power supply control signal VRMP. Further, the DC-DC converter 10 provides the first power supply output signal PS 1 , which is the first envelope power supply signal, to the final stage 50 based on the power supply control signal VRMP. The driver stage 48 receives and amplifies the RF input signal RFI to provide a driver stage output signal DSO using the second envelope power supply signal, which provides power for amplification. Similarly, the final stage 50 receives and amplifies the driver stage output signal DSO to provide the RF transmit signal RFT using the first envelope power supply signal, which provides power for amplification.

An analysis of improved ripple cancellation performance of the DC-DC converter 10 illustrated in FIG. 2 is presented. In general, the first power supply output signal PS 1 is fed to a load (not shown) having a load resistance RL, such as the RF PA 40 ( FIG. 3 ). The switching output voltage VS has a DC component called a DC voltage VD and a ripple component called an AC voltage VA given by EQ. 1, as shown below.

VS=VD+VA.   EQ. 1:

Further, the inductor current IL has a DC current ID and an AC current IA given by EQ. 2, as shown below.

IL=ID+IA.   EQ. 2:

The DC-DC converter 10 regulates the DC voltage VD to be about equal to the voltage setpoint. The first inductive element L 1 and the second inductive element L 2 appear approximately as short circuits to the DC component. Further, the filter capacitive element CF appears approximately as an open circuit to the DC component. Therefore, the DC voltage VD is approximately applied to the load resistance RL, as intended. As a result, the DC current ID is based on the DC voltage VD and the load resistance RL, as shown in EQ. 3 below.

ID=VD/RL.   EQ. 3:

Most of the ripple components of the switching output voltage VS is filtered out from the first voltage V 1 by the first low-pass filter 22 and the second low-pass filter 24 . As a result, most of the AC voltage VA is across the series combination of the first inductive element L 1 and the second inductive element L 2 . The first inductive element L 1 has a first inductance 11 and the second inductive element L 2 has a second inductance I 2 . Therefore, the AC current IA is based on the AC voltage VA, the first inductance 11 and the second inductance I 2 , where s=j2πf, j=√−1, and f=frequency, as shown in EQ. 4 below.

IA=VA/[s ( I 1+ I 2)].  EQ. 4:

Much of what remains of the ripple component is cancelled out from the first voltage V 1 by the parallel amplifier 14 . However, to the extent that the parallel amplifier 14 cannot completely cancel out the remains of the ripple component, the first voltage V 1 has a first residual ripple voltage VR 1 and the second voltage V 2 has a second residual ripple voltage VR 2 . Two approaches to ripple cancellation will be compared against one another. In the first approach, the DC-DC converter 10 is the DC-DC converter 10 illustrated in FIG. 2 , such that the second voltage V 2 is fed to the feedback input FBI, as shown. In this regard, the second residual ripple voltage VR 2 drives the parallel amplifier 14 to provide a ripple cancellation current, which is the parallel amplifier output current IP. In the second approach, the DC-DC converter 10 is similar to the DC-DC converter 10 illustrated in FIG. 2 , except the first voltage V 1 is fed to the feedback input FBI instead of the second voltage V 2 , such that the first residual ripple voltage VR 1 drives the parallel amplifier 14 to provide the ripple cancellation current, which is the parallel amplifier output current IP.

In the following analysis, the parallel amplifier 14 has a DC open loop gain GO and an open loop bandwidth factor T. As a result, the parallel amplifier 14 has a gain G, as shown in EQ. 5 below.

G=GO /(1 +sT ).  EQ. 5:

As a result, at frequencies significantly below an open loop bandwidth of the parallel amplifier 14 , the open loop bandwidth factor T is small compared to one, such that the gain G approaches the DC open loop gain GO. Conversely, at frequencies significantly above the open loop bandwidth of the parallel amplifier 14 , the open loop bandwidth factor T is large compared to one, such that the gain G approaches GO/sT.

In the first approach, described above wherein the second residual ripple voltage VR 2 drives the parallel amplifier 14 and at frequencies significantly above the open loop bandwidth of the parallel amplifier 14 , the parallel amplifier output current IP is based on the second residual ripple voltage VR 2 , as shown in EQ. 6 below.

IP=G*VR 2≈( GO*VR 2)/ sT.   EQ. 6:

In the second approach described above, when the first residual ripple voltage VR 1 drives the parallel amplifier 14 and at frequencies significantly above the open loop bandwidth of the parallel amplifier 14 , the parallel amplifier output current IP is based on the first residual ripple voltage VR 1 , as shown in EQ. 7 below.

IP=G*VR 1≈( GO*VR 1)/ sT.   EQ. 7:

However, a difference between the first residual ripple voltage VR 1 and the second residual ripple voltage VR 2 is based on the AC current IA and the second inductance I 2 , as shown in EQ. 8 and EQ. 9 below.

( VR 2 −VR 1)=( s )( IA )( I 2),  EQ. 8: or

VR 2=( s )( IA )( I 2)+ VR 1.  EQ. 9:

Substituting EQ. 9 into EQ. 6 provides EQ. 10 and EQ. 11, as shown below.

IP ≈( GO )( VR 1)/ sT +( GO )( s )( IA )( I 2)/ sT,   EQ. 10: or

IP ≈( GO )( VR 1)/ sT +( GO )( IA )( I 2)/ T.   EQ. 11:

EQ. 11 is representative of the first approach and EQ. 7 is representative of the second approach. As a reminder, in the first approach, the second residual ripple voltage VR 2 drives the parallel amplifier 14 and in the second approach, the first residual ripple voltage VR 1 drives the parallel amplifier 14 . In both equations, a smaller first residual ripple voltage VR 1 represents better ripple cancellation performance. For comparison purposes, both approaches are assumed to provide the same magnitude of parallel amplifier output current IP. However, in the second approach, the parallel amplifier output current IP is phase-shifted from the first residual ripple voltage VR 1 by about 90 degrees. As such, the parallel amplifier output current IP is phase-shifted from the ripple current it is trying to cancel by about 90 degrees, thereby degrading ripple cancellation performance. However, in the first approach, according to EQ. 11, the parallel amplifier output current IP has two terms, namely the (GO)(VR 1 )/sT term and the (GO)(IA)(I 2 )/T term. The (GO)(VR 1 )/sT term has the same phase-alignment shortcoming as in the second approach. But the (GO)(IA)(I 2 )/T term phase-aligns the parallel amplifier output current IP with the ripple current it is trying to cancel. Overall, the phase-alignment in the first approach is improved over the second approach. Additionally, to the extent that the (GO)(VR 1 )/sT term is smaller than the (GO)(IA)(I 2 )/T term, the first residual ripple voltage VR 1 is reduced, thereby improving ripple cancellation. In this regard, if the (GO)(IA)(I 2 )/T term is equal to the (GO)(VR 1 )/sT term in EQ. 7, then in the (GO)(VR 1 )/sT term in EQ. 11, the first residual ripple voltage VR 1 is equal to about zero, such that the first approach is greatly improved over the second approach.

Some of the circuitry previously described may use discrete circuitry, integrated circuitry, programmable circuitry, non-volatile circuitry, volatile circuitry, software executing instructions on computing hardware, firmware executing instructions on computing hardware, the like, or any combination thereof. The computing hardware may include mainframes, micro-processors, micro-controllers, DSPs, the like, or any combination thereof.

None of the embodiments of the present disclosure are intended to limit the scope of any other embodiment of the present disclosure. Any or all of any embodiment of the present disclosure may be combined with any or all of any other embodiment of the present disclosure to create new embodiments of the present disclosure.

Those skilled in the art will recognize improvements and modifications to the embodiments of the present disclosure. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.