Low noise amplifiers for carrier aggregation

CLAIM OF PRIORITY UNDER 35 U.S.C. §119

The present Application for Patent claims priority to Provisional U.S. Application Ser. No. 61/652,064, entitled “LOW NOISE AMPLIFIERS FOR CARRIER AGGREGATION,” filed May 25, 2012, assigned to the assignee hereof, and expressly incorporated herein by reference.

I. Field

The present disclosure relates generally to electronics, and more specifically to low noise amplifiers (LNAs).

II. Background

A wireless device (e.g., a cellular phone or a smartphone) in a wireless communication system may transmit and receive data for two-way communication. The wireless device may include a transmitter for data transmission and a receiver for data reception. For data transmission, the transmitter may modulate a radio frequency (RF) carrier signal with data to obtain a modulated RF signal, amplify the modulated RF signal to obtain an amplified RF signal having the proper output power level, and transmit the amplified RF signal via an antenna to a base station. For data reception, the receiver may obtain a received RF signal via the antenna and may amplify and process the received RF signal to recover data sent by the base station.

A wireless device may support carrier aggregation, which is simultaneous operation on multiple carriers. A carrier may refer to a range of frequencies used for communication and may be associated with certain characteristics. For example, a carrier may be associated with system information describing operation on the carrier. A carrier may also be referred to as a component carrier (CC), a frequency channel, a cell, etc. It is desirable to efficiently support carrier aggregation by the wireless device.

FIG. 1 shows a wireless device communicating with a wireless system.

FIGS. 2A to 2D show four examples of carrier aggregation (CA).

FIG. 3 shows a block diagram of the wireless device in FIG. 1 .

FIGS. 4A and 4B show a receiver supporting intra-band CA.

FIGS. 5A and 5B show a receiver supporting intra-band CA and inter-band CA.

FIGS. 6A to 6C show an LNA with inductive degeneration and cascode shutoff.

FIG. 7 shows an LNA with inductive degeneration, cascode shutoff, and resistive feedback.

FIG. 8A shows an LNA with a separate input attenuation circuit for each amplifier stage.

FIG. 8B shows an LNA with a shared input attenuation circuit for two amplifier stages.

FIG. 9 shows an LNA with a tunable input matching circuit.

FIGS. 10 to 11C show several exemplary designs of a multiple-input multiple-output (MIMO) LNA.

FIGS. 12A to 12F show six exemplary designs of a tunable input matching circuit.

FIG. 13 shows a process for receiving signals in a wireless system.

The detailed description set forth below is intended as a description of exemplary designs of the present disclosure and is not intended to represent the only designs in which the present disclosure can be practiced. The term “exemplary” is used herein to mean “serving as an example, instance, or illustration.” Any design described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other designs. The detailed description includes specific details for the purpose of providing a thorough understanding of the exemplary designs of the present disclosure. It will be apparent to those skilled in the art that the exemplary designs described herein may be practiced without these specific details. In some instances, well-known structures and devices are shown in block diagram form in order to avoid obscuring the novelty of the exemplary designs presented herein.

LNAs supporting carrier aggregation are disclosed herein. These LNAs may have better performance and may be used for various types of electronic devices such as wireless communication devices.

FIG. 1 shows a wireless device 110 communicating with a wireless communication system 120 . Wireless system 120 may be a Long Term Evolution (LTE) system, a Code Division Multiple Access (CDMA) system, a Global System for Mobile Communications (GSM) system, a wireless local area network (WLAN) system, or some other wireless system. A CDMA system may implement Wideband CDMA (WCDMA), cdma2000, or some other version of CDMA. For simplicity, FIG. 1 shows wireless system 120 including two base stations 130 and 132 and one system controller 140 . In general, a wireless system may include any number of base stations and any set of network entities.

Wireless device 110 may also be referred to as a user equipment (UE), a mobile station, a terminal, an access terminal, a subscriber unit, a station, etc. Wireless device 110 may be a cellular phone, a smartphone, a tablet, a wireless modem, a personal digital assistant (PDA), a handheld device, a laptop computer, a smartbook, a netbook, a cordless phone, a wireless local loop (WLL) station, a Bluetooth device, etc. Wireless device 110 may be capable of communicating with wireless system 120 . Wireless device 110 may also be capable of receiving signals from broadcast stations (e.g., a broadcast station 134 ), signals from satellites (e.g., a satellite 150 ) in one or more global navigation satellite systems (GNSS), etc. Wireless device 110 may support one or more radio technologies for wireless communication such as LTE, cdma2000, WCDMA, GSM, 802.11, etc.

Wireless device 110 may support carrier aggregation, which is operation on multiple carriers. Carrier aggregation may also be referred to as multi-carrier operation. Wireless device 110 may be able to operate in low-band from 698 to 960 megahertz (MHz), mid-band from 1475 to 2170 MHz, and/or high-band from 2300 to 2690 and 3400 to 3800 MHz. Low-band, mid-band, and high-band refer to three groups of bands (or band groups), with each band group including a number of frequency bands (or simply, “bands”). Each band may cover up to 200 MHz and may include one or more carriers. Each carrier may cover up to 20 MHz in LTE. LTE Release 11 supports 35 bands, which are referred to as LTE/UMTS bands and are listed in 3GPP TS 36.101. Wireless device 110 may be configured with up to 5 carriers in one or two bands in LTE Release 11.

In general, carrier aggregation (CA) may be categorized into two types—intra-band CA and inter-band CA. Intra-band CA refers to operation on multiple carriers within the same band. Inter-band CA refers to operation on multiple carriers in different bands.

FIG. 2A shows an example of contiguous intra-band CA. In the example shown in FIG. 2A , wireless device 110 is configured with four contiguous carriers in the same band, which is a band in low-band. Wireless device 110 may receive transmissions on multiple contiguous carriers within the same band.

FIG. 2B shows an example of non-contiguous intra-band CA. In the example shown in FIG. 2B , wireless device 110 is configured with four non-contiguous carriers in the same band, which is a band in low-band. The carriers may be separated by 5 MHz, 10 MHz, or some other amount. Wireless device 110 may receive transmissions on multiple non-contiguous carriers within the same band.

FIG. 2C shows an example of inter-band CA in the same band group. In the example shown in FIG. 2C , wireless device 110 is configured with four carriers in two bands in the same band group, which is low-band. Wireless device 110 may receive transmissions on multiple carriers in different bands in the same band group (e.g., low-band in FIG. 2C ).

FIG. 2D shows an example of inter-band CA in different band groups. In the example shown in FIG. 2D , wireless device 110 is configured with four carriers in two bands in different band groups, which include two carriers in one band in low-band and two additional carriers in another band in mid-band. Wireless device 110 may receive transmissions on multiple carriers in different bands in different band groups (e.g., low-band and mid-band in FIG. 2D ).

FIGS. 2A to 2D show four examples of carrier aggregation. Carrier aggregation may also be supported for other combinations of bands and band groups. For example, carrier aggregation may be supported for low-band and high-band, mid-band and high-band, high-band and high-band, etc.

FIG. 3 shows a block diagram of an exemplary design of wireless device 110 in FIG. 1 . In this exemplary design, wireless device 110 includes a transceiver 320 coupled to a primary antenna 310 , receivers 322 coupled to a secondary antenna 312 , and a data processor/controller 380 . Transceiver 320 includes multiple (K) receivers 330 aa to 330 ak and multiple (K) transmitters 360 a to 360 k to support multiple bands, carrier aggregation, multiple radio technologies, etc. Receivers 322 include multiple (M) receivers 330 ba to 330 bm to support multiple bands, carrier aggregation, multiple radio technologies, receive diversity, MIMO transmission, etc.

In the exemplary design shown in FIG. 3 , each receiver 330 includes input circuits 332 , an LNA 340 , and receive circuits 342 . For data reception, antenna 310 receives signals from base stations and/or other transmitter stations and provides a received RF signal, which is routed through switches/duplexers 324 and provided to a selected receiver. The description below assumes that receiver 330 aa is the selected receiver. Within receiver 330 aa , the received RF signal is passed through input circuits 332 aa , which provides an input RF signal to an LNA 340 aa . Input circuits 332 aa may include a matching circuit, a receive filter, etc. LNA 340 aa amplifies the input RF signal and provides an output RF signal. Receive circuits 342 aa amplify, filter, and downconvert the output RF signal from RF to baseband and provide an analog input signal to data processor 380 . Receive circuits 332 aa may include mixers, a filter, an amplifier, a matching circuit, an oscillator, a local oscillator (LO) generator, a phase locked loop (PLL), etc. Each remaining receiver 330 in transceiver 320 and each receiver 330 in receivers 322 may operate in similar manner as receiver 330 aa in transceiver 320 .

In the exemplary design shown in FIG. 3 , each transmitter 360 includes transmit circuits 362 , a power amplifier (PA) 364 , and output circuits 366 . For data transmission, data processor 380 processes (e.g., encodes and modulates) data to be transmitted and provides an analog output signal to a selected transmitter. The description below assumes that transmitter 360 a is the selected transmitter. Within transmitter 360 a , transmit circuits 362 a amplify, filter, and upconvert the analog output signal from baseband to RF and provide a modulated RF signal. Transmit circuits 362 a may include mixers, an amplifier, a filter, a matching circuit, an oscillator, an LO generator, a PLL, etc. A PA 364 a receives and amplifies the modulated RF signal and provides an amplified RF signal having the proper output power level. The amplified RF signal is passed through output circuits 366 a , routed through switches/duplexers 324 , and transmitted via antenna 310 . Output circuits 366 a may include a matching circuit, a transmit filter, a directional coupler, etc.

FIG. 3 shows an exemplary design of receivers 330 and transmitters 360 . A receiver and a transmitter may also include other circuits not shown in FIG. 3 , such as filters, matching circuits, etc. All or a portion of transceiver 320 and receivers 322 may be implemented on one or more analog integrated circuits (ICs), RF ICs (RFICs), mixed-signal ICs, etc. For example, LNAs 340 , receive circuits 342 , and transmit circuits 362 may be implemented on one module, which may be an RFIC, etc. Switches/duplexers 324 , switches/filters 326 , input circuits 332 , output circuits 366 , and PAs 364 may be implemented on another module, which may be a hybrid module, etc. The circuits in receivers 330 and transmitters 360 may also be implemented in other manners.

Data processor/controller 380 may perform various functions for wireless device 110 . For example, data processor 380 may perform processing for data being received via receivers 330 and data being transmitted via transmitters 360 . Controller 380 may control the operation of switches/duplexers 324 , switches/filters 326 , input circuits 332 , LNAs 340 , receive circuits 342 , transmit circuits 362 , PAs 364 , output circuits 366 , or a combination thereof. A memory 382 may store program codes and data for data processor/controller 380 . Data processor/controller 380 may be implemented on one or more application specific integrated circuits (ASICs) and/or other ICs.

Wireless device 110 may receive multiple transmissions from one or more cells/base stations on multiple carriers at different frequencies for carrier aggregation. For intra-band CA, the multiple transmissions are sent on multiple carriers in the same band. For inter-band CA, the multiple transmissions are sent on multiple carriers in different bands.

FIG. 4A shows a block diagram of an exemplary design of a receiver 400 that includes a CA LNA 440 supporting no CA and intra-band CA. CA LNA 440 may be used for one or more LNAs 340 within wireless device 110 in FIG. 3 .

At receiver 400 , an antenna 410 receives transmissions on multiple carriers in the same band and provides a received RF signal. The received RF signal is routed through switches/duplexers 424 and provided as a receiver input signal, RXin, to an input matching circuit 432 . Matching circuit 432 performs power and/or impedance matching between CA LNA 440 and either switches/duplexers 424 or antenna 410 for one or more bands of interest. Matching circuit 432 , which may be part of one of input circuits 332 in FIG. 3 , provides an input RF signal, RFin, to CA LNA 440 .

CA LNA 440 receives the input RF signal from matching circuit 432 , amplifies the input RF signal, and provides up to M output RF signals, RFout 1 to RFoutM, via up to M LNA outputs, where M>1. M load circuits 490 a to 490 m are coupled to the M LNA outputs. Each load circuit 490 may include one or more inductors, capacitors, transistors, mixers, etc. Each load circuit 490 may be part of one of receive circuits 342 in FIG. 3 . Each output RF signal may be provided to one or more mixers within one load circuit 490 and may be downconverted by the associated mixer(s) such that transmissions on one or more carriers of interest are downconverted from RF to baseband.

A CA LNA, such as CA LNA 440 in FIG. 4A , may operate in a non-CA mode or a CA mode at any given moment. In the non-CA mode, the CA LNA operates in a 1-input 1-output (1×1) configuration, receives one input RF signal comprising one or more transmissions on one set of carriers, and provides one output RF signal to one load circuit. In the CA mode, the CA LNA operates in a 1×M configuration, receives one input RF signal comprising multiple transmissions on M sets of carriers, and provides M output RF signals to M load circuits, one output RF signal for each set of carriers, where M>1. Each set of carriers may include one or more carriers in one band.

FIG. 4B shows a schematic diagram of an exemplary design of a CA LNA 440 x supporting no CA and intra-band CA on two sets of carriers in the same band. CA LNA 440 x is one exemplary design of CA LNA 440 in FIG. 4A . In the exemplary design shown in FIG. 4B , CA LNA 440 x receives an input RF signal from input matching circuit 432 and provides up to two output RF signals, RFout 1 and RFout 2 , for up to two sets of carriers. The first output RF signal is provided to a load circuit 490 x , and the second output RF signal is provided to a load circuit 490 y.

In the exemplary design shown in FIG. 4B , load circuit 490 x includes two mixers 492 a and 492 b coupled to two baseband filters 494 a and 494 b , respectively. Mixers 492 a and 492 b implement a quadrature downconverter for a first set of carriers. Mixer 492 a receives the first output RF signal from CA LNA 440 x and an inphase LO signal, ILO 1 , at a first mixing frequency for the first set of carriers. Mixer 492 a downconverts the first output RF signal with the ILO 1 signal and provides an inphase (I) downconverted signal. Mixer 492 b receives the first output RF signal from CA LNA 440 x and a quadrature LO signal, QLO 1 , at the first mixing frequency for the first set of carriers. Mixer 492 b downconverts the first output RF signal with the QLO 1 signal and provides a quadrature (Q) downconverted signal. Filters 494 a and 494 b receive and filter the I and Q downconverted signals from mixers 492 a and 492 b , respectively, and provide I and Q baseband signals, Vout 1 , for the first set of carriers.

Mixers 492 c and 492 d and filters 494 c and 494 d within load circuit 490 y similarly process the second output RF signal from CA LNA 440 x and provide I and Q baseband signals for a second set of carriers. Mixers 492 c and 492 d receive the second RF signal and I and Q LO signals, respectively, at a second mixing frequency for the second set of carriers. Mixers 492 c and 492 d downconvert the second output RF signal with the I and Q LO signals and provide the I and Q downconverted signals, respectively. Filters 494 c and 494 d receive and filter the I and Q downconverted signals from mixers 492 c and 492 d , respectively, and provide I and Q baseband signals, Vout 2 , for the second set of carriers.

FIG. 4B shows an exemplary design of load circuits 490 x and 490 y . A load circuit may also comprise different and/or additional circuits. For example, a load circuit may include an amplifier coupled before the mixers, or between the mixers and the filters, or after the filters.

FIG. 5A shows a block diagram of an exemplary design of a receiver 500 that includes a MIMO LNA 540 supporting no CA, intra-band CA, and inter-band CA. MIMO LNA 540 may be used for one or more LNAs 340 within wireless device 110 in FIG. 3 .

At receiver 500 , an antenna 510 receives transmissions on one or more carriers in the same band or different bands and provides a received RF signal to switches/duplexers 524 . Switches/duplexers 524 provide up to N receiver input signals, RXin 1 to RXinN, to up to N input matching circuits 532 a to 532 n , respectively, where N>1. Matching circuits 532 a to 532 n may be part of one or more input circuits 332 in FIG. 3 . Each matching circuit 532 performs power and/or impedance matching between MIMO LNA 540 and either switches/duplexers 524 or antenna 510 for one or more bands of interest. The N matching circuits 532 a to 532 n may be designed for different bands and may provide up to N input RF signals, RFin 1 to RFinN.

MIMO LNA 540 receives up to N input RF signals and amplifies (i) one input RF signal for no CA or intra-band CA or (ii) multiple input RF signals for inter-band CA. MIMO LNA 540 provides up to M output RF signals, RFout 1 to RFoutM, via up to M LNA outputs. M load circuits 590 a to 590 m are coupled to the M LNA outputs. Each load circuit 590 may include one or more inductors, capacitors, transistors, mixers, etc. Each output RF signal may be provided to one or more mixers within one load circuit 590 and may be downconverted by the associated mixer(s) such that one or more transmissions on one or more carriers of interest are downconverted from RF to baseband.

A MIMO LNA, such as MIMO LNA 540 in FIG. 5A , may operate in a non-CA mode, an intra-band CA mode, or an inter-band CA mode at any given moment. In the non-CA mode, the MIMO LNA operates in a 1×1 configuration, receives one input RF signal comprising one or more transmissions on one set of carriers, and provides one output RF signal to one load circuit. In the intra-band CA mode, the MIMO LNA operates in a 1×M configuration, receives one input RF signal comprising multiple transmissions on M sets of carriers in the same band, and provides M output RF signals to M load circuits, one output RF signal for each set of carriers, where M>1. In the inter-band CA mode, the MIMO LNA operates in an N×M configuration, receives N input RF signals comprising multiple transmissions on M sets of carriers in up to N different bands, and provides M output RF signals to M load circuits, where M>1 and N>1. The N input RF signals may correspond to up to N different bands.

A MIMO LNA, such as MIMO LNA 540 in FIG. 5A , may be used to receive transmissions on multiple carriers at different frequencies. A MIMO LNA may include multiple outputs providing multiple output RF signals for different carriers or different sets of carriers of interest. A MIMO LNA is different from LNAs used to receive a MIMO transmission sent from multiple transmit antennas to multiple receive antennas. An LNA for a MIMO transmission typically has (i) one input receiving one input RF signal from one receive antenna and (ii) one output providing one output RF signal. The multiple outputs of a MIMO LNA thus cover frequency dimension whereas the outputs of LNAs used for a MIMO transmission cover spatial dimension.

FIG. 5B shows a schematic diagram of an exemplary design of a MIMO LNA 540 x supporting no CA, intra-band CA, and inter-band CA on two sets of carriers in different bands. Each set of carriers may include one or more carriers in one band. MIMO LNA 540 x is one exemplary design of MIMO LNA 540 in FIG. 5A . Matching circuits 532 a and 532 b may receive (i) the same receiver input signal from one antenna or (ii) different receiver input signals from one or more antennas. Hence, the RXin 2 signal may or may not be equal to the RXin 1 signal in FIG. 5B . Each matching circuit 532 performs power and/or impedance matching for one or more bands of interest.

In the exemplary design shown in FIG. 5B , MIMO LNA 540 x includes two amplifier stages 550 a and 550 b for two sets of carriers. Amplifier stage 550 a receives and amplifies the first input RF signal from matching circuit 532 a and provides a first output RF signal, RFout 1 , for a first set of carriers. Amplifier stage 550 b receives and amplifies the second input RF signal from matching circuit 532 b and provides a second output RF signal, RFout 2 , for a second set of carriers. Although not shown in FIG. 5B for simplicity, MIMO LNA 540 x may include circuitry to route an output RF signal from each amplifier stage 550 to any one of load circuits 590 x and 590 y.

In the exemplary design shown in FIG. 5B , load circuit 590 x includes two mixers 592 a and 592 b coupled to two baseband filters 594 a and 594 b , respectively. Mixer 592 a receives the first output RF signal from amplifier stage 550 a and an inphase LO signal, ILO 1 , at a first mixing frequency for the first set of carriers. Mixer 592 a downconverts the first output RF signal with the ILO 1 signal and provides an I downconverted signal. Mixer 592 b receives the first output RF signal from amplifier stage 550 b and a quadrature LO signal, QLO 1 , at the first mixing frequency for the first set of carriers. Mixer 592 b downconverts the first output RF signal with the QLO 1 signal and provides a Q downconverted signal. Filters 594 a and 594 b receive and filter the I and Q downconverted signals from mixers 592 a and 592 b , respectively, and provide I and Q baseband signals, Vout 1 , for the first set of carriers.

Mixers 592 c and 592 d and filters 594 c and 594 d within load circuit 590 y similarly process the second output RF signal from amplifier stage 550 b and provide I and Q baseband signals, Vout 2 , for a second set of carriers.

CA LNA 440 in FIG. 4A may be implemented in various manners. Some exemplary designs of CA LNA 440 are described below. CA LNA 440 may also be implemented with transistors of various types. Some exemplary designs of CA LNA 440 using N-channel metal oxide semiconductor (NMOS) transistors are described below.

FIG. 6A shows a schematic diagram of an exemplary design of a CA LNA 640 a with inductive degeneration and cascode shutoff CA LNA 640 a is one exemplary design of CA LNA 440 in FIG. 4A . CA LNA 640 a includes two amplifier stages 650 a and 650 b coupled to a common input matching circuit 632 and to two load circuits 690 a and 690 b . Matching circuit 632 receives a receiver input signal, RXin, performs input matching for CA LNA 640 a , and provides an input RF signal, RFin. Matching circuit 632 may correspond to matching circuit 432 in FIG. 4A . Load circuits 690 a and 690 b may correspond to load circuits 490 a and 490 m in FIG. 4A . CA LNA 640 a receives the input RF signal, which may include transmissions on two sets of carriers, with each set including one or more carriers.

In the exemplary design shown in FIG. 6A , amplifier stage 650 a includes a source degeneration inductor 652 a , a gain transistor 654 a , and a cascode transistor 656 a . Gain transistor 654 a and cascode transistor 656 a may be implemented with NMOS transistors (as shown in FIG. 6A ) or with transistors of other types. Gain transistor 654 a has its gate coupled to matching circuit 632 and its source coupled to one end of inductor 652 a . The other end of inductor 652 a is coupled to circuit ground. Cascode transistor 656 a has its source coupled to the drain of gain transistor 654 a and its drain coupled to load circuit 690 a . A switch 658 a has its input port coupled to the gate of cascode transistor 656 a , its first output port coupled to a bias voltage, Vcasc, and its second output port coupled to circuit ground. Amplifier stage 650 b includes a source degeneration inductor 652 b , a gain transistor 654 b , a cascode transistor 656 b , and a switch 658 b , which are coupled in similar manner as inductor 652 a , gain transistor 654 a , cascode transistor 656 a , and switch 658 a in amplifier stage 650 a.

For simplicity, FIG. 6A shows CA LNA 640 a including two amplifier stages 650 a and 650 b for two sets of carriers. Amplifier stages 650 a and 650 b may be independently enabled or disabled via switches 658 a and 658 b , respectively. CA LNA 640 a may include more than two amplifier stages 650 for more than two sets of carriers.

An input RF signal may include transmissions on multiple sets of carriers in the same band and may be referred to as a carrier-aggregated RF signal. The carrier-aggregated RF signal may be downconverted using LO signals at different frequencies corresponding to the center frequencies of the multiple sets of carriers on which the transmissions are sent. The carrier-aggregated RF signal may be split at the LNA input in order to achieve good LO-LO isolation between the LO signals for the multiple sets of carriers. CA LNA 640 a includes two amplifier stages 650 a and 650 b to amplify the carrier-aggregated RF signal and provide two output RF signals to two separate downconverters in the two load circuits 690 a and 690 b.

CA LNA 640 a may operate in a non-CA mode or a CA mode at any given moment. In the non-CA mode, CA LNA 640 a receives transmissions on one set of carriers and provides one output RF signal to one load circuit. In the CA mode, CA LNA 640 a receives transmissions on two sets of carriers and provides two output RF signals to two load circuits, one output RF signal for each set of carriers.

FIG. 6B shows operation of CA LNA 640 a in the CA mode. In the CA mode, both amplifier stages 650 a and 650 b are enabled by connecting the gate of cascode transistor 656 a to the Vcasc voltage via switch 658 a and coupling the gate of cascode transistor 656 b to the Vcasc voltage via switch 658 b . Amplifier stage 650 a amplifies the input RF signal and provides the first output RF signal to load circuit 690 a . Amplifier stage 650 b amplifies the input RF signal and provides the second output RF signal to load circuit 690 b.

FIG. 6C shows operation of CA LNA 640 a in the non-CA mode. In the non-CA mode, only one amplifier stage is enabled, and the other amplifier stage is disabled. In the example shown in FIG. 6C , amplifier stage 650 a is enabled by connecting the gate of cascode transistor 656 a to the Vcasc voltage via switch 658 a , and amplifier stage 650 b is disabled by shorting the gate of cascode transistor 656 b to circuit ground via switch 658 b . Amplifier stage 650 a amplifies the input RF signal and provides an output RF signal to load circuit 690 a.

In another configuration of the non-CA mode, amplifier stage 650 b is enabled, and amplifier stage 650 a is disabled (not shown in FIG. 6C ). In this configuration, amplifier stage 650 b amplifies the input RF signal and provides an output RF signal to load circuit 690 b.

In the exemplary design shown in FIG. 6A , separate source degeneration inductors 652 a and 652 b are used for amplifier stages 650 a and 650 b in order to reduce interaction between the two amplifier stages and to help reduce noise figure (NF) degradation. Source degeneration inductors 652 a and 652 b may also improve linearity of amplifier stages 650 a and 650 b and help input impedance matching of CA LNA 640 a . Inductors 652 a and 652 b may have the same value or different values. The values of inductors 652 a and 652 b may be selected (e.g., independently) based on a trade-off between voltage gain and linearity in the CA mode and the non-CA mode.

As shown in FIG. 6A , a variable capacitor 668 a may be present across the gate and source of gain transistor 654 a . Capacitor 668 a may include parasitic of gain transistor 654 a . Capacitor 668 a may also include a bank of switchable capacitors, which may be coupled between the gain and source of gain transistor 654 a and may be used to fine-tune the input impedance of CA LNA 640 a . Each switchable capacitor may be implemented with a capacitor coupled in series with a switch. Similarly, a variable capacitor 668 b may be present across the gate and source of gain transistor 654 b . Capacitor 668 b may include a bank of switchable capacitors, which may be coupled between the gain and source of gain transistor 654 b and may be used to fine-tune the input impedance.

Input matching circuit 632 is common to both amplifier stages 650 a and 650 b and is used in both the CA mode and the non-CA mode. In the CA mode, both amplifier stages 650 a and 650 b are enabled, and gain transistors 654 a and 654 b operate in a saturation region, as shown in FIG. 6B . In the non-CA mode, one amplifier stage (e.g., amplifier stage 650 a ) is enabled, and the other amplifier stage (e.g., amplifier stage 650 b ) is disabled. However, the gain transistor in the disabled amplifier stage (e.g., gain transistor 654 b in amplifier stage 650 b ) is turned On by the input RF signal that is applied to both gain transistors 654 a and 654 b . Since the cascode transistor in the disabled amplifier stage (e.g., cascode transistor 656 b ) is turned Off, the gain transistor in the disabled amplifier stage operates in a linear region. Hence, a gain transistor may operate in the saturation region when an amplifier stage is enabled and may operate in the linear region when the amplifier stage is disabled. Operating the gain transistor of the disabled amplifier stage in the linear region may help to reduce changes in the input impedance of CA LNA 640 a between the CA mode and the non-CA mode, without a current penalty in the disabled amplifier stage. In particular, the input capacitance, C IN , of a given gain transistor (e.g., gain transistor 654 b ) in an enabled amplifier stage and in a disabled amplifier stage may be expressed as:

⁢ C IN = 2 3 · W · L C OX amplifer ⁢ ⁢ stage ⁢ ⁢ is ⁢ ⁢ enabled , and Eq ⁢ ⁢ ( 1 ) C IN = 1 2 · W · L C OX amplifer ⁢ ⁢ stage ⁢ ⁢ is ⁢ ⁢ disabled , Eq ⁢ ⁢ ( 2 )

where W is the width and L is the length of gain transistor 654 b , and

C OX is a gate oxide capacitance of gain transistor 654 b.

As shown in equations (1) and (2), there may be a finite change in the input impedance of a gain transistor depending on whether an amplifier stage is enabled or disabled. However, the input impedance of CA LNA 640 a may be maintained within tolerable limits even with the change in the input impedance of the gain transistor.

CA LNA 640 a splits the carrier-aggregated RF signal at the “gate” level by having the carrier-aggregated RF signal applied to two gain transistors 654 a and 654 b . The carrier-aggregated RF signal may also be split at the “cascode” level by having the carrier-aggregated RF signal applied to a single gain transistor driving two cascode transistors. Splitting the carrier-aggregated RF signal at the gate level (as shown in FIG. 6A ) may provide better performance (e.g., better gain, noise figure, linearity, and isolation) than splitting the carrier-aggregated RF signal at the cascode level. For example, splitting the carrier-aggregated RF signal at the gate level may provide good LO-LO isolation of about 35 dB whereas splitting the carrier-aggregated RF signal at the cascode level may provide LO-LO isolation of only about 15 dB.

FIG. 7 shows a schematic diagram of an exemplary design of a CA LNA 640 b with inductive degeneration, cascode shutoff, and resistive feedback. CA LNA 640 b is another exemplary design of CA LNA 440 in FIG. 4A . CA LNA 640 b includes two amplifier stages 650 a and 650 b coupled to a common input matching circuit 632 and to two load circuits 690 a and 690 b , similar to CA LNA 640 a in FIG. 6A . CA LNA 640 b further includes a feedback circuit 660 coupled between the drains of cascode transistors 656 a and 656 b and the gates of gain transistors 654 a and 654 b , i.e., between the input and output of amplifier stages 650 a and 650 b.

In the exemplary design shown in FIG. 7 , feedback circuit 660 includes switches 662 a and 662 b , a resistor 664 , and a capacitor 666 . Resistor 664 and capacitor 666 are coupled in series, with the bottom terminal of capacitor 666 being coupled to the gates of gain transistors 654 a and 654 b . Switch 662 a is coupled between the drain of cascode transistor 656 a and the top terminal of resistor 664 . Switch 662 b is coupled between the drain of cascode transistor 656 b and the top terminal of resistor 664 . Switches 662 a and 662 b may each be closed to connect feedback circuit 660 to its associated cascode transistor 656 and may be opened to disconnect feedback circuit 660 from the associated cascode transistor 656 . A feedback path from RFout 1 to feedback circuit 660 may be formed by closing switch 662 a . A feedback path from RFout 2 to feedback circuit 660 may be formed by closing switch 662 b . Feedback circuit 660 may also include one or more active circuits such as a transistor. In an exemplary design, feedback circuit 660 may be used/enabled for low-band to provide input power match. For mid-band and high-band, feedback circuit 660 may be disabled, and source degeneration inductors 652 a and 652 b may be used with matching circuit 632 for input power match. Feedback circuit 660 may also be used in other manners.

Input matching circuit 632 is common to both amplifier stages 650 a and 650 b and is used in both the CA mode and the non-CA mode. In the CA mode, input matching for CA LNA 640 b may be achieved with feedback circuit 660 around amplifier stages 650 a and 650 b as well as source degeneration inductors 652 a and 652 b . In the non-CA mode, input matching for CA LNA 640 b may be achieved with feedback circuit 660 and source degeneration inductor 652 a and 652 b . Feedback circuit 660 may help with input matching for the entire LNA 640 b in both the CA mode and the non-CA mode. Input matching for CA LNA 640 b may be achieved (i) with feedback circuit 660 and source degeneration inductor 652 a for RFout 1 and (ii) with feedback circuit 660 and source degenerated inductor 652 b for RFout 2 .

Amplifier stage 650 a may be linearized by (i) both source degeneration inductor 652 a and feedback circuit 660 when feedback circuit 660 is selected or (ii) only source degeneration inductor 652 a when feedback circuit 660 is not selected. Feedback circuit 660 may improve the linearity of amplifier stage 650 a in both the CA mode and the non-CA mode. This may allow a smaller inductor 652 a to be used for amplifier stage 650 a to obtain the desired linearity. Similarly, amplifier stage 650 b may be linearized by (i) both source degeneration inductor 652 b and feedback circuit 660 when feedback circuit 660 is selected or (ii) only source degeneration inductor 652 b when feedback circuit 660 is not selected. A smaller inductor may be used for inductor 652 a and/or 652 b to obtain the desired linearity for amplifier stage 650 b with feedback circuit 660 enabled.

FIG. 8A shows a schematic diagram of an exemplary design of a CA LNA 840 a with a separate input attenuation circuit for each amplifier stage. CA LNA 840 a is yet another exemplary design of CA LNA 440 in FIG. 4A . CA LNA 840 a includes two amplifier stages 850 a and 850 b coupled to two input attenuation circuits 860 a and 860 b and to two load circuits 890 a and 890 b.

An input RF signal is provided to the input of CA LNA 840 a , which is node X. Amplifier stage 850 a is coupled to node X via an NMOS transistor 842 a operating as a switch, attenuation circuits 860 a , and an AC coupling capacitor 844 a . NMOS transistor 842 a has its source coupled to node X, its gate receiving a first control signal, Enb 1 , and its drain coupled to the input of attenuation circuit 860 a . Attenuation circuit 860 a includes (i) a resistor 862 a coupled between the input and output of attenuation circuit 860 a and (ii) a variable resistor 864 a coupled between the output of attenuation circuit 860 a and circuit ground. AC coupling capacitor 844 a is coupled between the output of attenuation circuit 860 a and the input of amplifier stage 850 a . Amplifier stage 850 b is coupled to node X via an NMOS transistor 842 b , an attenuation circuit 860 b , and an AC coupling capacitor 844 b , which are coupled in similar manner as NMOS transistor 842 a , attenuation circuit 860 a , and AC coupling capacitor 844 a.

Amplifier stage 850 a includes a gain transistor 854 a and a cascode transistor 856 a . Gain transistor 854 a has its gate coupled to AC coupling capacitor 844 a and its source coupled to circuit ground (as shown in FIG. 8A ) or to a source degeneration inductor (not shown in FIG. 8A ). Cascode transistor 856 a has its gate receiving a first bias voltage, Vcasc 1 , its source coupled to the drain of gain transistor 854 a , and its drain coupled to load circuit 890 a Amplifier stage 850 b includes a gain transistor 854 b and a cascode transistor 856 b , which are coupled in similar manner as gain transistor 854 a and cascode transistor 856 a Amplifier stages 850 a and 850 b may be independently selected via NMOS transistor 842 a and 842 b , respectively, and independently enabled or disabled via the Vcasc 1 and Vcasc 2 voltages, respectively.

FIG. 8A shows an exemplary design in which a signal path from the LNA input (node X) to each amplifier stage 850 includes NMOS transistor 842 , attenuation circuit 860 , and AC coupling capacitor 844 . A signal path may also include fewer, different, and/or additional circuits. Furthermore, the signal path for amplifier stage 850 a may or may not match the signal path for amplifier stage 850 b . For example, NMOS transistor 842 a may be omitted whereas NMOS transistor 842 b may be retained. Attenuation circuits 860 a and 860 b for the two signal paths may be identical, or may have the same circuit design but different values, or may have different circuit designs with different circuit topologies.

NMOS transistor 842 a operates as a switch that can pass the input RF signal to amplifier stage 850 a when NMOS transistor 842 a is enabled by the Enb 1 signal. Similarly, NMOS transistor 842 b operates as a switch that can pass the input RF signal to amplifier stage 850 b in the CA mode when NMOS transistor 842 b is enabled by the Enb 2 signal. In one design, NMOS transistors 842 a may be enabled in both the CA mode and the non-CA mode, and NMOS transistors 842 b may be enabled only in the CA mode. Separate NMOS transistors 842 a and 842 b and separate attenuation circuits 860 a and 860 b may be used to allow the input RF signal to encounter only one series switch prior to hitting gain transistor 854 a or 854 b.

In the CA mode, both NMOS transistors 842 a and 842 b are turned On, and the input RF signal is provided to both attenuation circuits 860 a and 860 b and amplifier stages 850 a and 850 b . Each amplifier stage 850 amplifies the input RF signal and provides a respective output RF signal to its load circuit 890 . In the non-CA mode, one amplifier stage 850 a or 850 b may be selected. NMOS transistor 842 for the selected amplifier stage 850 is turned On, and the input RF signal is provided to attenuation circuit 860 and the selected amplifier stage 850 . NMOS transistor 842 for the unselected amplifier stage 850 is turned Off, and attenuation circuit 860 and the unselected amplifier stage 850 are disconnected from node X, thereby reducing loading on the signal path for the selected amplifier stage 850 . The selected amplifier stage 850 amplifies the input RF signal and provides an output RF signal to the associated load circuit 890 .

CA LNA 840 a may be advantageously used in a scenario in which the input RF signal includes jammers, which are undesired signals of large amplitude and close in frequency to the desired signals. Input attenuation circuits 860 a and 860 b may be programmable (e.g., as shown in FIG. 8A ) or may be fixed (not shown in FIG. 8A ) and may serve a dual purpose of attenuating the jammers in the input RF signal and providing a good input impedance match for CA LNA 840 a . Attenuation circuits 860 a and 860 b may be designed differently and/or may have different settings/values for the CA mode and the non-CA mode in order to obtain a good input impedance match in both modes.

For simplicity, FIG. 8A shows CA LNA 840 a including two amplifier stages 850 a and 850 b for two sets of carriers. CA LNA 840 a may include more than two amplifier stages for more than two sets of carriers.

FIG. 8B shows a schematic diagram of an exemplary design of a CA LNA 840 b with a shared input attenuation circuit for both amplifier stages 850 a and 850 b . CA LNA 840 b is yet another exemplary design of CA LNA 440 in FIG. 4A . CA LNA 840 b includes two amplifier stages 850 a and 850 b coupled to a shared input attenuation circuit 860 a and to two load circuits 890 a and 890 b . Sharing input attenuation circuit 860 a between amplifier stages 850 a and 850 b may reduce circuit area and may also provide other advantages.

CA LNA 840 b includes all of the circuit components in CA LNA 840 a in FIG. 8A except for input attenuation circuit 860 b and AC coupling capacitor 844 b . All of the circuit components within CA LNA 840 b are coupled as described above for CA LNA 840 a in FIG. 8A except for NMOS transistor 842 b . NMOS transistor 842 b has its source coupled to the input of amplifier stage 850 a , its gate receiving an Enb 2 control signal, and its drain coupled to the input of amplifier stage 850 b Amplifier stages 850 a and 850 b may be independently selected via NMOS transistor 842 a and 842 b , respectively, and may be independently enabled or disabled via the Vcasc 1 and Vcasc 2 voltages, respectively.

In the exemplary design shown in FIG. 8B , amplifier stages 850 a and 850 b share a common input switch implemented with NMOS transistor 842 a and a common input attenuation circuit 860 a . Input attenuation circuit 860 a may be programmable (e.g., as shown in FIG. 8B ) or may be fixed (not shown in FIG. 8B ) and may serve a dual purpose of attenuating incoming jammers in the input RF signal and providing a good input impedance match for CA LNA 840 b . Attenuation circuit 860 a may have different settings for the CA mode and the non-CA mode in order to obtain a good input impedance match in both modes.

In the CA mode, both NMOS transistors 842 a and 842 b are turned On, and the input RF signal is provided via attenuation circuit 860 a to both amplifier stages 850 a and 850 b . The input RF signal passes through a single series switch prior to hitting gain transistor 854 a . The input RF signal passes through two series switches prior to hitting gain transistor 854 b , which may result in a small degradation in performance of amplifier stage 850 b . In the non-CA mode, NMOS transistor 842 a is turned On, and the input RF signal is provided to attenuation circuit 860 a and amplifier stage 850 a . NMOS transistor 842 b is turned Off, and amplifier stage 850 b is disconnected from node Y, thereby reducing capacitive loading on the signal path for amplifier stage 850 a . The input RF signal passes through a single series switch implemented with NMOS transistor 842 a prior to hitting gain transistor 854 a in the non-CA mode.

FIG. 9 shows a schematic diagram of an exemplary design of a CA LNA 940 with a tunable matching circuit. CA LNA 940 is another exemplary design of CA LNA 440 in FIG. 4A . CA LNA 940 includes two amplifier stages 950 a and 950 b coupled to a shared tunable matching circuit 932 and to two load circuits 990 a and 990 b . Amplifier stage 950 a includes a source degeneration inductor 952 a , a gain transistor 954 a , and a cascode transistor 956 a , which are coupled in similar manner as inductor 652 a , gain transistor 654 a , and cascode transistor 656 a in FIG. 6A . Amplifier stage 950 b includes a source degeneration inductor 952 b , a gain transistor 954 b , and a cascode transistor 956 b , which are also coupled in similar manner as inductor 652 a , gain transistor 654 a , and cascode transistor 656 a in FIG. 6A . Cascode transistor 956 a has its gate receiving a first control voltage, Vcasc 1 . Cascode transistor 956 b has its gate receiving a second control voltage, Vcasc 2 . Each amplifier stage 950 may be independently enabled or disabled based on its Vcasc control voltage. Amplifier stages 950 a and 950 b may be independently enabled or disabled via the Vcasc 1 and Vcasc 2 voltages, respectively.

Matching circuit 932 receives an input RF signal and performs input matching for CA LNA 940 . An AC coupling capacitor 944 a has one end coupled to the output of matching circuit 932 and the other end coupled to the gate of gain transistor 954 a . An AC coupling capacitor 944 b has one end coupled to the output of matching circuit 932 and the other end coupled to the gate of gain transistor 954 b.

CA LNA 940 supports the CA mode and the non-CA mode. In the CA mode, both amplifier stages 950 a and 950 b are enabled with the Vcasc 1 and Vcasc 2 voltages applied to cascode transistors 956 a and 956 b , respectively. In the non-CA mode, only one of the two amplifier stages 950 a and 950 b is enabled, depending on the particular load circuit to which the input RF signal is to be routed. Matching circuit 932 may be adjusted based on the number of enabled amplifier stages and/or which amplifier stage(s) are enabled in order to obtain good noise/power match in both the CA modes and the non-CA mode.

MIMO LNA 540 in FIG. 5A may be implemented with various circuit architectures. Some exemplary designs of MIMO LNA 540 are described below. MIMO LNA 540 may also be implemented with transistors of various types. Some exemplary designs of MIMO LNA 540 using NMOS transistors are described below.

FIG. 10 shows a schematic diagram of an exemplary design of a 2×2 MIMO LNA 1040 based on a cascode shutoff architecture. MIMO LNA 1040 is an exemplary design of MIMO LNA 540 in FIG. 5A and includes (i) two LNA inputs receiving two input RF signals, RFin 1 and RFin 2 , and (ii) two LNA outputs providing two output RF signals, RFout 1 and RFout 2 .

MIMO LNA 1040 includes four amplifier stages 1050 a to 1050 d coupled to two load circuits 1090 a and 1090 b . Each amplifier stage 1050 a includes a source degeneration inductor 1052 , a gain transistor 1054 , and a cascode transistor 1056 , which are coupled in similar manner as inductor 652 a , gain transistor 654 a , and cascode transistor 656 a in FIG. 6A . Gain transistors 1054 a and 1054 b within amplifier stages 1050 a and 1050 b have their gates receiving the first input RF signal. Gain transistors 1054 c and 1054 d within amplifier stages 1050 c and 1050 d have their gates receiving the second input RF signal. Cascode transistors 1056 a and 1056 c within amplifier stages 1050 a and 1050 c have their drains coupled to load circuit 1090 a and providing the first output RF signal. Cascode transistors 1056 b and 1056 d within amplifier stages 1050 b and 1050 d have their drains coupled to load circuit 1090 b and providing the second output RF signal.

Amplifier stages 1050 a and 1050 b may be designed to provide good performance for a first set of one or more bands, e.g., as described above for CA LNA 640 a in FIG. 6A . Similarly, amplifier stages 1050 c and 1050 d may be designed to provide good performance for a second set of one or more bands.

MIMO LNA 1040 may operate in a 1×2 configuration for intra-band CA. In the 1×2 configuration, an input RF signal, RFin 1 or RFin 2 , may be provided via one LNA input to two gain transistors 1054 in two amplifier stages 1050 . The input RF signal is amplified by the two gain transistors, buffered by the two cascode transistors coupled to the two gain transistors, and provided to load circuits 1090 a and 1090 b . MIMO LNA 1040 can support intra-band CA with the input RF signal provided to either of the two LNA inputs.

MIMO LNA 1040 may operate in a 2×2 configuration for inter-band CA. In the 2×2 configuration, a first input RF signal may be amplified by a first selected gain transistor 1054 in one amplifier stage 1050 , buffered by cascode transistor 1056 coupled to the first selected gain transistor 1054 , and provided to load circuit 1090 a . A second input RF signal may be amplified by a second selected gain transistor 1054 in another amplifier stage 1050 , buffered by cascode transistor 1056 coupled to the second selected gain transistor 1054 , and provided to load circuit 1090 b.

FIG. 11A shows a schematic diagram of an exemplary design of a 2×2 MIMO LNA 1140 a based on the cascode shutoff architecture. MIMO LNA 1140 a is another exemplary design of MIMO LNA 540 in FIG. 5A and includes (i) two LNA inputs receiving two input RF signals, RFin 1 and RFin 2 , and (ii) two LNA outputs providing two output RF signals, RFout 1 and RFout 2 .

MIMO LNA 1140 a includes two amplifier stages 1150 a and 1150 b coupled to two load circuits 1190 a and 1190 b , respectively. Amplifier stage 1150 a includes a source degeneration inductor 1152 a , a gain transistor 1154 a , and a cascode transistor 1156 a , which are coupled in similar manner as inductor 652 a , gain transistor 654 a , and cascode transistor 656 a in FIG. 6A . Gain transistor 1154 a has its gate receiving the first input RF signal. Cascode transistor 1156 a has its gate receiving a Vcasc 1 a voltage and its drain coupled to load circuit 1190 a . Amplifier stage 1150 a further includes a gain transistor 1164 a and a cascode transistor 1166 a . Gain transistor 1164 a has its source coupled to inductor 1152 a and its gate receiving the second input RF signal. Cascode transistor 1166 a has its gate receiving a Vcasc 2 a voltage and its drain coupled to load circuit 1190 a.

Amplifier stage 1150 b includes a source degeneration inductor 1152 b , two gain transistors 1154 b and 1164 b , and two cascode transistors 1156 b and 1166 b , which are coupled in similar manner as inductor 1152 a , gain transistors 1154 a and 1164 a , and cascode transistors 1156 a and 1166 a in amplifier stage 1150 a . Gain transistors 1154 b and 1164 b have their gates receiving the first and second input RF signals, respectively. Cascode transistors 1156 b and 1166 b have their gates receiving Vcasc 1 b and Vcasc 2 b voltages, respectively, and their drains coupled to load circuit 1190 b.

MIMO LNA 1140 a may operate in a 1×2 configuration for intra-band CA. In the 1×2 configuration, an input RF signal, RFin 1 or RFin 2 , may be provided via one LNA input to two gain transistors 1154 a and 1154 b (or to gain transistors 1164 a and 1164 b ) in the two amplifier stages 1150 a and 1150 b . The input RF signal is amplified by the two gain transistors, buffered by the two cascode transistors coupled to the two gain transistors, and provided to load circuits 1190 a and 1190 b . MIMO LNA 1140 a can support intra-band CA with the input RF signal applied to either of the two LNA inputs.

MIMO LNA 1140 a may operate in a 2×2 configuration for inter-band CA. In the 2×2 configuration, a first input RF signal (e.g., RFin 1 ) may be received by amplifier stage 1150 a or 1150 b and may be amplified by gain transistor 1154 a or 1154 b , buffered by cascode transistor 1156 a or 1156 b , and provided to load circuit 1190 a or 1190 b . A second input RF signal (e.g., RFin 2 ) may be received by amplifier stage 1150 a or 1150 b , amplified by gain transistor 1164 a or 1164 b , buffered by cascode transistor 1166 a or 1166 b , and provided to load circuit 1190 a or 1190 b . Each amplifier stage 1150 would receive only one of the two input RF signals and would provide its output RF signal to load circuit 1190 coupled to that amplifier stage 1150 .

FIG. 11B shows a schematic diagram of an exemplary design of a 4×2 MIMO LNA 1140 b based on the cascode shutoff architecture. MIMO LNA 1140 b is yet another exemplary design of MIMO LNA 540 in FIG. 5A and includes (i) four LNA inputs receiving four input RF signals, RFin 1 to RFin 4 , and (ii) two LNA outputs providing two output RF signals, RFout 1 and RFout 2 . MIMO LNA 1140 b includes four amplifier stages 1150 a to 1150 d coupled to two load circuits 1190 a and 1190 b . Amplifier stages 1150 a and 1150 b include source degeneration inductors, gain transistors, and cascode transistors that are coupled as described above for FIG. 11A .

Amplifier stage 1150 c includes a source degeneration inductor 1152 c , two gain transistors 1154 c and 1164 c , and two cascode transistors 1156 c and 1166 c , which are coupled in similar manner as inductor 1152 a , gain transistors 1154 a and 1164 a , and cascode transistors 1156 a and 1166 a in amplifier stage 1150 a . Gain transistors 1154 c and 1164 c have their gates receiving the third and fourth input RF signals, respectively. Cascode transistors 1156 c and 1166 c have their gates receiving Vcasc 3 a and Vcasc 4 a voltages, respectively, and their drains coupled to load circuit 1190 a.

Amplifier stage 1150 d includes a source degeneration inductor 1152 d , two gain transistors 1154 d and 1164 d , and two cascode transistors 1156 d and 1166 d , which are coupled in similar manner as inductor 1152 a , gain transistors 1154 a and 1164 a , and cascode transistors 1156 a and 1166 a in amplifier stage 1150 a . Gain transistors 1154 d and 1164 d have their gates receiving the third and fourth input RF signals, respectively. Cascode transistors 1156 d and 1166 d have their gates receiving Vcasc 3 b and Vcas 4 b voltages, respectively, and their drains coupled to load circuit 1190 b.

MIMO LNA 1140 b may operate in a 1×2 configuration for intra-band CA. In the 1×2 configuration, an input RF signal (RFin 1 , RFin 2 , RFin 3 or RFin 4 ) may be provided via one LNA input to two gain transistors in two amplifier stages 1150 . The input RF signal is amplified by the two gain transistors, buffered by the two cascode transistors coupled to the two gain transistors, and provided to load circuits 1190 a and 1190 b . MIMO LNA 1140 b can support intra-band CA with the input RF signal applied to any one of the four LNA inputs.

MIMO LNA 1140 b may operate in a 2-input 2-output (2×2) configuration for inter-band CA. In the 2×2 configuration, a first input RF signal (e.g., RFin 1 or RFin 2 ) may be received by amplifier stage 1150 a or 1150 b , amplified by a first selected gain transistor in one amplifier stage 1150 a or 1150 b , buffered by the cascode transistor coupled to the first selected gain transistor, and provided to load circuit 1190 a or 1190 b . A second input RF signal (e.g., RFin 3 or RFin 4 ) may be received by amplifier stage 1150 c or 1150 d , amplified by a second selected gain transistor in amplifier stage 1150 c or 1150 d , buffered by the cascode transistor coupled to the second selected gain transistor, and provided to load circuit 1190 a or 1190 b . Only two amplifier stages 1150 are enabled to amplify the two input RF signals. Each enabled amplifier stage 1150 would receive only one of the two input RF signals and would provide its output RF signal to load circuit 1190 coupled to that amplifier stage 1150 .

FIG. 11C shows a schematic diagram of another view of 4×2 MIMO LNA 1140 b in FIG. 11B . MIMO LNA 1140 b in FIG. 11C include all of the degeneration inductors, gain transistors, and cascode transistors shown in FIG. 11B , which are arranged differently in FIG. 11C . FIG. 11C shows that MIMO LNA 1140 b can receive a single input RF signal provided to any LNA input (e.g., RFin 1 , RFin 2 , RFin 3 , or RFin 4 ) and can provide two output RF signals to two load circuits 1190 a and 1190 b for intra-band CA. FIG. 11C also shows that MIMO LNA 1140 b can receive two input RF signals provided to two LNA inputs (e.g., RFin 1 and RFin 3 , or RFin 1 and RFin 4 , or RFin 2 and RFin 3 , or RFin 2 and RFin 4 ) and can provide two output RF signals to two load circuits 1190 a and 1190 b for inter-band CA. If a separate source degeneration inductor is used for each gain transistor (for a total of eight source degeneration inductors), then two input RF signals may be applied to any two LNA inputs.

FIGS. 10 and 11A show two exemplary designs of a 2×2 MIMO LNA. FIG. 11B shows an exemplary design of a 4×2 MIMO LNA. A MIMO LNA may also be implemented in other manners. For example, a MIMO LNA may include one or more feedback circuits, with each feedback circuit being coupled between the input and output of one or more amplifier stages, e.g., as shown in FIG. 7 .

In general, a MIMO LNA with any number of inputs and any number of outputs may be implemented based on the cascode shutoff architecture. More LNA inputs for more bands may be supported with more amplifier stages and/or more gain and cascode transistors in each amplifier stage. More LNA outputs for more sets of carriers may also be supported with more amplifier stages and/or more gain and cascode transistors in each amplifier stage.

A matching circuit and a tunable matching circuit may be implemented in various manners. Some exemplary designs of a tunable matching circuit are described below.

FIG. 12A shows an exemplary design of a tunable matching circuit 1210 based on an L topology. The L topology includes a series circuit component coupled to a shunt circuit component. A series circuit component is a circuit component connected between two nodes. A shunt circuit component is a circuit component connected between a node and circuit ground. A circuit component may be an inductor, a capacitor, a resistor, etc. Matching circuit 1210 includes (i) a series inductor 1212 coupled between the input and output of matching circuit 1210 and (ii) a tunable shunt capacitor 1214 coupled between the output of matching circuit 1210 and circuit ground.

FIG. 12B shows an exemplary design of a tunable matching circuit 1220 based on the L topology. Matching circuit 1220 includes (i) a tunable series capacitor 1222 coupled between the input and output of matching circuit 1220 and (ii) a shunt inductor 1224 coupled between the output of matching circuit 1220 and circuit ground.

FIG. 12C shows an exemplary design of a tunable matching circuit 1230 based on an R topology. The R topology includes a shunt circuit component coupled to a series circuit component. Matching circuit 1230 includes (i) a tunable shunt capacitor 1232 coupled between the input of matching circuit 1230 and circuit ground and (ii) a series inductor 1234 coupled between the input and output of matching circuit 1230 .

FIG. 12D shows an exemplary design of a tunable matching circuit 1240 based on a Pi topology. The Pi topology includes a shunt circuit component coupled to a series circuit component, which is coupled to another shunt circuit component. Matching circuit 1240 includes (i) a shunt capacitor 1242 coupled between the input of matching circuit 1240 and circuit ground, (ii) a series inductor 1244 coupled between the input and output of matching circuit 1240 , and (iii) a tunable shunt capacitor 1246 coupled between the output of matching circuit 1240 and circuit ground.

FIG. 12E shows an exemplary design of a tunable matching circuit 1250 with two R sections. Matching circuit 1250 includes (i) a shunt inductor 1252 coupled between the input of matching circuit 1250 and a power supply, Vdd, (ii) a series capacitor 1254 coupled between the input of matching circuit 1250 and node E, (iii) a tunable shunt capacitor 1256 coupled between node E and circuit ground, and (iv) a series inductor 1258 coupled between node E and the output of matching circuit 1250 .

FIG. 12F shows an exemplary design of a tunable matching circuit 1260 based on the Pi topology. Matching circuit 1260 includes (i) a shunt inductor 1262 coupled between the input of matching circuit 1260 and the Vdd supply, (ii) a series capacitor 1264 coupled between the input and output of matching circuit 1260 , (iii) a tunable shunt capacitor 1266 coupled between the output of matching circuit 1260 and circuit ground, and (iv) a shunt inductor 1268 coupled between the output of matching circuit 1260 and circuit ground.

A fixed matching circuit may also be implemented based on any of the exemplary designs shown in FIGS. 12A to 12F . In this case, each adjustable circuit component (e.g., each adjustable capacitor) may be replaced with a fixed circuit component (e.g., a fixed capacitor).

In an exemplary design, an apparatus (e.g., a wireless device, an IC, a circuit module, etc.) may include first and second amplifier stages (e.g., for a CA LNA or a MIMO LNA). The first amplifier stage (e.g., amplifier stage 650 a in FIG. 6A ) may receive and amplify an input RF signal and provide a first output RF signal to a first load circuit (e.g., load circuit 690 a ) when the first amplifier stage is enabled. The input RF signal may comprise transmissions sent on multiple carriers at different frequencies to a wireless device. The second amplifier stage (e.g., amplifier stage 650 b in FIG. 6A ) may receive and amplify the input RF signal and provide a second output RF signal to a second load circuit (e.g., load circuit 690 b ) when the second amplifier stage is enabled. Each load circuit may comprise at least one mixer (e.g., as shown in FIG. 4B or 5 B) and/or other circuits. The first output RF signal may be processed (e.g., downconverted) for transmissions on a first set of at least one carrier. The second output RF signal may be processed for transmissions on a second set of at least one carrier.

In an exemplary design, the first amplifier stage may comprise a first gain transistor (e.g., gain transistor 654 a in FIG. 6A ) coupled to a first cascode transistor (e.g., cascode transistor 656 a ). The second amplifier stage may comprise a second gain transistor (e.g., gain transistor 654 b ) coupled to a second cascode transistor (e.g., cascode transistor 656 b ). The input RF signal may be provided to both the first and second gain transistors. In an exemplary design, the first amplifier stage may further comprise a first inductor (e.g., inductor 652 a ) coupled to the first gain transistor. The second amplifier stage may further comprise a second inductor (e.g., inductor 652 b ) coupled to the second gain transistor. In another exemplary design, the first and second gain transistors may have their sources coupled to circuit ground (e.g., as shown in FIGS. 8A and 8B ).

In an exemplary design, the first and second amplifier stages may (i) provide the first and second output RF signals in a first/CA mode and (ii) provide the first output RF signal but not the second output RF signal in a second/non-CA mode. The first and second cascode transistors may both be enabled in the first/CA mode. Only one of the first and second cascode transistors may be enabled in the second/non-CA mode. The first and second gain transistors may be applied the input RF signal in both the first/CA mode and the second/non-CA mode. One of the first and second gain transistors may operate in a saturation region and the other one of the first and second gain transistors may operate in a linear region in the second/non-CA mode. Each amplifier stage may be enabled or disabled by providing one or more appropriate voltages to one or more cascode transistors in the amplifier stage.

In an exemplary design, a feedback circuit (e.g., feedback circuit 660 in FIG. 7 ) may be coupled between an output and an input of at least one of the first and second amplifier stages. The feedback circuit may comprise a resistor, or a capacitor, or an active circuit such as a transistor, or some other circuit, or any combination thereof.

In an exemplary design, separate attenuation circuits may be used for the amplifier stages, e.g., as shown in FIG. 8A . A first attenuation circuit (e.g., attenuation circuit 860 a in FIG. 8A ) may be coupled to the first amplifier stage and may receive the input RF signal and provide a first attenuated input RF signal to the first amplifier stage. A second attenuation circuit (e.g., attenuation circuit 860 b in FIG. 8A ) may be coupled to the second amplifier stage and may receive the input RF signal and provide a second attenuated input RF signal to the first amplifier stage.

In another exemplary design, a shared/common attenuation circuit may be used for all amplifier stages, e.g., as shown in FIG. 8B . The share attenuation circuit (e.g., attenuation circuit 860 a in FIG. 8B ) may be coupled to the first and second amplifier stages and may receive the input RF signal and provide an attenuated input RF signal to both amplifier stages.

In an exemplary design, an input matching circuit may be used for the amplifier stages. The input matching circuit (e.g., input matching circuit 632 in FIG. 6A ) may be coupled to the first and second amplifier stages and may receive a receiver input signal and provide the input RF signal. The input matching circuit may be fixed (e.g., as shown in FIG. 6A ) and may comprise one or more fixed circuit components. Alternatively, the input matching circuit may be tunable (e.g., as shown in FIG. 9 ) and may comprise at least one adjustable circuit component.

In an exemplary design, the apparatus may further component third and fourth amplifier stages (e.g., amplifier stages 1050 c and 1050 d in FIG. 10 for a MIMO LNA). The third amplifier stage (e.g., amplifier stage 1050 c ) may receive and amplify a second input RF signal and provide the first output RF signal to the first load circuit when the third amplifier stage is enabled. The fourth amplifier stage (e.g., amplifier stage 1050 d ) may receive and amplify the second input RF signal and provide the second output RF signal to the second load circuit when the fourth amplifier stage is enabled.

In another exemplary design, the first amplifier stage (e.g., amplifier stage 1150 a in FIG. 11A ) may receive and amplify the input RF signal or a second input RF signal and may provide the first output RF signal to the first load circuit when the first amplifier stage is enabled. The second amplifier stage (e.g., amplifier stage 1150 b ) may receive and amplify the input RF signal or the second input RF signal and may provide the second output RF signal to the second load circuit when the second amplifier stage is enabled. The first amplifier stage may further comprise a third gain transistor (e.g., gain transistor 1164 a ) coupled to a third cascode transistor (e.g., cascode transistor 1166 a ). The second amplifier stage may further comprise a fourth gain transistor (e.g., gain transistor 1164 b ) coupled to a fourth cascode transistor (e.g., cascode transistor 1166 b ). The second input RF signal may be provided to both the third and fourth gain transistors.

In another exemplary design, the apparatus may further include third and fourth amplifier stages (e.g., amplifier stages 1150 c and 1150 d in FIG. 11B for a MIMO LNA). The third amplifier stage (e.g., amplifier stage 1150 c ) may receive and amplify a third input RF signal or a fourth input RF signal and may provide the first output RF signal to the first load circuit when the third amplifier stage is enabled. The fourth amplifier stage (e.g., amplifier stage 1150 d ) may receive and amplify the third input RF signal or the fourth input RF signal and may provide the second output RF signal to the second load circuit when the fourth amplifier stage is enabled.

FIG. 13 shows an exemplary design of a process 1300 for receiving signals in a wireless system. Process 1300 may be performed by a wireless device (as described below) or by some other entity. The wireless device may enable first and second amplifier stages in a first/CA mode (block 1312 ). The wireless device may enable the first amplifier stage and disable the second amplifier stage in a second/non-CA mode (block 1314 ). The wireless device may amplify a first input RF signal with the first amplifier stage to obtain a first output RF signal when the first amplifier stage is enabled (block 1316 ). The wireless device may amplify the first input RF signal or a second input RF signal with the second amplifier stage to obtain a second output RF signal when the second amplifier stage is enabled (block 1318 ). The first and second input RF signals may comprise transmissions sent on multiple carriers at different frequencies to the wireless device. The first and second input RF signals may be for different bands.

The LNAs described herein may be implemented on an IC, an analog IC, an RFIC, a mixed-signal IC, an ASIC, a printed circuit board (PCB), an electronic device, etc. The LNAs may also be fabricated with various IC process technologies such as complementary metal oxide semiconductor (CMOS), N-channel MOS (NMOS), P-channel MOS (PMOS), bipolar junction transistor (BJT), bipolar-CMOS (BiCMOS), silicon germanium (SiGe), gallium arsenide (GaAs), heterojunction bipolar transistors (HBTs), high electron mobility transistors (HEMTs), silicon-on-insulator (SOI), etc.

An apparatus implementing the LNAs described herein may be a stand-alone device or may be part of a larger device. A device may be (i) a stand-alone IC, (ii) a set of one or more ICs that may include memory ICs for storing data and/or instructions, (iii) an RFIC such as an RF receiver (RFR) or an RF transmitter/receiver (RTR), (iv) an ASIC such as a mobile station modem (MSM), (v) a module that may be embedded within other devices, (vi) a receiver, cellular phone, wireless device, handset, or mobile unit, (vii) etc.

In one or more exemplary designs, the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code in the form of instructions or data structures and that can be accessed by a computer. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media.

The previous description of the disclosure is provided to enable any person skilled in the art to make or use the disclosure. Various modifications to the disclosure will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other variations without departing from the scope of the disclosure. Thus, the disclosure is not intended to be limited to the examples and designs described herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.