Ladder filter, duplexer and module

This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2011-100193, filed on Apr. 27, 2011, the entire contents of which are incorporated herein by reference.

A certain aspect of the present invention relates to ladder filters, duplexers and modules.

Recently, portable telephones and portable information terminals have rapidly spread due to the development of mobile communication systems. Thus, there is an increasing demand for RF filters, particularly, those having a compact size and a sharp characteristic. The filters are required to generate an attenuation pole and have sufficient suppression over a broad band in order to handle a new communication system. Such filters may be a ladder filter in which multiple resonators are connected. A duplexer or module equipped with filters is required to have transmission and reception filters configured so that the transmission filter has high suppression in the pass band of the reception filter and the reception filter has high suppression in the pass band of the transmission filter. There is a proposal of a filter equipped with an inductor in order to adjust the filter characteristics.

Japanese Patent Application Publication No. 6-350390 (Document 1) discloses a resonator with which an LC resonance circuit is connected in parallel. Japanese Patent Application Publication No. 11-55067 (Document 2) discloses a ladder filter to which an inductor is connected. The arts disclosed in Documents 1 and 2 are capable of generating an attenuation pole outside of the pass band.

According to an aspect of the present invention, there are provided a ladder filter, a duplexer and a module having high suppression over a broad band.

According to another aspect of the present invention, there is provided a ladder filter including: at least one series resonator connected in series between an input terminal and an output terminal; at least one parallel resonator connected in parallel with the at least one series resonator; an additional resonator connected in series between the at least one series resonator and one of the input terminal and the output terminal; and an inductor connected to the additional resonator in series, the additional resonator having a resonance frequency higher than an anti-resonance frequency of the at least one series resonator.

FIG. 1A is a circuit diagram of a series resonator, FIG. 1B is a circuit diagram of a parallel resonator, and FIG. 1C is a diagram of a pass characteristic of the series resonator and that of the parallel resonator;

FIG. 2A is a circuit diagram of a single-stage ladder filter, and FIG. 2B is a diagram of a pass characteristic of the single-stage ladder filter;

FIG. 3A is a circuit diagram of a ladder filter having attenuation poles at the high-frequency side of the ladder filter, and FIG. 3B is a diagram of an attenuation characteristic of the ladder filter;

FIG. 4 is a circuit diagram of a ladder filter in accordance with a comparative example;

FIG. 5A is a diagram of an attenuation characteristic of the ladder filter in accordance with the comparative example, and FIG. 5B is an enlarged view of a pass band illustrated in FIG. 5A ;

FIG. 6 is a circuit diagram of a duplexer using a ladder filter having an inductor in a series arm;

FIG. 7 is a diagram of a reactance characteristic of a series resonator Sx;

FIG. 8A is a diagram of an attenuation characteristic of a transmission filter F 13 ; and FIG. 8B is an enlarged view of a pass band illustrated in FIG. 8A ;

FIG. 9 is a plan view of a surface acoustic wave resonator;

FIG. 10A is a diagram of a pass characteristic of a surface acoustic wave resonator; and FIG. 10B is a diagram of a Q value of the surface acoustic wave resonator;

FIG. 11A is a block diagram of an exemplary duplexer, and FIG. 11B is a block diagram of an exemplary RF (Radio Frequency) module in accordance with a first embodiment;

FIG. 12 is a circuit diagram of a duplexer in accordance with the first embodiment;

FIG. 13A is a plan view of a duplexer chip in accordance with the first embodiment, FIG. 13B is a cross-sectional view taken along a line A 1 -A 1 in FIG. 13A , and FIG. 13C is a plan view of the duplexer;

FIG. 14 is a transparent view of a transmission filter chip installed in the duplexer of the first embodiment; and

FIGS. 15A , 15 B and 15 C are respectively plan views of a package substrate.

A ladder filter is now described. FIG. 1A is a circuit diagram of a series resonator, FIG. 1B is a circuit diagram of a parallel resonator, and FIG. 1C is a diagram of a pass characteristic of the series resonator and that of the parallel resonator.

As illustrated in FIG. 1A , the series resonator has a resonator S 21 having a pair of ports (signal terminals), one of which is an input terminal In and the other is an output terminal Out. As illustrated in FIG. 1B , the parallel resonator has a resonator P 21 having a pair of ports, one of which is connected to a ground terminal and the other is connected to a short-circuit line between the input terminal In and the output terminal Out.

The horizontal axis of FIG. 1C denotes the frequency, and the vertical axis thereof denotes the amplitude or magnitude. The pass characteristic of the series resonator is indicated by a solid line, and that of the parallel resonator is indicated by a broken line. As illustrated in FIG. 1C , the pass characteristic of the series resonator has one resonance point (resonance frequency) f rs and one anti-resonance point (anti-resonance frequency) f as . The amplitude is the largest at the resonance point f rs , and is the smallest at the anti-resonance point f as . The pass characteristic of the parallel resonator has one resonance point f rp and one anti-resonance point f ap . The amplitude is the smallest at the resonance point frp and is the largest at the anti-resonance point f ap .

FIG. 2A is a circuit diagram of a single-stage ladder filter, and FIG. 2B illustrates a pass characteristic of the ladder filter illustrated in FIG. 2A .

As illustrated in FIG. 2A , a series resonator S 22 is connected in series to the input terminal In and the output terminal Out, and a parallel resonator P 22 is connected between the output terminal Out and ground. The resonance point f rs of the series resonator S 22 and the anti-resonance point f ap of the parallel resonator P 22 are designed to be approximately equal to each other.

The horizontal axis of FIG. 2B is the frequency, and the vertical axis thereof is the amplitude. The ladder filter in FIG. 2A has a pass characteristic as illustrated in FIG. 2B , which is a combination of the pass characteristic of the series resonator and that of the parallel resonator. The amplitude is the largest around the resonance point f rs of the series resonator S 22 and the anti-resonance point f ap of the parallel resonator P 22 , and is the smallest around the anti-resonance point f as of the series resonator S 22 and the resonance point f rp of the parallel resonator P 22 . The ladder filter in FIG. 2A has a pass band between the resonance frequency f rp of the parallel resonator P 22 and the anti-resonance point f as of the series resonator S 22 , and has an attenuation range equal to or lower than the resonance frequency f rp of the parallel resonator P 22 and that equal to or higher than the anti-resonance point f as of the series resonator S 22 . Thus, the ladder filter functions as a bandpass filter. The frequency f up at the high-frequency end of the pass band is the frequency f as of the anti-resonance point (anti-resonance frequency) of the series resonator S 22 . The frequency f down of the low-frequency end of the pass band is the frequency of the resonance point (resonance frequency) f rp of the parallel resonator P 22 .

A ladder filter having an attenuation pole at a harmonic is now described, and a description of a multi-stage ladder filter is followed. FIG. 3A is a circuit diagram of a multi-stage ladder filter, and FIG. 3B is a diagram of an attenuation characteristic of the multi-stage ladder filter. The horizontal axis of FIG. 3B is the frequency, and the vertical axis thereof is the amplitude.

As illustrated in FIG. 3A , a ladder filter F 11 is composed of series resonators S 1 , S 2 and S 3 , parallel resonators P 1 and P 2 , and inductors L 1 and L 2 . The series resonators S 1 , S 2 and S 3 are connected in series to each other between the input terminal In and the output terminal Out. The series resonator S 1 is connected to the input terminal In, and the series resonator S 3 is connected to the output terminal Out.

One end of the parallel resonator P 1 is connected to a node between the series resonators S 1 and S 2 . One end of the parallel resonator P 2 is connected to a node between the series resonators S 2 and S 3 . The other end of the parallel resonator P 1 and one end of the inductor L 1 are connected in series to each other, and the other end of the inductor L 1 is grounded. The other end of the parallel resonator P 2 and one end of the inductor L 2 are connected in series to each other, and the other end of the inductor L 2 is grounded. The resonators function as capacitors outside of the pass band. Thus, the parallel resonator P 1 and the inductor L 1 function as an LC resonance circuit, and the parallel resonator P 2 and the inductor L 2 function as an LC resonance circuit. It is assumed that the capacitance of each of the parallel resonators P 1 and P 2 is denoted as C p , and the inductance of the inductor L 1 is denoted as L 1 . In this case, the resonance frequency f 1 of the LC resonance circuit formed by the parallel resonator P 1 and the inductor L 1 is expressed as follows:

f 1 = 1 2 ⁢ π ⁢ L 1 ⁢ C P

The resonance frequency f 2 of the LC resonance circuit formed by the parallel resonator P 2 and the inductor L 2 is described in a similar way to the above expression.

As illustrated in FIG. 3B , attenuation poles are formed at the frequencies f 1 and f 2 . By adjusting the capacitance value of the parallel resonator and the inductance value of the inductor, the frequencies at which the attenuation poles appear can be adjusted. For example, the frequencies f 1 and f 2 may be selected so as to be respectively included in a frequency band corresponding to the second harmonic of the pass band of the filter and a frequency band corresponding to the third harmonic thereof. It is thus possible to form the attenuation poles in the second and third harmonics.

Although the ladder filter F 11 has two parallel arms, more attenuation poles may be formed by employing more parallel arms and associated inductors. However, an increased number of parallel arms may increase the signal loss in the pass band, and may prevent downsizing of the filter. Further, in a case where an attenuation pole is formed in a low-frequency range, the inductors L 1 and L 2 may be varied to have larger inductance values. For example, in a case where an attenuation pole is formed at the low-frequency side of the pass band (left-hand side in FIG. 3B ), inductance values as large as 1˜5 nH may be used. The use of inductors having large inductance values may prevent downsizing of the ladder filter. At the high-frequency side of the pass band, the inductor L 1 or L 2 may function as a choke coil, and sufficient suppression may not be obtained at frequencies higher than those of the pass band. As described above, it is difficult to realize high suppression over the broad band.

A comparative example is now described. A duplexer 100 R in accordance with the comparative example corresponds to, for example, the W-CDMA (Wideband Code Division Multiple Access) Band 2 system. The pass band of a transmission filter F 11 overlaps the transmission band of the W-CDMA Band 2 system that ranges from 1850 MHz to 1910 MHz. The pass band of a reception filter F 12 overlaps the reception band of the W-CDMA Band 2 system that ranges from 1930 MHz to 1990 MHz. For example, a system in which the transmission band and the reception band are close to each other, such as the W-CDMA Band 2 system, may be designed to generate an attenuation pole around the high-frequency end of the pass band of the transmission filter to improve the sharpness, whereby high suppression is ensured in the pass band of the reception filter. It is thus possible to suppress the interference between the filters. In order to generate the attenuation pole, an inductor may be connected to the parallel arm, as illustrated in FIG. 3A .

FIG. 4 is a circuit diagram of a duplexer with the ladder filter of the comparative example. FIG. 5A illustrates an attenuation characteristic of the ladder filter of the comparative example, and FIG. 5B illustrates an enlarged view of a pass band illustrated in FIG. 5A .

Referring to FIG. 4 , the duplexer 100 R of the comparative example includes the transmission filter F 11 , the reception filter F 12 , an antenna terminal Ant, a transmission terminal Tx, reception terminals Rx 1 and Rx 2 , and inductors La and Lr. The inductor La, the transmission filter F 11 and the reception filter F 12 are connected in parallel with the antenna terminal Ant.

One end of the reception filter F 12 is connected to the antenna terminal Ant and one end of the transmission filter F 11 , and the other end thereof is connected to the two reception terminals Rx 1 and Rx 2 . The inductor Lr is connected between the reception terminals Rx 1 and Rx 2 . The inductors La and Lr function as those for impedance matching. The reception filter F 12 may be a ladder filter.

The transmission filter F 11 is the ladder filter of the comparative example. The transmission filter F 11 is composed of series resonators S 11 ˜S 13 , S 2 , S 3 , S 41 ˜S 42 , S 5 and S 6 , and parallel resonators P 1 ˜P 3 , and inductors L 1 , L 2 and Lg. The series resonators S 11 ˜S 6 are connected in series to each other. One end of the series resonator S 11 is connected to the antenna terminal Ant and one end of the reception filter F 12 . One end of the series resonator S 6 is connected in series to the transmission terminal Tx.

One end of the parallel resonator P 1 is connected in parallel with a node between the series resonator S 13 and the series resonator S 2 , and the other end thereof is connected in series to one end of the inductor L 1 . One end of the parallel resonator P 2 is connected in parallel with a node between the series resonator S 3 and the series resonator S 41 . One end of the parallel resonator P 3 is connected in parallel with a node between the series resonator S 42 and the series resonator S 5 . The other end of the parallel resonator P 2 and the other end of the parallel resonator P 3 are connected in parallel with one end of the inductor L 2 . The other end of the inductor L 1 and the other end of the inductor L 2 are connected in parallel with one end of the inductor Lg. The other end of the inductor Lg is grounded. Next, a description is given of the capacitances and resonance frequencies of the resonators included in the ladder filter of the comparative example (transmission filter F 11 ).

Table 1 indicates the capacitance values and resonance frequencies of the resonators of the ladder filter of the first comparative example.

As indicated in Table 1, the capacitance values of the series resonators S 11 ˜S 13 are 3.879 pF, and the resonance frequencies thereof are 1881 MHz. The capacitance value of the series resonator S 2 is 7.8 pF, and the resonance frequency thereof is 1968 MHz. The capacitance of the series resonator S 3 is 1.67 pF, and the resonance frequency thereof is 1901 MHz. The capacitance values of the series resonators S 41 and S 42 are 1.491 pF, and the resonance frequencies thereof are 1889 MHz. The capacitance value of the series resonator S 5 is 6.49 pF, and the resonance frequency thereof is 1876 MHz. The capacitance value of the series resonator S 6 is 9.23 pF, and the resonance frequency thereof is 1881 MHz. The capacitance value of the parallel resonator P 1 is 2.317 pF, and the resonance frequency thereof is 1816 MHz. The capacitance value of the parallel resonator P 2 is 1.979 pF, and the resonance frequency thereof is 1819 MHz. The capacitance value of the parallel resonator P 3 is 2.243 pF, and the resonance frequency thereof is 1820 MHz. The inductance values of the inductors L 1 , L 2 and Lg are 0.5˜2 nH, for example.

As described above, outside of the pass band, the resonator functions as a capacitor. By adjusting the capacitance of the resonator, the attenuation characteristic may be varied over a broader band. For example, the amount of attenuation may be increased in a broader band by increasing the ratio C p /C s where C s is the combined capacitance of the series resonator and C p is the combined capacitance of the parallel resonator. Such a resonator having increased suppression is referred to as a second comparative example.

Table 2 illustrates exemplary capacitance values and resonance frequencies of the resonators of a ladder filter in accordance with the second comparative example. The circuit configuration of the ladder filter of the second comparative example is the same as illustrated in FIG. 4 , and a description thereof is omitted here.

As indicated in Table 2, the capacitance values of the series resonators S 1 ˜S 3 are 3.618 pF, and the capacitance value of the series resonator S 2 is 7.43 pF. The capacitance of the series resonator S 3 is 1.558 pF, and the capacitance values of the series resonators S 41 and S 42 are 1.393 pF. The capacitance value of the series resonator S 5 is 6.05 pF, and the capacitance value of the series resonator S 6 is 8.93 pF. The capacitance value of the parallel resonator P 1 is 2.485 pF. The capacitance value of the parallel resonator P 2 is 2.122 pF. The capacitance value of the parallel resonator P 3 is 2.405 pF. The resonance frequencies of the resonators are the same as those illustrated in Table 1. The ratio C p /C s of the combined capacitance of the parallel resonators P 1 ˜P 3 to that of the series resonators S 1 ˜S 6 of the second comparative example is approximately 10% greater than that of the first comparative example.

A description is now given of results of calculation of the attenuation characteristics of the first and second comparative examples. FIG. 5A is a diagram of attenuation characteristics of the ladder filters in accordance with the first and second comparative examples, and FIG. 5B is an enlarged view of pass bands of the first and second comparative examples. In FIGS. 5A and 5B , the horizontal lines are the frequency, and the vertical lines are the amount of attenuation. Solid lines are the attenuation characteristics of the ladder filter in accordance with the first embodiment, and dotted lines are the attenuation characteristics of the ladder filter in accordance with the second embodiment. In FIG. 5A , frequency ranges A˜E are defined by broken lines. The frequency range A is a band used for a portable telephone system. The frequency range B is a band used for GPS (Global Positioning System). The frequency range C is a band used for a wireless LAN (Local Area Network) and Bluetooth (BT). The frequency range D is a band of second harmonics of the transmission filter F 11 . The frequency range E is a band of third harmonics of the transmission filter F 11 . A range defined by broken lines in FIG. 5B is a transmission band of the W-CDMA band 2 system.

As illustrated in FIG. 5A , the first and second comparative examples have attenuation poles generated around the high-frequency ends of the pass bands. This is because the inductors L 1 , L 2 and Lg having high inductance values are connected to the parallel arms. In the first comparative example, the degree of suppress deteriorates at frequencies lower than and higher than those of the pass band. Thus, the first comparative example has a difficulty in realizing a high degree of suppression over broad bands. The second comparative example has a higher degree of suppression than that of the first comparative example without increasing the number of attenuation poles. This is because the capacitance ratio C p /C s is increased, as illustrated in Table 2.

However, as illustrated in FIG. 5B , the second comparative example has a deterioration of about 0.15˜0.2 dB even in the pass band. Thus, the second comparative example has a difficulty in realizing both high suppression over broad bands and low insertion loss. Further, the first and second comparative examples, the suppression is easily degraded at high frequencies. This is because the inductors L 1 , L 2 and Lg used to form the attenuation poles function as choke coils. The behaviors of the inductors L 1 , L 2 and Lg as choke coils are conspicuous when these inductors have large inductance values. As has been described, in the duplexers or modules that conform to a system in which the reception band and the transmission band are close to each other like the W-CDMA band 2 system, an attenuation pole may be formed in the pass band of the other filter. When the inductors having large inductance values are used to form the attenuation pole, there is a high possibility that realization of high suppression over broad bands may have a difficulty.

A description is now given of a ladder filter in which an inductor is connected to a series arm. FIG. 6 is a circuit diagram of a duplexer using a ladder filter with an inductor being connected to a series arm. A description of parts similar to those in FIG. 4 is omitted here.

As illustrated in FIG. 6 , a transmission filter F 13 of a duplexer 100 L is configured to replace the aforementioned series resonator S 6 with a series resonator Sx connected in series and to have an inductor Ls connected in series to the series resonator Sx. One end of the series resonator Sx is connected in series to the series resonator S 5 , and the other end thereof is connected in series to one end of the inductor Ls. The other end of the inductor Ls is connected in series to the transmission terminal Tx. The series resonator Sx is an additional resonator connected in series between the series resonators S 11 ˜S 5 and the transmission terminal Tx. The inductor Ls is connected in series to the series resonator Sx and the transmission terminal Tx. The transmission terminal Tx corresponds to an input terminal of the transmission filter F 13 , and the antenna terminal Ant corresponds to an output terminal of the transmission filter F 13 . Next, the reactance characteristic of the series resonator Sx is described.

FIG. 7 is a diagram of a reactance characteristic of the series resonator Sx. The horizontal axis is the frequency, and the vertical axis is the reactance. Referring to FIG. 7 , the reactance is zero at the resonance frequency f rx . At the anti-resonance frequency f ax , the reactance diverges. In a range G 1 equal or lower than the resonance frequency f rx and a range G 2 equal to or higher than the anti-resonance frequency f ax , the series resonator Sx functions as a capacitor. In a range G 3 between the resonance frequency f rx and the anti-resonance frequency f ax , the series resonator Sx functions as an inductor. Now, a description is given of the capacitance and the resonance frequency of the resonators and the inductance value of the inductor included in the transmission filter F 13 .

Table 3 indicates the capacitance values and resonance frequencies of the resonators included in the transmission filter F 13 .

As indicated in FIG. 3 , the capacitance values and resonance frequencies of the series resonators S 11 ˜S 42 are the same as those of the comparative example 1 indicated in Table 1. The capacitance value of the series resonator S 5 is 4.964 pF, and the resonance frequency thereof is 1870 MHz. The capacitance values and resonance frequencies of the parallel resonators P 1 ˜P 3 are the same as those of the second comparative example indicated in Table 2. The capacitance value of the series resonator Sx is 3.2 pF, and the resonance frequency thereof is 2400 MHz. The inductance value of the inductor Ls is 2.2 nH, and the Q value thereof is 40.

In the transmission filter F 13 , the series resonator S 2 has the highest anti-resonance frequency among the series resonators S 11 ˜S 5 except the series resonator Sx. As indicated in Table 3, the resonance frequency f rx of the series resonator Sx is 2400 MHz, which is higher than the anti-resonance frequency of the series resonator S 2 . In other words, the resonance frequency f rx of the series resonator Sx is higher than the anti-resonance frequencies of the series resonators S 11 ˜S 5 . As indicated in FIG. 2B , the frequency of the high-frequency end of the pass band of the ladder filter is defined by the anti-resonance frequency of the series resonator. Thus, the resonance frequency fr x is located at the high-frequency side of the pass band of the transmission filter F 13 . In this case, the pass band of the transmission filter F 13 corresponds to the range G 1 illustrated in FIG. 7 . Therefore, the series resonator Sx functions as a capacitance in the pass band of the transmission filter F 13 . As a result, the series resonator Sx and the inductor Ls function as an LC resonance circuit in the passband.

It is assumed that the capacitance of the series resonator Sx is Cx and the inductance of the inductor Ls is Lx. The resonance frequency f lc of an LC resonance circuit LC 1 formed by the series resonator Sx and the inductor Ls is expressed as follows:

f lc = 1 2 ⁢ π ⁢ L x ⁢ C x

The resonance frequency f lc of the LC resonance circuit LC 1 is higher than the frequency f down of the low-frequency end of the pass band and is lower than the frequency f up of the high-frequency end thereof. That is, the following relationship stands:

f down <1/(2π( C x ·L x ) 1/2 )< f up

Next, a description is given of results of calculation of the attenuation characteristic of the transmission filter F 13 . FIG. 8A is a diagram of an exemplary attenuation characteristic of the transmission filter F 13 , and FIG. 8B is an enlarged view of the pass band in FIG. 8A . Solid lines are the attenuation characteristic of the transmission filter F 13 , and broken lines are the attenuation characteristic of the ladder filter of the first comparative example. Dotted lines are the attenuation characteristics of the ladder filter of the second comparative example.

As illustrated in FIG. 8A , the transmission filter F 13 has high suppression over a broad band including the frequency ranges A through E. As illustrated in FIG. 8B , the transmission filter F 13 has good characteristics that are as good as or better than those of the transmission filter F 11 of the first comparative example in the pass band, particularly, at the edges of the reception bands of the W-CDMA Band 2 system (around 1850 MHz and around 1910 MHz).

A description is given of a mechanism obtained from the attenuation characteristics illustrated in FIGS. 8A and 8B . As has been described, the resonance frequency f rx of the series resonator Sx is higher than the resonance frequencies of the series resonators S 11 ˜S 5 . In other words, the resonance frequency f rx is located at the high-frequency side of the pass band of the transmission filter F 13 . Thus, the series resonator Sx functions as a capacitor in the pass band of the transmission filter F 13 . The series resonator Sx and the inductor Ls function as the LC resonance circuit LC 1 in the pass band. From the relationship f down <1/(2π(C x ·L x ) 1/2 )<f up described above, the insertion loss in the pass band can be canceled by the resonance of the LC resonance circuit. The resonance frequency f lc may be set around the center frequency of the pass band, for example.

In the frequency range lower than the pass band, the LC resonance circuit LC 1 functions as a capacitor, and the capacitance ratio Cp/Cs of the transmission filter F 13 has a larger value. Thus, in the low-frequency range, high suppression is available. In the frequency range higher than the pass band, the LC resonance circuit LC 1 functions as an inductor, and plays a role of choke coil. Thus, high suppression is available even in the high-frequency range. Attenuation poles are generated around the high-frequency end of the pass band. As described above, the transmission filter F 13 is capable of generating the attenuation poles and realizing high suppression over the broad band while suppressing increase in the insertion loss in the pass band.

As has been described, in order to have the series resonator Sx and the inductor Ls function as an LC resonance circuit in the pass band of the transmission filter F 13 , the series resonator Sx is required to function as a capacitor in the pass band. Therefore, the resonance frequency f rx of the series resonator Sx is higher than or lower than the pass band. However, as has been described with reference to FIG. 7 , the series resonator Sx functions as an inductor in the frequency range G 3 between the resonance frequency f rx and the anti-resonance frequency f ax . Thus, even when the resonance frequency f rx of the series resonator Sx is lower than the pass band, in a case where the pass band and the frequency range G 3 overlap with each other, it may be difficult for the series resonator Sx to function as a capacitor. It is therefore preferable that the anti-resonance frequency f ax of the series resonator Sx is lower than the resonance frequency of the parallel resonator of the transmission filter F 13 . In other words, the anti-resonance frequency f ax is preferably lower than the frequency at the low-frequency end of the pass band. In this case, the pass band corresponds to the frequency range G 2 in FIG. 7 , and the series resonator Sx functions as a capacitor.

A description is now given of an example in which surface acoustic wave (SAW) resonators are used. FIG. 9 is a plan view of an exemplary SAW resonator. The view of FIG. 9 is schematic, and the SAW resonator is not limited to the number of electrode fingers illustrated in FIG. 9 .

Referring to FIG. 9 , a SAW resonator 10 includes a piezoelectric substrate 12 , an IDT (InterDigital Transducer) 14 and two reflectors 16 . The IDT 14 and the reflectors 16 are provided on the piezoelectric substrate 12 . The IDT 14 includes a pair of comb-like electrodes 14 a and 14 b that are interleaved with each other, and excites acoustic waves. The two reflectors 16 are arranged at both sides of the IDT 14 in directions in which the acoustic waves are propagated. The wavelength λ of the acoustic waves depends on the pitch between the electrode fingers of the IDT 14 , and may be 1.62 μm, for example. The aperture length W of the interleaved electrode fingers of the IDT 14 may be 19.9λ, for example. The number of pairs of electrode fingers of the IDT 14 may be 59.5, for example. The capacitance of the SAW resonator may be adjusted by varying the pitch between the electrode fingers or the aperture length W of the interleaved electrode fingers. The piezoelectric substrate 12 may be made of a piezoelectric substance such as lithium tantalate (LiTaO 3 ) or lithium niobate (LiNbO 3 ), for example. The IDT 14 and the reflectors 16 may be made of a metal such as aluminum (Al).

A description is now given of the pass characteristic and Q value of the SAW resonator 10 . The Q value of the SAW resonator is defined as a Q value obtained when the SAW resonator 10 functions as a capacitor.

FIG. 10A is a diagram of an exemplary pass characteristic of the SAW resonator, and FIG. 10B is a diagram of an exemplary Q value thereof. In FIGS. 10A and 10B , the horizontal axis is the frequency, and the vertical axis of FIG. 10A is the amplitude, while the vertical axis of FIG. 10B is the Q value.

As illustrated in FIG. 10A , the resonance frequency f r of the SAW resonator 10 is 2332 MHz, and the amplitude is the largest at the resonance frequency f r . The anti-resonance frequency f a is 2405 MHz, and the amplitude is the smallest at the anti-resonance frequency f a . As illustrated in FIG. 10B , the Q value is zero around the resonance frequency f r . At the high-frequency side of the resonance frequency f r , the Q value is about 10. At the high-frequency side of the resonance frequency, there are bulk waves that are excited by the IDT 14 and are emitted in the piezoelectric substrate 12 . The bulk waves result in loss and causes a decrease in the Q value.

In contrast, the Q value increases at the low-frequency side of the resonance frequency fr. As illustrated in a left-hand side in FIG. 10B , the Q value is equal to or larger than 40 at frequencies lower than f q , where fq is 2050 MHz. Thus, the frequency ratio f r /f q =1.138 and f r =1.138×f q

A case is considered where the SAW resonator 10 is employed as the series resonator Sx. As has been described, in the transmission filter F 13 illustrated in FIG. 6 , the series resonator Sx functions as a capacitor in the pass band of the transmission filter F 13 . As the inductor Ls illustrated in FIG. 6 , a chip inductor may be used. The Q value of the chip inductor included in the ladder filter is about 40, for example. In order to prevent the characteristics of the ladder filter from deteriorating, the Q value of the series resonator Sx is equal to or larger than 40.

As has been described with reference to FIG. 10B , the Q value is equal to or larger than 40 at the low-frequency side of the resonance frequency f q . It is therefore preferable that the frequency f up of the high-frequency end of the pass band of the transmission filter F 13 is lower than f q , that is, lower than 2050 MHz. In other words, it is preferable that the resonance frequency f rx of the series resonator Sx meets the following condition with respect to the high-frequency end of the pass band of the transmission filter F 13 :

f rx >1.138 ×f up

Thus, the range on the left-hand side of FIG. 10B corresponds to the pass band. The frequency f up may be equal to or lower than f q .

As described above, the series resonator Sx functions as a capacitor in the pass band, and forms the LC resonance circuit LC 1 together with the inductor Ls. It is thus possible to obtain high suppression over the broad bands. Since the resonance frequency f lc of the LC resonance circuit LC 1 is located within the pass band, the loss in the pass band can be reduced. The resonance frequency f rx of the series resonator Sx may be either higher than the anti-resonance frequency of the series resonator or lower than the resonance frequency of the parallel resonator. However, in the case where the resonator with the IDTs is used for the filter, it is preferable that the resonance frequency f rx is higher than the anti-resonance frequency of the series resonator and satisfies the condition f rx >1.138×f up .

First Embodiment

Embodiments according to the aspects described above are described below. A first embodiment uses the SAW resonators. FIG. 11A is a block diagram of an exemplary duplexer in accordance with the first embodiment, and FIG. 11B is a block diagram of an RF module equipped with the duplexer in accordance with the first embodiment.

Referring to FIG. 11A , a duplexer 100 has a reception filter 100 a and a transmission filter 100 b . The reception filter 100 a and the transmission filter 100 b are connected to a common terminal (antenna terminal) 102 . The common terminal 102 is connected to an antenna 104 . The reception filter 100 a receives a signal from the antenna 104 , and the filtered signal is applied to an amplifier, for example. The transmission filter 100 b filters a signal from an amplifier, for example, and the filtered signal is applied to the antenna 104 via which the signal is transmitted.

Referring to FIG. 11B , an RF module 110 is equipped with an antenna 104 , an antenna switch 112 , a duplexer bank 114 and an amplifier module 116 . The RF module 110 corresponds to a plurality of communication systems such as the GSM (Global System for Mobile Communication) system and the W-CDMA system, and is installed in the portable telephone or the like. The GSM system handles the 850 MHz band (GSM 850), 900 MHz band (GSM 900), 1800 MHz band (GSM 1800), and 1900 MHz band (GSM 1900). The antenna 104 is capable of handling the GSM system and the W-CDMA system. The duplexer bank 114 includes a plurality of duplexers 114 a , 114 b and 114 c . Each of the duplexers corresponds to one of the communication systems. The duplexer bank 114 may have two duplexers or more than three duplexers. The antenna switch 112 selects the duplexer that conforms to the communication system used for signal transmission and reception from among the duplexers of the duplexer bank 114 , and makes a connection between the selected duplexer and the antenna 104 . The duplexers 114 a , 114 b and 114 c of the duplexer bank 114 are connected to the amplifier module 116 . The amplifier module 116 amplifies the signals received by the reception filters of the duplexers 114 a ˜ 114 c , and outputs the amplified signal to a next-stage processing part. Further, the amplifier module 116 amplifiers signals generated by a processing part (not illustrated), and outputs the amplified signals to the transmission filters of the duplexers 114 a ˜ 114 c.

A description is given of the configuration of the duplexer in accordance with the first embodiment. FIG. 12 is a circuit diagram of the configuration of the duplexer in accordance with the first embodiment.

Referring to FIG. 12 , the circuit configuration of the duplexer 100 in accordance with the first embodiment is the same as illustrated in FIG. 6 . The duplexer 100 has a duplexer 101 , which includes structural components other than the inductors La, Lr and Ls. The duplexer chip 101 includes a reception filter chip 100 c and a transmission filter chip 100 d . The detailed configuration of the duplexer chip 101 will be described later. Structural components of the transmission filter other than inductors L 1 , L 2 , Lg and Ls are realized as a transmission filter chip 100 d . The inductors La, Lr and Ls are realized by chip inductors.

The resonance frequencies and capacitance values of the resonators are the same as those described in Table 3. The resonance frequency f rx of the series resonator Sx is higher than the anti-resonance frequencies of the series resonators S 11 ˜S 5 . The resonance frequency f rx satisfies the aforementioned condition f rx >1.138×f up . The inductance value of the inductor Ls may be 2.2 nH, and the Q value thereof may be 40, for example.

FIG. 13A is a plan view of the duplexer chip 101 , and FIG. 13B is a cross-sectional view taken along a line A 1 -A 1 illustrated in FIG. 13A . FIG. 13C is a plan view of the duplexer chip 101 .

As illustrated in FIGS. 13A and 13B , the duplexer chip 101 includes the reception filter chip 100 c , the transmission filter chip 100 d , a package substrate 120 and a seal member 122 . The view of FIG. 13A is seen through the seal member 122 . As illustrated in FIG. 13B , the reception filter chip 100 c and the transmission filter chip 100 d are flip-chip mounted on the package substrate 120 by bumps 124 . The package substrate 120 has a two-layer structure composed of a first layer 120 - 1 and a second layer 120 - 2 located below the first layer 120 - 1 . Interconnection layers 120 a , 120 b and 120 c are formed on the upper, intermediate and lower surfaces of the package substrate 120 , which are connected together by via interconnections 26 . The interconnection layer 120 b is located between the first layer 120 - 1 and the second layer 120 - 2 . The interconnection layer 120 c functions as footpads. Some interconnections provided on or in the package substrate 120 function as inductors. The seal member 122 may be made of an insulative material such as epoxy resin or solder, and seals the reception filter chip 100 c and the transmission filter chip 100 d . The package substrate 120 may be made of an insulative material such as ceramic. The interconnection layers 120 a ˜ 120 c are made of a metal such as aluminum. The bumps 124 may be made of a metal such as gold.

As illustrated in FIG. 13C , the duplexer chip 101 is flip-chip mounted on a printed-circuit board 130 via the interconnection layer 120 c . The inductors Lr, La and Ls, which are chip inductors, are mounted on the printed-circuit board 130 . Although not illustrated, the antenna terminal Ant, the reception terminals Rx 1 and Rx 2 and the transmission terminal Tx are provided on the printed-circuit board 130 . The duplexer chip 101 and the inductor Lr are connected together by interconnection lines 131 r - 1 and 131 r - 2 . The interconnection line 131 r - 1 is connected to the reception terminal Rx 1 . The interconnection line 131 r - 2 is connected to the reception terminal Rx 2 . The duplexer chip 101 and the inductor La are connected together via an interconnection line 131 a . The interconnection line 131 a is connected to the antenna terminal Ant. The duplexer chip 101 and the inductor Ls are connected together by an interconnection line 131 s . The interconnection line 131 s is connected to the transmission terminal Tx.

In FIG. 13C , the printed-circuit board 130 may be replaced with a module board. The RF module 110 illustrated in FIG. 11B is formed by mounting the duplexer chips and the chip inductors on the printed-circuit board 130 or the module board.

The filters of the duplexer are now described. FIG. 14 is a perspective view of the transmission filter chip 100 d of the duplexer configured in accordance with the first embodiment.

Referring to FIG. 14 , the transmission filter chip 100 d is composed of the piezoelectric substrate 12 , the series resonators S 11 ˜Sx, the parallel resonators P 1 ˜P 3 , the antenna terminal Ant 1 , the transmission terminal Tx 1 and terminals 18 a and 18 b . The series resonators S 11 ˜Sx and the parallel resonators P 1 ˜P 3 are SAW resonators in which the IDT and the reflectors of each SAW filter are arranged on the piezoelectric substrate 12 . The transmission filter chip 100 d is flip-chip mounted so that the surface of the piezoelectric substrate 12 on which the SAW resonators are formed and the upper surface of the package substrate 120 face each other. As illustrated in FIG. 13B , since a spacing is formed between the transmission filter chip 100 d and the package substrate 120 , the acoustic waves are excited without any obstacles.

One of the comb-like electrodes of the series resonator S 11 is connected to the antenna terminal Ant 1 . The antenna terminal Ant 1 is connected to the antenna 104 illustrated in FIG. 11A or 11 B. One of the comb-like electrodes of the parallel resonator P 1 is connected to the terminal 18 a . The terminal 18 a is connected to the inductor L 1 illustrated in FIG. 12 . One of the comb-like electrodes of the parallel resonator P 2 and one of the comb-like electrodes of the parallel resonator P 3 are connected to the terminal 18 b . The terminal 18 b is connected to the inductor L 2 illustrated in FIG. 12 . One of the comb-like electrodes of the series resonator Sx is connected to the transmission terminal Tx 1 . The transmission terminal Tx 1 is connected to the transmission terminal Tx via the inductor Ls illustrated in FIGS. 12 and 13C . Each terminal is used for flip-chip mounting on the package substrate 120 illustrated in FIG. 13B . Circular shapes on the solid patterns are areas in which the bumps 124 are connected.

The package substrate 120 is now described. FIGS. 15A through 15C are respectively plan views of the package substrate 120 . FIG. 15A illustrates the upper surface of the package substrate 120 . FIG. 15B illustrates the plan view of the second layer 120 - 2 of the package substrate 120 seen through the first layer 120 - 1 . FIG. 15C is a view seen through the first layer 120 - 1 and the second layer 120 - 2 .

As illustrated in FIG. 15A , the interconnection layer 120 a is provided on the first layer 120 - 1 . The interconnection layer 120 a includes an antenna terminal Ant 2 , a transmission terminal Tx 2 , interconnection lines 20 a , 20 b and 20 c , reception terminals Rx 1 a and Rx 2 a , and a ground terminal GND 1 . The transmission filter chip 100 d is mounted in an area surrounded by a broken line in FIG. 15A . The antenna terminal Ant 2 of the interconnection layer 120 a is connected to the antenna terminal Ant 1 of the transmission filter chip 100 d . The transmission terminal Tx 2 of the interconnection layer 120 a is connected to the transmission terminal Tx 1 of the transmission filter chip 100 d . The interconnection line 20 a of the interconnection layer 120 a is connected to the terminal 18 a of the transmission filter chip 100 d . The interconnection line 20 b of the interconnection layer 120 a is connected to the terminal 18 b of the transmission filter chip 100 d . The interconnection lines 20 a and 20 b are connected to the interconnection line 20 c . The interconnection line 20 a contributes to the generation of the inductor L 1 illustrated in FIG. 12 . The interconnection line 20 b contributes to the generation of the inductor L 2 . The interconnection line 20 c contributes to the generation of the inductor Lg. As described above, the inductors L 1 , L 2 and Lg are provided on the package substrate 120 . The reception terminals Rx 1 a and Rx 2 a are connected to reception terminals included in the footpads of the reception filter chip 100 c . The ground terminal GND 1 is connected to a ground terminal included in the footpads of the reception filter chip 100 c.

Referring to FIG. 15B , the interconnection layer 120 b is provided on the upper surface of the second layer 120 - 2 . The interconnection layer 120 b includes an antenna pattern Ant 3 , a transmission pattern Tx 3 , reception patterns Rx 1 b and Rx 2 b , and ground patterns GND 2 b and GND 3 . As has been described with reference to FIG. 12B , the interconnection layers 120 a and 120 b are connected together by the via interconnections 26 . Specifically, the antenna pattern Ant 3 of the interconnection layer 120 b is connected to the antenna terminal Ant 2 of the interconnection layer 120 a . The transmission pattern Tx 3 of the interconnection layer 120 b is connected to the transmission terminal Tx 2 of the interconnection layer 120 a . The ground pattern GND 2 b of the interconnection layer 120 b is connected to the interconnection line 20 c of the interconnection layer 120 a . The reception pattern Rx 1 b of the interconnection layer 120 b is connected to the reception terminal Rx 1 a of the interconnection layer 120 a . The reception pattern Rx 2 b of the interconnection layer 120 b is connected to the reception terminal Rx 2 a of the interconnection layer 120 a . The ground pattern GND 3 of the interconnection layer 120 b is connected to the ground terminal GND 1 of the interconnection layer 120 a.

Referring to FIG. 15C , the interconnection layer 120 c is provided on the lower surface of the second layer 120 - 2 . The interconnection layer 120 c includes an antenna terminal Ant 4 , a transmission terminal Tx 4 , reception terminals Rx 1 c and Rx 2 c , and ground terminals GND 2 c , GND 2 d , GND 4 , GND 5 and GND 6 . As has been described with reference to FIG. 12B , the interconnection layers 120 b and 120 c are connected together by the via interconnections 26 . Specifically, the antenna terminal Ant 4 of the interconnection layer 120 c is connected to the antenna pattern Ant 3 of the interconnection layer 120 b . The transmission terminal Tx 4 of the interconnection layer 120 c is connected to the transmission pattern Tx 3 of the interconnection layer 120 b . The ground terminals GND 2 c and GND 2 d of the interconnection layer 120 c are connected to the ground pattern GND 2 b of the interconnection layer 120 b . The reception terminal Rx 1 c of the interconnection layer 120 c is connected to the reception pattern Rx 1 b of the interconnection layer 120 b . The reception terminal Rx 2 c of the interconnection layer 120 c is connected to the reception pattern Rx 2 b of the interconnection layer 120 c . The ground terminals GND 4 , GND 5 and GND 6 of the interconnection layer 120 c is connected to the ground pattern GND 3 of the interconnection layer 120 b.

The transmission filter 100 b of the duplexer 100 in accordance with the first embodiment includes the series resonators S 11 ˜S 5 , the parallel resonators P 1 ˜P 3 , the series resonator Sx (additional resonator), and the inductor Ls connected in series to the series resonator Sx. The resonance frequency f rx of the series resonator Sx is higher than the anti-resonance frequencies of the series resonators S 11 ˜S 5 . Thus, the series resonator Sx functions as a capacitor, and forms the LC resonance circuit LC 1 together with the inductor Ls. As a result, as illustrated in FIGS. 8A and 8B , the transmission filter 100 b realizes high suppression over the broad bands. The resonance frequency of the LC resonance circuit LC 1 is located in the pass band of the transmission filter 100 b . Thus, the insertion loss in the pass band can be reduced.

The anti-resonance frequency fax of the series resonator Sx may be lower than the resonance frequencies of the parallel resonators P 1 ˜P 3 . However, in the case where the resonator is the SAW resonator as illustrated in FIG. 14 , the resonance frequency f rx satisfies the aforementioned condition f rx >1.138×f up , the series resonator Sx functions as a capacitor having a Q value of 40 or more. Thus, deterioration of the frequency characteristic is suppressed. As described above, in the case where the SAW resonator is used, it is preferable that the resonance frequency f rx is higher than the anti-resonance frequencies of the series resonators S 11 ˜S 5 . Further, it is preferable that the aforementioned condition f rx >1.138×f up is satisfied. In a case where the resonator is not the SAW resonator but another type of resonator using IDT such as a Love wave resonator, a Lamb wave resonator, a boundary acoustic wave resonator, a high Q value is obtained by meeting the aforementioned condition f rx >1.138×f up . The resonator may be a resonator that does not use IDT, such as a piezoelectric thin-film resonator.

In the first embodiment, the series resonator Sx that functions as a capacitor is incorporated in the ladder filter. Thus, it is possible to realize compact ladder filters, duplexers and RF modules, as compared with those to which a capacitance element is externally connected. The inductors L 1 , L 2 and Lg are formed by the interconnection layers 120 a ˜ 120 c in the package substrate 120 . Thus, the duplexer 100 and the RF module 110 may be downsized. The inductors La, Ls and Lr may be formed by the interconnection layers 120 a ˜ 120 c . Thus, the duplexer 100 and the RF module 110 may further be downsized.

By using the inductors La and Lr, the attenuation poles are formed around the high-frequency end of the pass band of the transmission filter 100 b , as illustrated in FIG. 8A . Thus, the transmission filter 100 b has high suppression in the pass band of the reception filter 100 a , whereby interference between the transmission filter 100 b and the reception filter 100 a can be suppressed. Generally, among a first filter of the duplexer and a second filter having a pass band at the high-frequency side of the first filter, namely, having frequencies higher than the frequency at the high-frequency end of the first filter, the first filter is preferably the SAW filter. The first filter may be the transmission filter or the reception filter, which may be selected in accordance with the communication system, for example. The ladder filter of the first embodiment may be applied to both the filters of the duplexer or may be applied to the second filter. In other words, the duplexer has at least ladder filter of the first embodiment.

The RF module includes at least one ladder filter of the first embodiment. The module of the first embodiment is not limited to the RF module but may be another type of module. The module that includes multiple filters on the single module board and utilizes multiple bands is capable of ensuring suppression over the board bands and having improved characteristics.

As illustrated in FIGS. 11A and 11B , the first embodiment is not limited to the aforementioned duplexer and the module with the duplexer but may be a ladder filter alone. The ladder filter is not limited to the multiple-stage configuration but may have only a single stage. The ladder filter includes one or multiple series resonators and one or multiple parallel resonators. The resonance frequency f rx of the series resonator Sx is higher than the anti-resonance frequencies of one or the multiple series resonators. The anti-resonance frequency f ax of the series resonator Sx is lower than the resonance frequencies of one or the multiple parallel resonators. When these conditions are satisfied, the series resonator Sx functions as a capacitor in the pass band of the ladder filter. The series resonator Sx (additional resonator) and the inductor Ls may be provided between the series resonator and the antenna terminal Ant (output terminal).

The present invention is not limited to the specifically disclosed embodiments but may include other embodiments and variations without departing from the scope of the present invention.