Switching power supply device and lighting apparatus

This application is based upon and claims the benefit of priority from Japanese Patent Application No. 2011-071105, filed on Mar. 28, 2011; the entire contents of which are incorporated herein by reference.

Embodiments described herein relate generally to a switching power supply device and a lighting apparatus using the same.

A switching power supply is a power supply device using a switching element to convert and regulate electrical power in a power conversion device for obtaining a desired output power from an input power. The switching power supply includes a power supply device including a DC-DC converter that converts a direct-current power to another direct-current power, and a direct-current power supply that includes a rectifying device to convert an alternating-current power to a direct-current power and supplies the direct-current input power to the DC-DC converter.

As a use of the switching power supply, it is already known that an LED lighting device having high circuit efficiency is obtained by turning on a light-emitting diode. For example, a DC-DC converter includes a step-down chopper, and an increasing current flows through an inductor via a switching element in an on state, so that a voltage is induced in a secondary winding magnetically coupled thereto. The voltage is fed back and the on operation of the switching element is continued. Besides, a resistance element for detecting the increasing current is inserted in series to the switching element, and a control circuit is added to turn off the switching element when the drop voltage of the resistance element exceeds a previously set threshold.

In the above structure, when the increasing current exceeds the threshold and the switching element is turned off, electromagnetic energy stored in the inductor is released, and a decreasing current flows via a diode and an output capacitor of an output end. When the decreasing current becomes 0, the switching element is turned on by counter electromotive force generated in the secondary winding of the inductor. The DC-DC conversion by self-excitation constant-current control is performed by repeating the circuit operation, and the light-emitting diode is turned on.

On the other hand, a semiconductor element using a wide band gap semiconductor or a compound semiconductor, such as III-V group semiconductor and IV semiconductor, having a large band gap, for example, silicon carbide (SiC), gallium nitride (GaN) or diamond attracts attention as a semiconductor element having the potential to greatly break through the performance limit of a Si power device. That is, in high speed switching, high temperature operation, large power operation and the like, the semiconductor element can operate in a region where a Si or GaAs element cannot operate. Thus, also in the field of a high frequency power device, the expectation for the semiconductor is high. Here, the wide band gap semiconductor is a semiconductor having a band gap wider than gallium arsenide (GaAs) having a band gap of about 1.4 eV. The wide band gap semiconductor is, for example, a semiconductor having a band gap of 1.5 eV or more, gallium phosphide (GaP, band gap: about 2.3 eV), gallium nitride (GaN, band gap: about 3.4 eV), diamond (C, band gap: about 5.27 eV), aluminum nitride (AlN, band gap: about 5.9 eV), silicon carbide (SiC) or the like.

A wide band gap semiconductor transistor is an element which can be used as a switching element, and more specifically, is a transistor fabricated using a wide band gap semiconductor. For example, the wide band gap semiconductor transistor is a field effect high-frequency transistor including a pair of main terminals (drain, source) and a control terminal (gate). The transistor is typically a high electron mobility transistor (HEMT). Since the semiconductor transistor has the excellent characteristics as described above, when the wide band gap semiconductor transistor is used as a switching element of a switching power supply, a high frequency operation of 10 MHz or higher can be performed. As a result, the switching power supply, especially an inductor can be greatly miniaturized.

Incidentally, as the wide band gap semiconductor transistor, a JFET (junction FET), a SIT (Static Induction Transistor), a MESFET (metal-semiconductor FET: Metal-Semiconductor-Field-Effect-Transistor), a HFET (Heterojunction Field Effect Transistor), a HEMT (High Electron Mobility Transistor) and a storage FET are enumerated.

Besides, the wide band gap semiconductor transistor often has a normally-on characteristic in which a drain current flows when a gate voltage is 0. Accordingly, in order to certainly turn off the semiconductor element having the normally-on characteristic (hereinafter referred to as a normally-on switch), a drive circuit for negative gate voltage is required. Incidentally, a wide band gap semiconductor transistor having a normally-off characteristic is also obtained. In this case, a drive circuit for positive voltage is required.

When constant-current control is performed, a related art switching power supply requires a current feedback type feedback circuit including an impedance unit, such as a resistance element, that is inserted in series to a switching element and detects an increasing current flowing through an inductor, and a control circuit that turns off the switching element when the voltage drop of the impedance unit reaches a previously set threshold. Thus, the circuit structure becomes complicated, and miniaturization is difficult.

When the wide band gap semiconductor transistor is used as a switching element of a switching power supply, since the switching element can be operated at a high frequency of 10 MHz or higher as stated above, the whole apparatus can be remarkably miniaturized. In this case, when a constant current element including a wide band gap semiconductor transistor is connected in series to the source of the wide band gap semiconductor transistor, an impedance unit such as a resistance element for current detection can be omitted, and a structure of a gate drive circuit is simplified, and therefore, further miniaturization can be realized.

Although the switching power supply using the wide band gap semiconductor transistor can be further miniaturized by using the two wide band gap semiconductor transistors for switching and for constant current control as stated above, there is a limit in meeting a request for further cost reduction and miniaturization.

An exemplary embodiment provides a switching power supply device in which cost reduction and miniaturization are realized by using one switching element for switching and for constant current control, and a lighting apparatus using the same.

FIG. 1 is a block diagram of a switching power supply device of a first embodiment.

FIG. 2 is a graph showing a current-voltage characteristic of a switching element.

FIG. 3A to FIG. 3F are current-voltage waveform views of respective parts.

FIG. 4 is a main part circuit view of a second embodiment.

FIG. 5 is a main part circuit view of a third embodiment.

FIG. 6 is a main part circuit view of a fourth embodiment.

FIG. 7A to FIG. 7C are current-voltage waveform views of respective parts.

FIG. 8 is a circuit view of a switching power supply device of a fifth embodiment.

FIG. 9 is a circuit view of a switching power supply device of a sixth embodiment.

In general, according to one embodiment, a switching power supply device includes a direct-current power supply and a switching operation conversion circuit. The switching operation conversion circuit is connected to the direct-current power supply and includes an inductor, a switching element, a diode, a gate drive circuit, an input end and an output end. The switching element allows an increasing current to flow to the inductor from the direct-current power supply at an on time of the switching element. The diode allows a decreasing current to flow to the inductor at an off time of the switching element. The gate drive circuit controls a gate voltage of the switching element to turns off the switching element when the increasing current reaches a saturated state relative to a source-drain voltage of the switching element. The input end is connected to the direct-current power supply. The output end is connected with a load.

According to another embodiment, a lighting apparatus includes a switching power supply device and an illumination load connected to an output end of the switching power supply device. The switching power supply device includes a direct-current power supply and a switching operation conversion circuit. The switching operation conversion circuit is connected to the direct-current power supply and includes an inductor, a switching element, a diode, a gate drive circuit, an input end and an output end. The switching element allows an increasing current to flow to the inductor from the direct-current power supply at an on time of the switching element. The diode allows a decreasing current to flow to the inductor at an off time of the switching element. The gate drive circuit controls a gate voltage of the switching element to turns off the switching element when the increasing current reaches a saturated state relative to a source-drain voltage of the switching element. The input end is connected to the direct-current power supply. The output end is connected with the illumination load.

First Embodiment

FIG. 1 is a block diagram of a switching power supply device of a first embodiment.

In the first embodiment, a switching power supply device SPS includes a direct-current power supply DC and a switching operation conversion circuit SC, and the output thereof is supplied as direct-current power to a load LC.

The direct-current power supply DC is a unit to input a direct-current power before conversion to the switching operation conversion circuit SC described later. Although any structure may be used as long as the direct-current power before conversion is outputted, for example, the direct-current power supply is mainly composed of a rectifying circuit, and a smoothing circuit including a smoothing capacitor or the like may be added if desired. In this embodiment, the rectifying circuit is preferably constituted by a bridge type rectifying circuit. Besides, input terminals T 1 and T 2 of the direct-current power supply DC are connected to an alternating-current power supply AC, for example, a commercial alternating-current power supply, and a direct-current voltage is obtained by full-wave rectifying the alternating-current voltage. However, if desired, the direct-current power supply DC may be constructed such that an active filter such as a step-up chopper is added to the output side of the rectifying circuit.

The switching operation conversion circuit SC can also be called a DC-DC converter, and includes a switching element Q 1 , an inductor L 1 , a diode D 1 , a gate drive circuit GD, input ends T 3 and T 4 , and output ends T 5 and T 6 . The switching operation conversion circuit may be mainly composed of a chopper and a flyback converter. Incidentally, the chopper is the concept including various choppers such as a step-down chopper, a step-up chopper and a step-up/down chopper.

A first circuit A is a current path including the switching element Q 1 and the inductor L 1 connected in series. The switching element Q 1 is driven by the after-mentioned gate drive circuit GD and performs a switching operation. The structure is such that at the on time, an increasing current flows from the direct-current power supply DC through the transistor Q 1 to the inductor L 1 . By this, electromagnetic energy is stored in the inductor L 1 .

The switching element Q 1 is, for example, a wide band gap semiconductor transistor, for example, a gallium nitride high electron mobility transistor (GaN-HEMT). In this embodiment, the switching element Q 1 performs the switching operation. The switching element Q 1 is used, so that the switching characteristic in high frequency of 1 MHz or higher, for example, 10 MHz or higher is remarkably improved as described before, and switching loss is reduced, and further, the inductor L 1 is remarkably miniaturized. Accordingly, the switching power supply device can be greatly miniaturized.

Besides, the switching element Q 1 may have a normally-on characteristic or a normally-off characteristic. In general, the switching element having the normally-on characteristic is more easily obtained and the cost is low. However, the switching element having the normally-off characteristic can also be obtained and may be used. Besides, since the threshold of switching of the normally-on switch is negative, if the normally-on switch is used, off control using a drive winding L 2 magnetically coupled to the inductor L 1 , that is, a secondary winding becomes easy, and the normally-on switch is preferable.

Further, if the switching element Q 1 is, for example, the wide band gap semiconductor transistor, as shown in FIG. 2 , the current-voltage characteristic thereof has a saturation region R S continuous from a non-saturation region R L through pinch-off, and it is known that an excellent constant current characteristic is exhibited in the saturation region R S . Incidentally, in FIG. 2 , the horizontal axis indicates drain current I D , and the vertical axis indicates drain-source voltage V DS . The drawing shows plural characteristic curves changing according to gate voltage V G , that is, gate-source voltage as a parameter. In the present specification, the non-saturation region R L is a region in which the current-voltage characteristic becomes substantially linear. The saturation region R S is a region in which the current-voltage characteristic deviates from the straight line in the non-saturation region R L , and an increase in drain-source voltage with respect to an increase in drain current is larger than the increase indicated by the inclination of the straight line in the non-saturation region R L . That is, the saturation region R S is the region including a region in which the switching element Q 1 exhibits the constant current characteristic and a portion of a transition region between the non-saturation region R L and a region exhibiting the constant current characteristic.

As is understood from FIG. 2 , if the switching element Q 1 has the same chip area as a Si transistor, the increase in voltage with respect to the increase in current is very small in the non-saturation region R L . This indicates that in this region R L , the switching element Q 1 is in an on state where the on resistance is very low. In the saturation region R S , even if the voltage is raised, the current remains constant and is saturated. Besides, a saturation current value as a current value at saturation time changes according to the magnitude of the gate voltage. Accordingly, the saturation current value can be selected to a desired value by controlling the gate voltage V GS . Incidentally, in order to facilitate understanding of the above, FIG. 2 shows the current-voltage characteristic curves at plural gate voltages V G1 to V G4 different in voltage value.

In this embodiment, when the increasing current reaches a threshold, the switching element Q 1 is turned off by the control of the gate drive circuit GD. At that time, the gate voltage in which the threshold is selected to be located in the saturation region of the current-voltage characteristic is applied to the transistor Q 1 . As a result, when the increasing current flowing during the on period of the switching element Q 1 reaches the threshold (first threshold), the current flowing through the switching element Q 1 is abruptly saturated. Then, when the saturation current is abruptly changed from the increasing state to the constant threshold, the after-mentioned gate drive circuit GD applies the gate voltage selected to turn off the switching element Q 1 and turns off the switching element Q 1 .

A second circuit B is a current path including the inductor L 1 and the diode D 1 connected in series. When the switching element Q 1 is turned off by the switching operation, the inductor L 1 discharges the electromagnetic energy stored therein. Thus, a decreasing current flows in the second circuit B through the diode D 1 . When the decreasing current reduces a threshold (second threshold), the after-mentioned drive circuit GD again turns on the switching element Q 1 . Incidentally, a synchronous rectifier using a field effect transistor FET or the like as a diode may be used.

The gate drive circuit GD is a circuit unit to control at least turning-off of the switching element Q 1 when the on current reaches the first threshold. Besides, also when the switching element is turned on, the gate drive circuit GD can be made to function if desired. In this case, the switching element Q 1 is normally-on, and also when the gate drive circuit GD removes application of the gate voltage in order to turn off the switching element, the gate drive circuit GD can be said to function.

The off operation by the gate drive circuit GD in switching of the switching element Q 1 may be self-excited control or separately-excited control.

In the case of the self excitation, the structure can be made such that electric change caused by the circuit operation of the first and the second circuit A and B, such as the change of the drain-source voltage V DS of the switching element Q 1 or the change of the terminal voltage V L1 of the inductor L 1 , is detected and is fed back to turn off. By this, both the off timing and the gate voltage by which the current-voltage characteristic is saturated at the first threshold of current, in other words, the switching element Q 1 exhibits the constant current characteristic can be obtained.

In the case of the separate excitation, an oscillator for switching is provided in the switching operation conversion circuit SC, and the timing of on and off of the switching element Q 1 is set. The application of a required gate voltage can be controlled so that when the increasing current reaches the first threshold in the on period, the switching element is positioned in the saturation region and is turned off, and is again turned on at the timing when the decreasing current reaches the second threshold.

The input ends T 3 and T 4 are a unit constituting the input part of the switching operation conversion circuit SC, and are connected to the output ends of the direct-current power supply DC. The input ends T 3 and T 4 may have a form of a connection unit such as a terminal, or may be simply a conductive wire connecting the direct-current power supply DC and the input part of the switching operation conversion circuit SC.

The output ends T 5 and T 6 are a unit constituting the output part of the switching operation conversion circuit SC, and the load LC is connected thereto. The output ends T 5 and T 6 may adopt a free form similarly to the input ends T 3 and T 4 .

The load LC is driven and operated by the output of the switching power supply device SPS. In this embodiment, the load LC may have any structure, and in the illustrated example, the load is made of an LED LD. Accordingly, the lighting apparatus can be constructed which includes a light source such as a lamp, that is, an illumination load, and the switching power supply device SPS of the embodiment as a part of or substantially the whole of a lighting circuit to directly or indirectly drive the illumination load.

In the switching operation conversion circuit SC, as another structure, an output capacitor can be connected between the output ends T 5 and T 6 . The output capacitor causes a high frequency component, which is generated by switching of the switching element Q 1 in the switching operation conversion circuit SC, to bypath, and can suppress the high frequency component from flowing out to the load LC side.

In the first embodiment, in the switching operation conversion circuit SC, the inductor L 1 repeats the storage and release of the electromagnetic energy by the switching of the switching element Q 1 , and as a result, the direct-current power inputted from the input ends T 3 and T 4 is converted to a desired power and is supplied from the output ends T 5 and T 6 to the load LC. Besides, when the increasing current reaches the first threshold, since the switching element Q 1 transitions to the saturation region and the constant current flows, the switching element Q 1 is turned off by indirectly or directly using the change of the circuit operation at that time. Accordingly, a constant current unit such as a constant current diode is not required to be connected in series to the switching element Q 1 , and the circuit structure can be simplified.

Next, the circuit operation in the first embodiment will be described in brief with reference to current-voltage waveform views of respective parts shown in FIGS. 3A to 3F . FIG. 3A shows a gate voltage waveform of the switching element of the first circuit A, FIG. 3B shows a waveform of the drain-source voltage V DS of the switching element Q 1 , FIG. 3C shows a waveform of an increasing current I U , FIG. 3D shows a waveform of a decreasing current I D , FIG. 3E shows an addition waveform of the increasing current and the decreasing current, and FIG. 3F shows a waveform of the voltage V L1 of the inductor L 1 .

The gate voltage V GS of the normally-on switching element Q 1 is, for example, −8 V in a low negative voltage period and −5 V in a high negative voltage period. Thus, the switching element Q 1 is brought into an off state in the low negative voltage period. Besides, although the switching element is brought into the on state in the high negative voltage period, when the increasing current I U reaches about 220 mA, the switching element exhibits the saturation characteristic. That is, the first threshold can be selected to 220 mA. Accordingly, when the gate voltage V GS of FIG. 3A is made −5 V, the switching element Q 1 is turned on, and the linearly increasing current I U flows through the inductor L 1 . When the increasing current reaches 220 mA of the first threshold, the switching element Q 1 is saturated, and the drain-source voltage V DS increases as shown in FIG. 3B similarly to the case where the transistor is brought into the off state. In synchronization with that time, the gate voltage V GS outputted by the gate drive circuit GD is changed to −8 V as shown in FIG. 3A , so that the switching element Q 1 is turned off. When the off period occurs, as shown in FIG. 3D , the decreasing current flows from the inductor L 1 . When the gate voltage V GS is changed to −5 V as shown in FIG. 3A when the decreasing current reaches the second threshold, for example, 0 mA as shown in FIG. 3D , the switching element Q 1 is again turned on. After that, the above operation is repeated. As a result, as shown in FIG. 3E , the increasing current I U and the decreasing current I D continuously flow through the inductor L 1 . During the period, the polarity of the voltage at both ends of the inductor L 1 is alternately inverted as shown in FIG. 3F .

Second Embodiment

A second embodiment will be described with reference to FIG. 4 . This embodiment is an example of a gate drive circuit GD of a normally-off switching element Q 1 . In this embodiment, the gate drive circuit GD is mainly composed of a switch element Q 2 , a voltage dividing circuit VD 1 and a buffer amplifier BA.

That is, the switch element Q 2 is made of, for example, a bipolar transistor, and is turned on and off so that the output voltage of the voltage dividing circuit VD 1 is changed to a high voltage or a low voltage. The base of the switch element Q 2 is connected to a switching period signal source (not shown). Incidentally, the switching period signal source is a signal source to determine the timing of on and off of the switching element Q 1 . The voltage dividing circuit VD 1 is composed of a series circuit of resistors R 1 and R 2 . In the drawing, the upper end is connected to a plus control power supply Vcc, and the lower end is connected to the source of the switching element Q 1 . The collector and emitter of the switch element Q 2 are connected in parallel to the resistor R 2 . The buffer amplifier BA intervenes between the output end of the voltage dividing circuit VD 1 and the gate of the switching element Q 1 , and the output is applied to the gate.

The circuit operation of this embodiment will be described. When the switching period signal source connected to the base of the switch element Q 2 is periodically turned on and off, the gate drive circuit GD switches on and off the switching element Q 1 in reverse phase relation to the switching period signal source. That is, when the switch element Q 2 is turned off, the short circuit of the resistor R 2 of the voltage dividing circuit VD 1 is released and the terminal voltage of the resistor R 2 becomes high. Thus, the gate voltage V GS applied between the gate and source of the switching element Q 1 becomes high and the switching element is turned on. Next, when the switch element Q 2 is turned on, the resistor R 2 is shorted, and the output of the voltage dividing circuit VD 1 becomes 0. Thus, the switching element Q 1 is turned off.

Accordingly, the switching element Q 1 can be switched according to the switching of the switch element Q 2 . However, as is understood from the above, with respect to the gate voltage when the switching element Q 1 is turned on, the value at the timing when the switching element Q 1 is turned off and at which the increasing current transitions into the saturation region Rs is selected to the first threshold. Incidentally, with respect to the switching period signal source to control the timing of the switching of the switch element Q 2 , as stated above, the switching operation conversion circuit SC may be self-excited and operated by using the feedback signal obtained from the switching operation conversion circuit SC, or may be separately excited and operated by using a separately provided oscillator.

Third Embodiment

A third embodiment will be described with reference to FIG. 5 . Incidentally, in the drawing, the same portion as FIG. 4 is denoted by the same reference numeral and the description thereof is omitted. This embodiment is an example in which a gate drive circuit GD includes a holding circuit HC. The holding circuit HC holds a switching element Q 1 in an off state during a current flowing period of a decreasing current flowing through a switching operation conversion circuit SC, and releases the holding of the off state when the switching element Q 1 is turned on.

In this embodiment, in comparison with the second embodiment, the holding circuit HC is mainly composed of a voltage dividing circuit VD 2 and a switch element Q 3 , and constitutes a part of the gate drive circuit GD.

That is, the voltage dividing circuit VD 2 of the holding circuit HC includes a series circuit of resistors R 3 and R 4 , and in the drawing, the upper end is connected to the drain of the switching element Q 1 , and the lower end is connected to the source. Besides, the switch element Q 3 is made of, for example, a bipolar transistor, the collector and the emitter are connected in parallel to the resistor R 4 of the voltage dividing circuit VD 2 , and an on signal source (not shown) of the switching element Q 1 is connected to the base. Incidentally, a resistor R 5 is connected in series between the base of the switch element Q 3 and the on signal source, and a resistor R 6 is connected between the base and the emitter to regulate the base current. Incidentally, the on signal source is a signal source to determine the timing of on of the switching element Q 1 .

The circuit operation of this embodiment will be described. In the off state of the switching element Q 1 , since the drain-source voltage V DS is high, the voltage is applied to the base of the switch element Q 2 through the voltage dividing circuit VD 2 of the holding circuit HC, so that the base current flows through the switch element Q 2 , and the switch element Q 2 is on. Since the gate voltage of the switching element Q 1 becomes low by this, the switching element Q 1 holds the off state.

In order to turn on the switching element Q 1 , the switch element Q 2 is periodically shorted between the base and emitter by the switch element Q 3 . Then, the output of a voltage dividing circuit VD 1 becomes high, the holding of the off state of the switching element Q 1 by the holding circuit HC is released, and the switching element Q 1 is turned on. In order to turn on the switch element Q 3 , the base potential is made high by the on signal source.

Accordingly, when the switching element Q 1 is periodically turned on by the on signal source for a specified period, the switching operation conversion circuit SC can be operated. Of course, the operation condition is naturally set so that the increasing current reaches the first threshold at the end of the on time, and the drain current I D transitions into the saturation region.

Fourth Embodiment

A fourth embodiment will be described with reference to FIG. 6 . Incidentally, in the drawing, the same portion as FIG. 4 is denoted by the same reference numeral, and the description thereof is omitted. In this embodiment, a gate drive circuit GD includes a protecting circuit PC which operates at the initial on stage of the switching element Q 1 . The protecting circuit PC is a unit to perform protection by changing a gate voltage so that at the initial on stage of the switching element Q 1 , the on operation is not distributed by a transient pulse current which is easily generated by a snubber capacitor or the like equivalently connected to both ends of the switching element. In this embodiment, the protecting circuit PC includes a timer TM and a switch element Q 4 .

The timer TM includes a switch element Q 5 and an integration circuit INT. The switch element Q 5 , for example, a bipolar transistor is connected between a resistor R 2 of a voltage dividing circuit VD 1 and the source of the switching element Q 1 . The integration circuit INT includes a series circuit of resistors R 7 to R 10 connected as shown in FIG. 6 and a capacitor C 1 . The series circuit of the resistors R 7 to R 10 is connected between a control power supply Vcc and the source of the switching element Q 1 , and the capacitor C 1 is connected in parallel to a series portion of the resistors R 9 and R 10 .

The switch element Q 4 is connected between a connection point between the resistor R 7 and the resistor R 8 of the integration circuit INT and the source of the switching element Q 1 , and resets the timer by shorting the integration circuit INT except for the resistor R 7 . Besides, the base of the switch element Q 4 is connected to the base of a switch element Q 2 . Accordingly, the switch elements Q 2 and Q 5 synchronously perform on and off operations according to a switching period signal from a switching period signal source.

The circuit operation of this embodiment will be described. At turning-on of the switching element Q 1 , when an off signal reaches from the switching period signal source, both the switch elements Q 2 and Q 4 are turned on. At this time, since the switch element Q 2 shorts the resistor R 2 of the voltage dividing circuit VD 1 and the switch element Q 5 , the gate potential of the switching element Q 1 becomes low, for example, 0 V and the switching element is turned on. Besides, at the same time, the switch element Q 4 resets the timer TM.

Next, when an on signal reaches from the switching period signal source, both the switch elements Q 2 and Q 4 are turned off. The short circuit of the resistor R 2 of the voltage dividing circuit VD 1 and the switch element Q 5 is released by turning-off of the switch element Q 2 , and the timer TM starts to operate by turning-off of the switch element Q 4 . However, since the integration operation of the integration circuit INT just starts, the switch element Q 5 is off. Thus, since the gate potential of the switching element Q 1 is increased to the voltage of the control power supply Vcc, the switching element Q 1 is sufficiently conductive, and the non-saturation region R L is significantly expanded. As stated above, even if a transient pulse current, which is easily generated in the switching element Q 1 at the initial on stage, flows, the switching element Q 1 is not saturated. That is, the saturation current value becomes high at the initial on stage of the switching element Q 1 . Thus, since the on operation of the switching element Q 1 is protected, an erroneous operation is hard to occur in the on operation at the initial on stage.

When the initial stage of the on period passes, since the base current flows and the switch element Q 5 is turned on by the integration operation of the integration circuit INT, the gate potential of the switching element Q 1 decreases to a middle level. In this state, although the transistor Q 1 remains on, since the saturation current value decreases to the first threshold, when the on period reaches the end, the drain current I D reaches the first threshold. When the switch element Q 2 is turned on by the off signal from the switching period signal source, the switching element Q 1 is turned off. Since the switch element Q 4 is tuned on at the same time as the switch element Q 2 , the integration circuit INT is reset, and the timer TM is reset for next turning-on of the switching element Q 1 .

FIGS. 7A to 7C are waveform views of respective parts for explaining the circuit operation of the protecting circuit PC in the fourth embodiment. FIG. 7A shows a drain current waveform of the switching element Q 1 , FIG. 7B shows a gate voltage waveform, and FIG. 7C shows a gate voltage waveform in another modified example.

As shown in FIGS. 7A to 7C , at the initial stage of the switching element Q 1 , a pulse current I P can occur in the drain current I D . For example, when the switching element Q 1 is changed from off to on when the current of the inductor L 1 has a finite value before the current becomes zero, the pulse current I P occurs at the initial on stage. However, since the gate voltage shown in FIG. 7B becomes sufficiently high at the initial on stage, the non-saturation region of the current-voltage characteristic of the switching element Q 1 is expanded. Thus, even if the pulse current I P occurs, the current can be sufficiently made to flow in the non-saturation region. Thus, the on operation is protected and the turning-on can be certainly performed.

FIG. 7C shows the modified example of the gate voltage at the on time. In this example, although the gate voltage is maximum at the initial stage, thereafter, the gate voltage is continuously decreased, and is set so that the saturation current value of the current-voltage characteristic immediately before turning-off becomes the first threshold. Also in this example, the circuit operation similarly to the above can be performed.

Fifth Embodiment

A fifth embodiment will be described with reference to FIG. 8 . Incidentally, the same portion as FIG. 1 is denoted by the same reference numeral and the description thereof is omitted. In this embodiment, a switching operation conversion circuit SC of a switching power supply device SPS is composed of a step-down chopper.

That is, input ends T 3 and T 4 of the switching operation conversion circuit SC are connected to a direct-current power supply DC. A first circuit A includes a series circuit of a switching element Q 1 , an inductor L 1 and an output capacitor C 2 , and both ends thereof are connected to the input ends T 3 and T 4 . In s second circuit B, the inductor L 1 , the output capacitor C 2 and a diode D 1 are connected in series and form a closed circuit. Output ends T 5 and T 6 are connected to both ends of the output capacitor C 2 . Besides, the output ends T 5 and T 6 are connected with a load LC. Incidentally, the switching element Q 1 is a normally-off element and is, for example, a GaN-HEMT.

A gate drive circuit GD is mainly composed of a voltage dividing circuit VD 1 , a buffer amplifier BA, a switch element Q 2 , a drive winding L 2 , diodes D 2 and D 3 , a capacitor C 3 and a Zener diode ZD 1 . The voltage dividing circuit VD 1 is connected in parallel to the capacitor C 3 and the Zener diode ZD 1 . The buffer amplifier BA is connected between a connection point between resistors R 1 and R 2 of the voltage dividing circuit VD 1 and the gate of the switching element Q 1 as shown in the drawing. The switch element Q 2 is similar to the second embodiment in that the switch element is connected in parallel to the resistor R 2 and can short the resistor R 2 . The drive winding L 2 is magnetically coupled to the inductor L 1 with the illustrated polarity, and is connected so that the induced voltage is applied to a series circuit of resistors R 11 and R 12 through a coupling capacitor C 4 , and is applied to the parallel circuit of the voltage dividing circuit VD 1 , the capacitor C 3 and the Zener diode ZD 1 through the diode D 2 . A connection point between the resistors R 11 and R 12 is connected to the base of the switch element Q 2 . Besides, the diode D 3 is connected in parallel to the series circuit of the resistors R 11 and R 12 in the illustrated polarity, and bypasses the voltage induced in the drive winding L 2 in the opposite polarity with respect to the diode D 3 .

Next, the circuit operation of the fifth embodiment will be described. When the direct-current power supply DC is applied between the input ends T 3 and T 4 , first, a positive potential is supplied to the gate of the switching element Q 1 through a not-shown starting circuit. The switching element Q 1 is turned on with a sufficient conductivity, and an increasing current flows through the inductor L 1 in the first circuit A.

In the first circuit A, when the increasing current flows through the inductor L 1 , a voltage with a polarity in the forward direction with respect to the diode D 3 is induced in the drive winding L 2 magnetically coupled thereto. Since the switch element Q 2 is turned off by this, both a voltage applied to a non-inversion input terminal of the buffer amplifier BA and an output voltage become high plus voltages. As a result, the gate potential becomes higher than the source potential. As a result, a saturation current value of the current-voltage characteristic of the switching element Q 1 becomes low.

Accordingly, when the increasing current reaches the saturation current value, the switching element Q 1 is abruptly saturated and is brought into the constant current state, and the induced voltage of the drive winding L 2 becomes 0. Then, the switch element Q 2 is biased by the resistors R 11 and R 12 and is turned on. As a result, the voltage at both ends of the resistor R 2 becomes zero, and the output voltage of the buffer amplifier BA becomes zero. The gate-source voltage of the switching element Q 1 becomes substantially zero, and the switching element Q 1 is turned off.

When the switching element Q 1 is turned off, electromagnetic energy stored in the inductor L 1 is released, and the decreasing current flows in the second circuit B. The switching element Q 1 is held in the off state during a specified period in which the decreasing current flows and the switch element Q 2 keeps the on state. When the decreasing current becomes 0, the switch element Q 2 is turned off, and the gate voltage of the switching element Q 1 becomes high. As a result, the switch element Q 2 is again turned on, and thereafter, the foregoing circuit operation is repeated.

In the above circuit operation, since the increasing current and the decreasing current continuously flow through the output capacitor C 2 and the capacitor is charged, the DC-DC converted direct-current power is outputted between the output ends T 5 and T 6 , and the load LC connected thereto is driven and operated.

Sixth Embodiment

A sixth embodiment will be described with reference to FIG. 9 . Incidentally, the same portion as FIG. 1 is denoted by the same reference numeral and the description thereof is omitted. In this embodiment, a switching operation conversion circuit SC of a switching power supply device SPS is composed of a step-down chopper.

That is, input ends T 3 and T 4 of the switching operation conversion circuit SC is connected to a direct-current power supply DC. A first circuit A includes a series circuit of a switching element Q 1 , an inductor L 1 and an output capacitor C 2 , and both ends thereof are connected to input ends T 3 and T 4 . In a second circuit B, the inductor L 1 , the output capacitor C 2 and a diode D 1 are connected in series and a closed circuit is formed. Output ends T 5 and T 6 are connected to both ends of the output capacitor C 2 . Besides, a load LC is connected to the output ends T 5 and T 6 . Incidentally, the switching element Q 1 is composed of a normally-on GaN-HEMT.

A gate drive circuit GD is mainly composed of a voltage dividing circuit VD 1 , a buffer amplifier BA, a switch element Q 2 , a drive winding L 2 , diodes D 2 and D 3 , a capacitor C 3 and a Zener diode ZD 1 . The voltage dividing circuit VD 1 is connected in parallel to the capacitor C 3 and the Zener diode ZD 1 . The buffer amplifier BA is connected between a connection point between resistors R 1 and R 2 of the voltage dividing circuit VD 1 and the source of the switching element Q 1 as shown in the drawing. The switch element Q 2 is similar to the second embodiment in that the switch element is connected in parallel to the resistor R 2 , and can short the resistor R 2 . The drive winding L 2 is magnetically coupled to the inductor L 1 in the illustrated polarity, and is connected so that the induced voltage is applied to the series circuit of resistors R 11 and R 12 through a coupling capacitor C 4 , and is applied to a parallel circuit of the voltage dividing circuit VD 1 , the capacitor C 3 and the Zener diode ZD 1 through the diode D 2 . A connection point between the resistors R 11 and R 12 of the series circuit is connected to the base of the switch element Q 2 . The diode D 3 is connected in parallel to the series circuit of the resistors R 11 and R 12 in the illustrated polarity, and bypasses the voltage induced in the drive winding L 2 in the opposite polarity with respect to the diode D 3 .

As stated above, this embodiment is different from the fifth embodiment in the structure of the switching element Q 1 , the buffer amplifier BA and the drive winding L 2 .

Next, the circuit operation of the sixth embodiment will be described. When the direct-current power supply DC is applied between the input ends T 3 and T 4 , since a potential is not applied to the gate of the switching element Q 1 at first, the switching element is turned on with a sufficient conductivity, and an increasing current flows through the inductor L 1 in the first circuit A. Incidentally, at the initial on stage, since the gate potential of the switching element Q 1 is 0, the switching element is turned on with the sufficient conductivity, and accordingly, even if a pulse voltage is generated in the switching element Q 1 , the on operation is not disturbed.

In the first circuit A, when the increasing current flows through the inductor L 1 , a voltage with the opposite polarity with respect to the diode D 3 is induced in the drive winding L 2 magnetically coupled thereto. Since the switch element Q 2 is turned on by this, both a voltage applied to the non-inversion input terminal of the buffer amplifier BA and an output voltage become low plus voltages. As a result, the gate potential becomes a relatively high negative potential, for example, −5 V. As a result, a saturation current value of current-voltage characteristic of the switching element Q 1 becomes small, for example, 220 mA.

Accordingly, since the saturation current value becomes the first threshold at the time of turning off the switching element Q 1 , when the increasing current reaches 220 mA, the switching element Q 1 is abruptly saturated and is brought into the constant current state, and the induced voltage of the drive winding L 2 becomes 0. Then, the switch element Q 2 is turned off since the forward bias thereof disappears. On the other hand, since the parallel circuit of the voltage dividing circuit VD 1 , the capacitor C 3 and the Zener diode ZD 1 is connected to the drive winding L 2 through the diode D 2 . Thus, since the output voltage of the buffer amplifier BA becomes high, the source of the switching element Q 1 becomes a high plus potential, while the gate potential becomes a relatively low negative potential, for example, −8 V, and the switching element Q 1 is turned off.

When the switching element Q 1 is turned off, since the capacitor C 3 has a remaining charge, the output of the voltage dividing circuit VD 1 continues, and accordingly, the switching element Q 1 is held in the off state for a specified period. When the switching element Q 1 is turned off, electromagnetic energy stored in the inductor L 1 is released, and the decreasing current flows in the second circuit B. When the decreasing current becomes 0, since the gate voltage of the switching element Q 1 disappears, the switching element is again turned on, and after that, the foregoing circuit operation is repeated.

In the above circuit operation, since the increasing current and the decreasing current continuously flow through the output capacitor C 2 , the DC-DC converted direct-current power is outputted between the output ends T 5 and T 6 , and the load LC connected thereto is driven and operated.

Although the exemplary embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the invention. Indeed, these novel embodiments can be carried out in a variety of other forms, and various omissions, substitutions and changes can be made within the scope not departing from the gist of the invention. These embodiments and modifications thereof fall within the scope and the gist of the invention and fall within the scope of the invention recited in the claims and their equivalents.