LED drive circuit and LED illumination component using the same

This application is based on Japanese Patent Application No. 2010-187697 filed on Aug. 24, 2010, the contents of which are hereby incorporated by reference.

1. Field of the Invention

The present invention relates to an LED drive circuit and an LED illumination component using the same.

2. Description of the Prior Art

An LED (light-emitting diode) is characterized by its low current consumption, long life, and so on, and its range of applications has been expanding not only to display devices but also to illumination apparatuses and the like. An LED illumination apparatus often uses a plurality of LEDs in order to attain desired illuminance (see, for example, JP-A-2004-327152, JP-A-2006-319172, and JP-A-2008-104273).

A general-use illumination apparatus often uses a commercial alternating current power source, and considering, for example, a case where an LED illumination component is used in place of a general-use illumination component such as an incandescent lamp, it is desirable that, similarly to a general-use illumination component, an LED illumination component also be configured to use a commercial alternating current power source.

Furthermore, in seeking to perform light control of an incandescent lamp, a phase-control light controller (referred to generally as an incandescent light controller) is used in which a switching element (generally, a thyristor element or a triac element) is switched on at a certain phase angle of an alternating current power source voltage and that thus allows light control through control of power supply to the incandescent lamp to be performed easily with a simple operation of a volume element (see, for example, JP-A-2005-26142). It is known, however, that in performing light control of a low-wattage incandescent lamp by use of a phase-control light controller, connecting the incandescent lamp to the light controller leads to the occurrence of flickering or blinking, so that the light control cannot be performed properly.

It is desirable that in seeking to perform light control of an LED illumination component that uses an alternating current power source, the LED illumination component be connectable as it is to an existing phase-control light controller for an incandescent lamp. By changing only an illumination component to an LED illumination component while using existing light control equipment therewith, compared with a case of using an incandescent lamp, power consumption can be reduced considerably. Furthermore, this can also secure compatibility without requiring the light control equipment to be changed to a type used exclusively for an LED illumination component and thus reduces equipment cost.

Now, FIG. 16 shows a conventional example of an LED illumination system capable of performing light control of an LED illumination component that uses an alternating current power source. An LED illumination system shown in FIG. 16 includes a phase-control light controller 200 , an LED drive circuit 300 having a diode bridge DB 1 and a current limitation portion IL, and an LED array 400 formed by connecting LEDs in series. The phase-control light controller 200 is connected in series between a commercial power source 100 that is an alternating current power source and the current limitation portion IL. In the phase-control light controller 200 , a resistance value of a variable resistor Rvar is made to vary, and a triac Tri is thus switched on at a power source phase angle depending on the resistance value. Typically, the variable resistor Rvar is built in the form of a rotary knob or a slider and so configured that changing an angle of rotation of the knob or the position of the slider allows light control of the illumination component. Moreover, in the phase-control light controller 200 , a capacitor Ca and an inductor L constitute a noise suppression circuit that reduces noise fed back into an alternating current power source line from the phase-control light controller 200 . FIG. 17 shows output waveforms of the light controller and those of the diode bridge DB 1 , which correspond to phase angles of 0°, 45°, 90°, and 135° of the phase-control light controller 200 , respectively. As the phase angle increases, an average value of a voltage of the output waveform of the diode bridge decreases. In a case where the LED illumination component is connected to the phase-control light controller 200 , as the phase angle of the light controller increases, resulting brightness decreases.

When the phase angle of the phase-control light controller 200 is increased to decrease resulting brightness of the LEDs, if an output voltage of the diode bridge DB 1 becomes smaller than a forward voltage (VF) obtained when the LED array 400 starts to glow, the LED array 400 no longer glows, and there occurs an abrupt decrease in current flowing through the light controller. Due to this abrupt decrease, the current flowing through the light controller falls below a level of an on-state holding current of the triac Tri in the light controller, so that the triac is switched off to halt the output of the light controller and thus to bring about an unstable state, which results in the occurrence of brightness flickering of the LED array 400 . Furthermore, when the triac Tri is switched from an off-state to an on-state through phase control of the output of the light controller, the LEDs are switched from an off-state to an on-state, so that there occurs an abrupt variation in impedance of the LEDs. This might cause ringing to occur at an edge of an output voltage of the light controller, where the output voltage varies abruptly, so that the triac Tri is brought to an unstable state to be switched off, which results in the occurrence of brightness flickering. For the above-described reason, in an LED illumination system adapted to the use of a phase-control light controller, when LEDs are not glowing, a current drawing circuit that forcibly passes a holding current is used to prevent a triac from being switched off. In this case, however, a drawn current is all converted to heat, which leads to a deterioration in efficiency of the LED illumination system and also requires heat radiation measures to be taken.

In a case of using a conventional incandescent lamp load, since a filament of tungsten or the like constitutes the load, even if a triac of a phase-control light controller is switched from an off-state to an on-state, there hardly occurs a variation in impedance, and thus a low impedance state is maintained. Thus, there occurs no abrupt variation in current flowing through the phase-control light controller, so that a stable light control operation can be performed as long as an alternating current power source has a voltage value of around 0 V or higher.

Furthermore, in a case of the conventional example shown in FIG. 16 , when an output voltage of the diode bridge DB 1 is lower than a forward voltage (VF) obtained when the LED array 400 starts to glow, the LEDs are switched off, and assuming that the alternating current power source is at a frequency of 60 Hz, since full-wave rectification is performed by the diode bridge, the LEDs are switched on/off repeatedly at a frequency of 120 Hz that is double the alternating current power source frequency. This switching on/off of the LEDs causes flickering and might disadvantageously make it more likely that such flickering is perceived by a user when the user quickly moves his/her line of sight in an attempt to follow a quick move in a sporting event or the like. In a case of using an incandescent lamp, due to a filament that has a response speed on the order of 0.1 seconds and thus does not respond to an on/off operation at 120 Hz, flickering as described above does not occur to a noticeable degree. On the other hand, in a case of using an LED, since its response speed is a million or more times higher than that of a filament used in an incandescent lamp, flickering tends to occur to a noticeable degree.

Moreover, FIG. 18 shows a relationship (light control curve) between a phase angle θ of the phase-control light controller and illumination brightness in each of a case of the conventional LED illumination system shown in FIG. 16 and a case of an incandescent lamp illumination system. In the conventional LED illumination system, there occurs no variation in brightness at the phase angle θ=0° to 45°, while at θ=45° or larger, a light amount varies linearly, and at θ=130°, the LED illumination system is turned off. The incandescent lamp exhibits a curve characterized in that a light amount decreases mildly starting at θ=0°, which at θ=50° to 100°, varies along with the light control curve of the conventional LED illumination system and at θ=120° to 150°, decreases mildly. Brightness is perceived logarithmically by human eyes, and thus a characteristic that a light amount decreases mildly with respect to the phase angle θ is the key to fine control of a light amount at low illuminance. The conventional LED illumination system has been disadvantageous in that since it dims abruptly at around θ=130°, a light amount at a phase angle of around 120° to 150° cannot be controlled finely compared with a case of an incandescent lamp.

It is an object of the present invention to provide an LED drive circuit and an LED illumination component that finely control a switching current and an LED current so as to maintain a holding current of a phase-control light controller, and thus a phenomenon can be suppressed in which the light controller is brought to an unstable state to cause brightness flickering, and increased efficiency can also be achieved. Furthermore, it is another object of the present invention to enable light control that allows the occurrence of flickering to be suppressed and a light control curve approximating that of an incandescent lamp to be obtained, thus causing no feeling of discomfort.

The present invention is to provide an LED drive circuit that can be connected to an alternating current power source via a phase-control light controller and drives an LED load. The LED drive circuit includes: a switching power source portion having a switching element, a switching current detection portion that detects a switching current, and an LED current detection portion that detects an LED current; a first subtraction portion that subtracts a given value from an output of the switching current detection portion; a first determination portion that determines, through computation on a plurality of externally inputted voltages, a target value to which the switching current is limited; a second subtraction portion that subtracts a given value from an output of the LED current detection portion; a second determination portion that determines, through computation on a plurality of externally inputted voltages, a target value of the LED current; a switching current limitation portion that, if a result of the subtraction by the first subtraction portion exceeds a result of the determination by the first determination portion, switches off the switching element so as to limit the switching current; and an LED current control portion that drives the switching element to control the LED current so that a result of the subtraction by the second subtraction portion coincides with a result of the determination by the second determination portion (first configuration).

According to this configuration, a switching current and an LED current are controlled finely so that a holding current is maintained. This can suppress a phenomenon in which a current holding unit of the phase-control light controller is switched off to bring about an unstable state, which results in the occurrence of brightness flickering of the LED load, and can also achieve increased efficiency.

Furthermore, in the above-described first configuration, the switching power source portion may be formed of a booster circuit (second configuration). This can stabilize a forward voltage to be applied to the LED load and thus can suppress the occurrence of flickering.

Furthermore, in the above-described first configuration, the following may be adopted. That is, the first subtraction portion is a first error amplifier that obtains a difference between an output voltage of the switching current detection portion and a reference voltage, and the first determination portion is a second error amplifier that obtains a difference between a switching current limit set voltage and a first offset voltage. The second subtraction portion is a third error amplifier that obtains a difference between an output voltage of the LED current detection portion and the reference voltage, and the second determination portion is a fourth error amplifier that obtains a difference between an LED current set voltage and a second offset voltage. The switching current limitation portion has a comparator that compares an output of the first error amplifier with an output of the second error amplifier, and switches off the switching element based on an output of the comparator. The LED current control portion has a fifth error amplifier that obtains a difference between an output of the third error amplifier and an output of the fourth error amplifier and a PWM signal generation portion that generates a PWM signal in accordance with an output of the fifth error amplifier, and drives the switching element based on the PWM signal (third configuration).

Furthermore, in the above-described third configuration, the following may be adopted. That is, there are further provided: a circuit current detection portion that detects currents in the entire circuit; and a current drawing portion that draws a current. An output of the circuit current detection portion is used as the reference voltage, and based on the output of the circuit current detection portion, the current drawing portion draws the current so that the currents in the entire circuit are made constant (fourth configuration).

According to this configuration, control is performed so that a total of a switching current, an LED current, and a drawn current is made constant, and thus a holding current of the phase-control light controller can be maintained.

Furthermore, in the above-described fourth configuration, the following may be adopted. That is, the circuit current detection portion has a resistor element that converts a circuit current into a voltage and an average voltage detection portion that averages the voltage resulting from the conversion. The current drawing portion draws a current so that the average voltage detection portion has an output equal to a reference voltage (fifth configuration).

This configuration suppresses useless current drawing and thus can provide improved efficiency.

Furthermore, in the above-described third configuration, the switching current limit set voltage may be a voltage proportional to an input voltage of the LED drive circuit. This makes a switching current have a waveform of a shape proportional to the input voltage and thus can provide an improvement in power factor (sixth configuration).

Furthermore in the above-described third configuration, the LED current set voltage may be a voltage proportional to an average voltage of an input voltage of the LED drive circuit (seventh configuration). This allows light control to be performed over a wide range of phase angles.

Furthermore, in the above-described seventh configuration, the second offset voltage may be a voltage proportional to an edge voltage of the input voltage of the LED drive circuit. This can provide a light control curve approximating that of an incandescent lamp and thus allows light control to be performed in such a manner as to prevent a feeling of discomfort from being caused.

Furthermore, in the above-described seventh configuration, the second offset voltage may be a voltage obtained by squaring a voltage proportional to an edge voltage of the input voltage of the LED drive circuit. This can provide a light control curve further approximating that of an incandescent lamp and thus allows light control to be performed in such a manner as to further prevent a feeling of discomfort from being caused.

The present invention is also to provide an LED illumination component including: an LED drive circuit that can be connected to an alternating current power source via a phase-control light controller and drives an LED load; and the LED load that is connected on an output side of the LED drive circuit. The LED drive circuit includes: a switching power source portion having a switching element, a switching current detection portion that detects a switching current, and an LED current detection portion that detects an LED current; a first subtraction portion that subtracts a given value from an output of the switching current detection portion; a first determination portion that determines, through computation on a plurality of externally inputted voltages, a target value to which the switching current is limited; a second subtraction portion that subtracts a given value from an output of the LED current detection portion; a second determination portion that determines, through computation on a plurality of externally inputted voltages, a target value of the LED current; a switching current limitation portion that, if a result of the subtraction by the first subtraction portion exceeds a result of the determination by the first determination portion, switches off the switching element so as to limit the switching current; and an LED current control portion that drives the switching element to control the LED current so that a result of the subtraction by the second subtraction portion coincides with a result of the determination by the second determination portion.

FIG. 1 is a diagram showing an entire configuration of an LED illumination system according to one embodiment of the present invention.

FIG. 2 is a diagram showing one configuration example of an LED drive circuit according to the present invention.

FIG. 3 is a diagram showing one configuration example of an oscillator.

FIG. 4 is a diagram showing one configuration example of error amplifiers and a comparator.

FIG. 5 is a diagram showing one configuration example of error amplifiers.

FIG. 6 is a diagram showing waveforms of a voltage inputted to the LED drive circuit in a case where a phase angle of a light controller is made to vary.

FIG. 7 is a graph showing a relationship between the phase angle of the light controller and an input voltage average value.

FIG. 8 is a diagram showing one example of a circuit that generates an LED current set voltage and a second offset voltage.

FIG. 9 is a graph showing a relationship between each of an input voltage average value, a voltage proportional to an edge voltage, and a difference between them, and the phase angle.

FIG. 10 is a graph showing a light control curve of an incandescent lamp and light control curves of the present invention.

FIG. 11 is a diagram showing another example of the circuit that generates the LED current set voltage and the second offset voltage.

FIG. 12 is a graph showing a relationship between each of an input voltage average value, a square value of a voltage proportional to an edge voltage, and a difference between them, and the phase angle.

FIG. 13 is a graph showing a light control curve of the incandescent lamp and light control curves of the present invention.

FIG. 14 is a diagram showing another configuration example of the LED drive circuit according to the present invention.

FIG. 15 is a diagram showing a specific configuration example of the LED drive circuit shown in FIG. 14 .

FIG. 16 is a diagram showing a conventional example of an LED illumination system.

FIG. 17 is a diagram showing output waveforms of the light controller and output waveforms of a diode bridge in a case where the phase angle of the light controller varies.

FIG. 18 is a graph showing light control curves of an incandescent lamp and the conventional LED illumination system, respectively.

Hereinafter, an embodiment of the present invention will be described with reference to the appended drawings. FIG. 1 shows an entire configuration of an LED illumination system according to one embodiment of the present invention. As shown in FIG. 1 , the LED illumination system according to the present invention includes a commercial power source 100 , a phase-control light controller 200 , a fuse F 1 , a surge protection element NR 1 , a diode bridge DB 1 , an LED drive circuit 500 , and an LED array 400 . The commercial power source 100 is connected to the diode bridge DB 1 via the phase-control light controller 200 and the fuse F 1 , and the surge protection element NR 1 is connected between one end of the commercial power source 100 and one end of the fuse F 1 . The LED drive circuit 500 is connected to an output side of the diode bridge DB 1 , and the LED array 400 is connected to an output side of the LED drive circuit 500 . The LED drive circuit 500 and the LED array 400 constitute an LED illumination component, examples of which include an LED lamp.

The commercial power source 100 outputs a sinusoidal alternating current voltage that varies from country to country between 100 V to 250 V, and a frequency of 50 Hz or 60 Hz is used for the power source 100 . When an alternating current voltage is inputted to the phase-control light controller 200 , in accordance with the rotation or sliding operation for light control of a volume element, a waveform is generated that has a shape obtained by cutting away a certain phase point of an alternating current waveform. By the diode bridge DB 1 , full-wave rectification of an output waveform of the phase-control light controller 200 is performed, and a pulsation waveform having a frequency (100 Hz in a case of an input frequency of 50 Hz, and 120 Hz in a case of an input frequency of 60 Hz) double an input frequency is inputted to an input terminal T 0 of the LED drive circuit 500 .

Now, FIG. 2 shows one configuration example of the LED drive circuit 500 . The LED drive circuit 500 shown in FIG. 2 has a switching power source portion 1 , an LED current detection portion 3 , a switching control portion 4 , and input terminals T 0 to T 5 . The switching power source portion 1 is composed of an inductor L 1 , a diode D 1 , a capacitor C 1 , and a switching element M 1 that is an nMOS transistor so as to form a booster circuit, and an input voltage VIN of the input terminal T 0 is boosted by the switching power source portion 1 . A forward voltage of the LED array 400 is set to be about double the input voltage, and the boosted voltage causes a current to flow through the LED array 400 , and thus the LED array 400 emits light.

A switching current detection portion 2 that detects a switching current is connected between a source side of the switching element M 1 and a ground, and the LED current detection portion 3 that detects a current flowing through LEDs is connected between a cathode side of the LED array 400 and the ground.

The switching control portion 4 has error amplifiers AMP 1 to AMP 5 , comparators CMP 1 and CMP 2 , an RS flip-flop SRFF 1 , an oscillator OS 1 , and an AND circuit AND 1 . An output voltage IDET 1 of the switching current detection portion 2 is inputted to a non-inverting terminal of the error amplifier AMP 1 , and a reference voltage SENS applied to the input terminal T 1 is inputted to an inverting terminal thereof. Furthermore, a switching current limit set voltage ILIM applied to the input terminal T 2 is inputted to a non-inverting terminal of the error amplifier AMP 2 , and a first offset voltage VOS 1 applied to the input terminal T 3 is inputted to an inverting terminal thereof. The error amplifier AMP 1 amplifies a difference of the inputs and outputs the amplified difference to a non-inverting terminal of the comparator CMP 1 , and the error amplifier AMP 2 amplifies a difference between the inputs and outputs the amplified difference to an inverting terminal of the comparator CMP 1 . The comparator CMP 1 outputs a result of a comparison made between the inputs to a set terminal of the RS flip-flop SRFF 1 . A rectangular wave signal outputted by the oscillator OS 1 is inputted to a reset terminal of the RS flip-flop SRFF 1 , and an output from a Q-bar output terminal thereof is inputted to the AND circuit AND 1 .

Furthermore, an output voltage IDET 2 of the LED current detection portion 3 is inputted to a non-inverting terminal of the error amplifier 3 , and the reference voltage SENS applied to the input terminal T 1 is inputted to an inverting terminal thereof. Furthermore, an LED current set voltage ILED applied to the input terminal T 4 is inputted to a non-inverting terminal of the error amplifier AMP 4 , and a second offset voltage VOS 2 applied to the input terminal T 5 is inputted to an inverting terminal thereof. The error amplifier AMP 3 amplifies a difference between the inputs and outputs the amplified difference to an inverting terminal of the error amplifier AMP 5 , and the error amplifier AMP 4 amplifies a difference between the inputs and outputs the amplified difference to a non-inverting terminal of the error amplifier AMP 5 . The error amplifier AMP 5 amplifies a difference between respective output voltages of the error amplifiers AMP 3 and AMP 4 and outputs the amplified difference to a non-inverting terminal of the comparator CMP 2 . A triangle wave signal outputted by the oscillator OS 1 is inputted to an inverting terminal of the comparator CMP 2 . The comparator CMP 2 outputs a result of a comparison made between the inputs to the AND circuit AND 1 . Then, an output of the AND circuit AND 1 is inputted to a gate side of the switching element M 1 .

With the above-described configuration, when

(IDET1−SENS)>(ILIM−VOS1),

an output of the comparator CMP 1 becomes high in level and thus the RS flip-flop SRFF 1 is set, so that an output of the AND circuit AND 1 becomes low in level. Because of this, a gate of the switching element M 1 becomes low in level and thus the switching element M 1 is switched off to halt switching. As a result, a switching current is limited.

Furthermore, a PWM function takes place in which a duty ratio of an output pulse of the comparator CMP 2 varies depending on an output level of the error amplifier AMP 5 , and thus a gate voltage of the switching element M 1 is controlled in such a manner that feedback is performed so that

(IDET2−SENS)=(ILED−VOS2).

As a result, an LED current is maintained constant.

Now, for the proper operation of the phase-control light controller 200 , it is required that a current flow constantly so as to prevent a triac included in the phase-control light controller 200 from being switched off. In the present invention, voltages set appropriately on the basis of the input voltage VIN are applied to the input terminals T 1 to T 5 , respectively, so that a switching current and an LED current are controlled finely so as to prevent the triac included in the phase-control light controller 200 from being switched off. This can suppress a phenomenon in which the light controller is brought to an unstable state to cause brightness flickering of the LEDs and can also achieve increased efficiency. Typically, the switching current limit set voltage ILIM to be applied to the input terminal T 2 is set to be a given voltage, and the first offset voltage VOS 1 to be applied to the input terminal T 3 is set to be at a ground level. Furthermore, the reference voltage SENS to be applied to the input terminal T 1 is set to be at the ground level. Voltages to be applied to the input terminals T 4 and T 5 , respectively, will be described later.

The switching current limit set voltage ILIM to be applied to the input terminal T 2 may be set to be a voltage proportional to the input voltage VIN. This makes a switching current have a waveform of a shape proportional to the input voltage VIN and thus provides an improvement in power factor.

Furthermore, there is also a characteristic that the boosting operation causes the capacitor C 1 to store energy, and thus voltage application to the LED array 400 is stabilized, so that flickering is unlikely to occur.

FIG. 3 shows a specific configuration example of the oscillator OS 1 . The oscillator OS 1 has a terminal T 6 to which a digital direct current power source voltage DVCC is applied, a terminal T 7 to which an analog direct current power source voltage AVCC is applied, a terminal T 8 to which a reference voltage VREF is applied, a terminal T 9 to which a reference voltage VOSCH is applied, a terminal T 10 to which a reference voltage VOSCL (<VOSCH) is applied, and output terminals T 11 and T 12 . The oscillator OS 1 further has constant-current pMOS transistors M 2 , M 3 , and M 5 , constant-current nMOS transistors M 4 and M 6 , and an nMOS transistor M 7 . In addition thereto, the oscillator OS 1 has an operational amplifier AMP, a resistor R, a capacitor C 2 , comparators CMP 3 and CMP 4 , and an RS flip-flop SRFF 2 .

If an output voltage OUT of the output terminal T 11 attains a value not lower than the reference voltage VOSCH (=3 V), the RS flip-flop SRFF 2 is set and thus a Q-bar output thereof becomes low in level, so that the nMOS transistor M 7 is switched off and thus the constant-current nMOS transistor M 6 is switched on to cause electric charge accumulated in the capacitor C 2 to be discharged at a constant current IO. If, as a result of the discharge, the voltage of the capacitor C 2 is lowered to a value not higher than the reference voltage VOSCL (=1V), the RS flip-flop SRFF 2 is reset and thus the Q-bar output thereof becomes high in level, so that the nMOS transistor M 7 is switched on and thus the constant-current nMOS transistor M 6 is switched off. Thus, an operation of charging the capacitor C 2 at the constant current IO is performed so as to cause oscillation at a constant frequency. As a result, a triangle wave signal is outputted from the output terminal T 11 , and a rectangular wave signal is outputted from the output terminal T 12 . The triangle wave signal is inputted to an inverting input of the comparator CMP 2 , and the rectangular wave signal is inputted to a reset terminal of the RS flip-flop SRFF 1 ( FIG. 2 ).

FIG. 4 shows a specific configuration example of the error amplifiers AMP 1 and AMP 2 . The error amplifier AMP 1 is a transconductance amplifier composed of MOS transistors M 8 to M 15 , resistors R 1 to R 5 , and a capacitor C 3 . In the error amplifier AMP 1 , a difference between the switching current detection portion output voltage IDET 1 and the reference voltage SENS is converted into a variation in current, which is then converted into a voltage by the resistor R 4 . The variation in current flowing through the resistor R 4 is expressed by

Δ I 1=(2/ gm+R 2)×(IDET1−SENS)

where gm=a transconductance of the MOS transistors M 11 and M 13 , and the voltage resulting from the conversion by the resistor 4 is expressed by

On the other hand, the error amplifier AMP 2 is a transconductance amplifier composed of MOS transistors M 16 to M 23 , resistors R 6 to R 10 , and a capacitor C 4 . In the error amplifier AMP 2 , a difference between the switching current limit set voltage ILIM and the first offset voltage VOS 1 is converted into a variation in current, which is then converted into a voltage by the resistor R 9 . The variation in current flowing through R 9 is expressed by

Δ I 2=(2/ gm+R 7)×(ILIM−VOS1)

where gm=a transconductance of the MOS transistors M 19 and M 21 , and the voltage resulting from the conversion by the resistor R 9 is expressed by

Assuming that R 3 =R 4 =R 8 =R 9 =RC, and R 2 =R 7 =RE,

ΔV 1 =RC ×(2/ gm+RE )×(IDET1−SENS)

ΔV 2 =RC ×(2/ gm+RE )×(ILIM−VOS1)

and a comparison is made between ΔV 1 and ΔV 2 by the comparator CMP 1 . In the configuration shown in FIG. 2 , if ΔV 1 >ΔV 2 , i.e. (IDET 1 −SENS)>(ILIM−VOS 1 ), an output of the comparator CMP 1 becomes high in level, so that the gate of the switching element M 1 becomes low in level, and thus a switching current is limited.

FIG. 5 shows a specific configuration example of the error amplifiers AMP 3 , AMP 4 , and AMP 5 . The error amplifier AMP 3 is a transconductance amplifier composed of MOS transistors M 16 to M 23 , resistors R 11 to R 15 , and a capacitor C 5 . In the error amplifier AMP 3 , a difference between the LED current detection portion output voltage IDET 2 and the reference voltage SENS is converted into a variation in current, which is then converted in to a voltage by the resistor R 14 . The variation in current flowing through the resistor R 14 is expressed by

ΔI 3=(2/ gm+R 12)×(IDET2−SENS)

where gm=a transconductance of the MOS transistors M 19 and M 21 , and the voltage resulting from the conversion by the resistor R 14 is expressed by

On the other hand, the error amplifier AMP 4 is a transconductance amplifier composed of MOS transistors M 24 to M 31 , resistors R 16 to R 20 , and a capacitor C 6 . In the error amplifier AMP 4 , a difference between the LED current set voltage ILED and the second offset voltage VOS 2 is converted into a variation in current, which is then converted into a voltage by the resistor R 19 . The variation in current flowing through the resistor R 19 is expressed by

ΔI 4=(2/ gm+R 17)×(ILED−VOS2)

where gm=a transconductance of the MOS transistors M 27 and M 29 , and the voltage resulting from the conversion by the resistor R 19 is expressed by

Assuming that R 13 =R 14 =R 18 =R 19 =RC, and R 12 =R 17 =RE,

ΔV 3 =RC×( 2 /gm+RE )×(IDET2−SENS)

ΔV 4=RC×(2 /gm+RE )×(ILED−VOS2).

The error amplifier AMP 5 is composed of buffers BUF 1 and BUF 2 , resistors R 21 to R 24 , capacitors C 7 and C 8 , and operational amplifiers AMP 6 and AMP 7 and amplifies a difference in output voltage between the error amplifiers AMP 3 and AMP 4 .

Assuming that R 21 =R 22 =RIN, R 23 =R 24 =RF, and VREF=1V, an output voltage VOUT is expressed by

In the configuration shown in FIG. 2 , through a boosting switching operation, feedback is performed so that VOUT=VREF, that is,

ΔV 3 =ΔV 4, i.e. (ILED−VOS2)=(IDET2−SENS),

and thus an LED current can be controlled so as to be constant.

As described above with reference to FIGS. 4 and 5 , a transconductance amplifier is used to which a signal at a level equal to or higher than a GND level can be inputted, and thus the circuit can be simplified.

Next, the following describes light control performed in the present invention. FIG. 6 shows waveforms of a voltage inputted to the LED drive circuit 500 in cases where the phase angle of the phase-control light controller 200 is 0°, 45°, 90°, and 135°, respectively. Assuming an input voltage of 100 Vrms, a voltage peak value Vo is about 141.4 V, and an average voltage at these phase angles has values of 90.0 V, 76.8 V, 45.0 V, and 13.2 V, respectively. A phase angle θ of the phase-control light controller and the average voltage has a relationship expressed by

Average voltage= Vo (1+cos(θ))/π.

FIG. 7 plots the relationship between the phase angle of the phase-control light controller and the average voltage. A voltage proportional to this average voltage is inputted as the LED current set voltage ILED to the input terminal T 4 (the second offset voltage VOS 2 is assumed to be at the ground level), so that an LED current commensurate with an average value of the input voltage is obtained, and thus light control of the LEDs can be performed so as to correspond to the phase angle of the light controller. A dotted line in FIG. 10 shows a light control curve obtained in this case. A comparison between this light control curve and the light control curve of the conventional LED illumination system shown in FIG. 18 indicates that the above-described case allows light control to be performed over a wider range of values of the phase angle θ. In order that a voltage proportional to the average value of the input voltage is outputted as the LED current set voltage ILED, the following configuration is adopted. That is, as shown in FIG. 8 , the input voltage VIN is divided between resistors R 25 and R 26 , and a resulting voltage is smoothed by the capacitor C 9 .

Moreover, it is possible that, as shown in FIG. 8 , the input voltage VIN is divided between R 27 and R 28 , and based on a resulting voltage, a voltage proportional to a voltage (edge voltage, see FIG. 6 ) of a rising part of the input voltage waveform is generated by an edge detection circuit 5 and a sample holding circuit 6 , which is then inputted as the second offset voltage VOS 2 to the input terminal T 5 .

The edge voltage is expressed by

Edge voltage=Vo×sin(θ), and a voltage proportional thereto is expressed by

A×edge voltage=A×Vo×sin(θ) (A represents a proportionality constant). Hence, a light control curve obtained in this case is proportional to

( Vo ×(1+cos(θ))/π)−( A×Vo ×sin(θ)).

FIG. 9 shows a relationship between each of the input voltage average value, a voltage proportional to the edge voltage, and a difference between them, and the phase angle in a case where A=0.15 and Vo=141.4 V. Furthermore, a solid line in FIG. 10 shows a light control curve obtained in a case where a voltage proportional to the edge voltage is used. As is indicated by this, a property further approximating that represented by a light control curve of an incandescent lamp is obtained.

In addition, it is also possible that, as shown in FIG. 11 , at an output of the sample holding circuit 6 , a square circuit 7 is provided so as to generate a voltage obtained by squaring a voltage proportional to the edge voltage of the input voltage waveform, which is then inputted as the second offset voltage VOS 2 to the input terminal T 5 .

A light control curve obtained in this case is proportional to

( Vo ×(1+cos(θ))/π)−( A×Vo ×sin(θ)) 2 .

FIG. 12 shows a relationship between each of the input voltage average value, a square value of a voltage proportional to the edge voltage, and a difference between them, and the phase angle in a case where A 2 =0.15/141.4 and Vo=141.4 V. Furthermore, a solid line in FIG. 13 shows a light control curve obtained in a case where a square value of a voltage proportional to the edge voltage is used. As is indicated by this, compared with the light control curve shown by the solid line in FIG. 10 , this light control curve, at a part thereof representing low illuminance at large phase angles, further approximates the light control curve of the incandescent lamp.

As discussed above, when mounted to an already-existing light controller for an incandescent lamp, the LED illumination component of the present invention can provide light control properties similar to those provided by an incandescent lamp, and thus light control thereof causes no feeling of discomfort.

Furthermore, FIG. 14 shows a configuration of an LED drive circuit according to another embodiment of the present invention. An LED drive circuit 500 ′ shown in FIG. 14 has a configuration in which a circuit current detection portion 9 is inserted between a ground and each of one end of a switching current detection portion 2 and one end of an LED current detection portion 3 , and a current drawing portion 8 is inserted between an input voltage line and one end of the circuit current detection portion 9 . Furthermore, a detection signal of the circuit current detection portion 9 is inputted as a reference voltage SENS to each of error amplifiers AMP 1 and AMP 3 . The circuit current detection portion 9 detects currents in the entire circuit, and based on a detection signal obtained as a result of the detection, the current drawing portion 8 draws a current so that a total of a switching current, an LED current, and the drawn current has a constant value. This can suppress a phenomenon in which the triac included in the phase-control light controller 200 is switched off to bring light emission of the LEDs to an unstable state.

FIG. 15 shows a more specific configuration example of the LED drive circuit 500 ′. As the switching current detection portion 2 , a resistor RSW is used to convert a switching current flowing through a switching element M 1 into a voltage that is then inputted as a switching current detection portion output voltage IDET 1 to the error amplifier AMP 1 . Furthermore, as the LED current detection portion 3 , a resistor RLED is used to convert an LED current flowing through an LED array 400 into a voltage that is then inputted as an LED current detection portion output voltage IDET 2 to the error amplifier AMP 3 . Furthermore, the circuit current detection portion 9 that detects currents in the entire circuit is composed of a resistor RSENS and an average voltage detection portion 10 . The current drawing portion 8 has a reference voltage source VREF, an error amplifier AMP 8 , and an nMOS transistor M 2 . The average voltage detection portion 10 detects an average value of a voltage resulting from the voltage conversion by the resistor RSENS. The error amplifier AMP 8 amplifies a difference between the average voltage detected by the average voltage detection portion 10 and a reference voltage VREF and passes a constant current through the nMOS transistor M 2 so that driving the nMOS transistor M 2 allows a current having a constant average value to flow through the resistor RSENS. Control is thus performed so as to prevent the triac in the phase-control light controller from being switched off.

Due to a switching current being a pulsing current, a voltage across the resistor RSENS is increased and decreased in a cycle synchronized with switching. A voltage detected by the resistor RSENS, however, is smoothed by the average voltage detection portion 10 , so that useless current drawing is prevented, and thus efficiency of the LED drive circuit can be improved.