Resonant power conversion apparatus

1. Field of the Invention

The present invention relates to a resonant power conversion apparatus applicable to, for example, a hybrid vehicle inverter and an electric vehicle inverter.

2. Description of the Related Art

Among power conversion apparatuses, the resonant power conversion apparatus is known to be capable of reducing a switching loss. A typical example of the resonant power conversion apparatus is an auxiliary resonant commutated pole three-phase voltage inverter that reduces a switching loss by soft-switching switches through an auxiliary resonant circuit that performs resonant operation.

FIG. 1 is a circuit diagram illustrating a one-phase portion of an auxiliary resonant commutated pole three-phase voltage inverter according to a related art disclosed in Japanese Unexamined Patent Application Publication No. H08-340676. In FIG. 1 , both ends of a DC power source Vdc are connected to a series circuit including capacitors C 1 and C 2 having the same capacitance. A connection point of the capacitors C 1 and C 2 generates a voltage that is half a voltage of the DC power source Vdc.

Switches S 1 and S 2 are insulated gate bipolar transistors (IGBTs) and are connected in series. Both ends of the series circuit of the switches S 1 and S 2 are connected to the ends of the DC power source Vdc. The switch S 1 has a collector and an emitter between which a capacitor Cr 1 and a diode D 1 are connected. The switch S 2 has a collector and an emitter between which a capacitor Cr 2 and a diode D 2 are connected. A connection point of the switches S 1 and S 2 is connected to a load RL.

Connected between the connection point of the switches S 1 and S 2 and the connection point of the capacitors C 1 and C 2 is a series circuit including a reactor Lr and switches Sa 1 and Sa 2 . The switches S 1 a and Sa 2 form an auxiliary resonant circuit and are connected to diodes Da 1 and Da 2 , respectively.

The switches S 1 and S 2 carry out a zero-voltage, zero-current turn-ON operation with a resonant current provided by the auxiliary resonant circuit, to greatly reduce a switching loss.

FIG. 2 is a timing chart illustrating operating waveforms of the switches in the auxiliary resonant commutated pole three-phase voltage inverter illustrated in FIG. 1 . FIG. 3A is a mode transition diagram of the inverter illustrated in FIG. 1 from mode M 0 of state A to mode M 2 of state A. FIG. 3B is a mode transition diagram of the same from mode M 3 - 1 of state A to mode M 3 of state B.

In FIG. 2 , S 1 g is a gate signal applied to a gate of the switch S 1 , S 2 g is a gate signal applied to a gate of the switch S 2 , Sa 1 g is a gate signal applied to a gate of the switch Sa 1 , and Sa 2 g is a gate signal applied to a gate of the switch Sa 2 .

Operation of the resonant power conversion apparatus of FIG. 1 will be explained with reference to FIGS. 2 to 3B . In mode M 0 of state A, the switch S 1 is ON and the diode D 1 turns on to pass a current through a path extending along the load RL, the diode D 1 , and the DC power source Vdc.

In mode M 1 - 1 of state A, the gate signal Sa 1 g turns on the switch Sa 1 to divide a current from the load RL. A divided current portion passes through the diode D 1 to the DC power source Vdc and the other divided current portion passes through the reactor Lr, diode Da 2 , and switch Sa 1 , to accumulate power in the reactor Lr.

In mode M 1 - 2 of state A, a current of the switch S 1 and a current of the load RL (each arrowed) pass through the reactor Lr, diode Da 2 , and switch Sa 1 , to accumulate power in the reactor Lr.

In mode M 2 of state A, the gate signal S 1 g turns off the switch S 1 and the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 2 connected in parallel with the switch S 2 is discharged and the capacitor Cr 1 connected in parallel with the switch S 1 is charged.

In modes M 3 - 1 and M 3 - 2 of state A, the capacitor Cr 2 completes the discharge and the diode D 2 passes a current. When a drain-source voltage of the switch S 2 is zero, the gate signal S 2 g turns on the switch S 2 , to realize the zero-voltage switching of the switch S 2 .

In mode M 0 of state B, a current from the load RL passes through the switch S 2 . In mode M 1 of state B, the gate signal Sa 2 g turns on the switch Sa 2 , so that a current passing through a path extending along Da 1 , Sa 2 , and Lr from the left to the right together with a current of the load RL pass through the switch S 2 .

In mode M 2 of state B, the gate signal S 2 g turns off the switch S 2 . Then, the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 1 connected in parallel with the switch S 1 is discharged and the capacitor Cr 2 connected in parallel with the switch S 2 is charged.

In mode M 3 of state B, the capacitor Cr 1 completes the discharge and the diode D 1 passes a current. When a drain-source voltage of the switch 51 is zero, the switch S 1 is turned on, to realize the zero-voltage switching of the switch S 1 .

The resonant power conversion apparatus according to the related art of FIG. 1 , however, contains the voltage dividing point between the capacitors C 1 and C 2 , to divide a voltage of the DC power source Vdc. Potential at the voltage dividing point fluctuates depending on load conditions and temperature and this results in shortening the service life of the capacitors C 1 and C 2 .

The present invention provides a resonant power conversion apparatus capable of securing long service life and high reliability.

According to an aspect of the present invention, the resonant power conversion apparatus includes a transformer having a primary winding, a secondary winding, a tertiary winding connected in series with the secondary winding, and a reset winding connected in series with the primary winding; a first series circuit connected between first and second ends of a DC power source and including a first switch and a second switch; a first capacitor and a first diode that are connected in parallel with the first switch; a second capacitor and a second diode that are connected in parallel with the second switch; a second series circuit connected between the first and second ends of the DC power source and including the primary winding of the transformer and a third diode; a third series circuit connected between the first and second ends of the DC power source and including the reset winding of the transformer and a fourth diode; a reactor connected between a connection point of the first and second switches and a connection point of the secondary and tertiary windings of the transformer; a first auxiliary switch connected between the first end of the DC power source and a first end of the secondary winding of the transformer; a second auxiliary switch connected between the second end of the DC power source and a first end of the tertiary winding of the transformer; and a controller configured to alternately turn on the first and second switches and alternately turn on the first and second auxiliary switches, so that the first and second switches each conduct a zero-voltage switching operation.

FIG. 1 is a circuit diagram illustrating a one-phase portion of an auxiliary resonant commutated pole three-phase voltage inverter according to a related art;

FIG. 2 is a timing chart illustrating operating waveforms of switches in the inverter of FIG. 1 ;

FIG. 3A is a mode transition diagram illustrating mode M 0 of state A to mode M 2 of state A of the inverter of FIG. 1 ;

FIG. 3B is a mode transition diagram illustrating mode M 3 - 1 of state A to mode M 3 of state B of the inverter of FIG. 1 ;

FIG. 4 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 1 of the present invention;

FIG. 5 is a timing chart illustrating operating waveforms in states A and B of Embodiment 1;

FIG. 6 is a mode transition diagram illustrating states A and B of Embodiment 1;

FIG. 7 is a timing chart illustrating operating waveforms in states C and D of Embodiment 1;

FIG. 8 is a mode transition diagram illustrating states C and D of Embodiment 1;

FIG. 9 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 2 of the present invention;

FIG. 10 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 3 of the present invention; and

FIG. 11 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 4 of the present invention.

Resonant power conversion apparatuses according to embodiments of the present invention will be explained in detail with reference to the drawings.

FIG. 4 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 1 of the present invention. This apparatus is an auxiliary resonant commutated pole three-phase voltage inverter employing transformers and FIG. 4 illustrates a one-phase portion of the apparatus.

In FIG. 4 , the transformer T 1 has a primary winding n 1 , a secondary winding n 2 , a tertiary winding n 3 connected in series with the secondary winding n 2 , and a reset winding nR connected in series with the primary winding n 1 . The reset, primary, secondary, and tertiary windings nR, n 1 , n 2 , and n 3 of the transformer T 1 have a turn ratio of 2:2:1:1.

Ends of a DC power source Vdc are connected to a first series circuit including a switch S 1 (first switch) of an insulated gate bipolar transistor and a switch S 2 (second switch) of an insulated gate bipolar transistor.

The switch S 1 has a collector and an emitter between which a capacitor Cr 1 (first capacitor) and a diode D 1 (first diode) are connected. The switch S 2 has a collector and an emitter between which a capacitor Cr 2 (second capacitor) and a diode D 2 (second diode) are connected.

The ends of the DC power source Vdc are also connected to a second series circuit including the primary winding n 1 of the transformer and a diode Dn 1 (third diode), as well as to a third series circuit including the reset winding nR of the transformer T 1 and a diode DR (fourth diode).

A connection point “b” of the switches S 1 and S 2 and a connection point “a” of the secondary and tertiary windings n 2 and n 3 of the transformer T 1 are connected to a reactor Lr. The connection point “b” of the switches S 1 and S 2 is connected to a load RL.

A positive electrode of the DC power source Vdc and a first end of the secondary winding n 2 of the transformer T 1 are connected to a switch S 10 (first auxiliary switch) that operates in a direction in which a current flows from a positive electrode line of the DC power source Vdc. The switch S 10 includes a series circuit of a switch Sa 1 (third switch) of an insulated gate bipolar transistor and a diode Da 1 (fifth diode).

A negative electrode of the DC power source Vdc and a first end of the tertiary winding n 3 of the transformer T 1 are connected to a switch S 20 (second auxiliary switch) that operates in a direction in which a current flows from a negative electrode line of the DC power source Vdc. The switch S 20 includes a series circuit of a switch Sa 2 (fourth switch) made of an insulated gate bipolar transistor and a diode Da 2 (sixth diode).

The reset winding nR and diode DR form a reset circuit to discharge energy from an exciting inductance of the transformer T 1 . With the reset circuit, counter electromotive force generated by the exciting inductance of the transformer during an OFF period of the switch Sa 1 or Sa 2 is returned to the DC power source Vdc, thereby greatly reducing a voltage applied to the switches Sa 1 and Sa 2 .

A controller 10 alternately turns on the switches S 1 and S 2 and alternately turns on the switches Sa 1 and Sa 2 . More precisely, before a period in which the switches S 1 and S 2 are both OFF, the controller 10 turns on one of the switches Sa 1 and Sa 2 , to realize a zero-voltage switching operation of the switches S 1 and S 2 by achieving a resonance of the capacitors Cr 1 and Cr 2 and reactor Lr during the OFF period of the switches S 1 and S 2 .

Operation of the resonant power conversion apparatus according to Embodiment 1 will be explained with reference to FIGS. 5 and 6 in which FIG. 5 illustrates operating waveforms of the apparatus in states A and B and FIG. 6 illustrates mode transition of the apparatus in states A and B. In FIG. 6 , only operating parts of the apparatus are illustrated with solid lines.

In FIG. 5 , S 1 g is a gate signal applied to a gate of the switch S 1 , S 2 g is a gate signal applied to a gate of the switch S 2 , Sa 1 g is a gate signal applied to a gate of the switch Sa 1 , Sa 2 g is a gate signal applied to a gate of the switch Sa 2 , Vdc is a voltage of the DC power source Vdc, iLr is a current passing through the reactor Lr, ix is a current of the load RL, Vs 1 is a collector-emitter voltage of the switch S 1 , Vs 2 is a collector-emitter voltage of the switch S 2 , is 1 is a current passing through the switch S 1 or the diode D 1 , is 2 is a current passing through the switch S 2 or the diode D 2 .

In mode M 0 of state A, the switch S 1 is OFF and the switch S 2 is ON. The current is 2 (negative current) passes through the diode D 2 to the load RL. The load current ix has a constant value.

In mode M 1 - 1 of state A, the gate signal Sa 1 g turns on the switch Sa 1 , to pass the current iLr through a path extending along Vdc, Sa 1 , Da 1 , n 2 , and Lr. The currents iLr and is 2 (negative current) pass through the load RL. At this time, a current passes counterclockwise through a path extending along n 1 , Vdc, Dn 1 , and n 1 . The primary winding n 1 receives a voltage from the DC power source Vdc. The primary, secondary, and tertiary windings n 1 , n 2 , and n 3 have a turn ratio of 2:1:1, and therefore, one half of the voltage of the DC power source Vdc is applied to the connection point “a” of the secondary and tertiary windings n 2 and n 3 . In mode M 1 - 2 of state A, the current iLr exceeds the current ix, and then, the current is 2 (positive current), i.e., the difference between the current iLr and the current ix passes through the switch S 2 .

In mode M 2 of state A, the gate signal S 2 g turns off the switch S 2 and the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 2 is charged and the capacitor Cr 1 is discharged. This results in increasing the voltage Vs 2 across the switch S 2 and decreasing the voltage Vs 1 across the switch S 1 .

In mode M 3 - 1 of state A, a voltage Vcr 2 across the capacitor Cr 2 becomes larger than the voltage Vdc, and then, the diode D 1 turns on to pass the current is 1 (negative current). At this time, the gate signal S 1 g turns on the switch 51 , thereby realizing the zero-voltage switching of the switch S 1 . In mode M 3 - 2 of state A, the current iLr becomes smaller than the current ix, and then, the current is 1 (positive current), which is obtained by subtracting the current iLr from the current ix, passes through the switch S 1 .

In mode M 0 of state B, the current iLr becomes zero. The gate signal Sa 1 g turns off the switch Sa 1 and only the current is 1 passes through the load RL. When the switch Sa 1 turns off, the exciting inductance of the transformer T 1 produces counter electromotive force, which is returned to the DC power source Vdc through the diode DR and reset winding nR, to greatly reduce a voltage applied to the switch Sa 1 .

In mode M 1 of state B, the gate signal Sa 2 g turns on the switch Sa 2 , to pass the current iLr through a path extending along S 1 , Lr, n 3 , Da 2 , and Sa 2 . This increases the current is 1 passing through the switch S 1 . At this time, a current passes counterclockwise through a path extending along n 1 , Vdc, Dn 1 , and n 1 . The primary winding n 1 receives the voltage of the DC power source Vdc. The primary, secondary, and tertiary windings n 1 , n 2 , and n 3 have a turn ratio of 2:1:1, and therefore, one half of the DC power source voltage Vdc is applied to the connection point “a” of the secondary and tertiary windings n 2 and n 3 .

In mode M 2 of state B, the gate signal S 1 g turns off the switch S 1 and the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 1 is charged and the capacitor Cr 2 is discharged. This results in increasing the voltage Vs 1 across the switch S 1 and decreasing the voltage Vs 2 across the switch S 2 .

In mode M 3 of state B, a voltage Vcr 1 across the capacitor Cr 1 becomes larger than the DC power source voltage Vdc and the diode D 2 turns on to pass the current is 2 (negative current). The gate signal S 2 g turns on the switch S 2 to pass the current is 2 . In mode M 0 of state A, the current iLr becomes zero and the gate signal Sa 2 g turns off the switch Sa 2 , so that only the current is 2 passes through the load RL. When the switch Sa 2 turns off, the exciting inductance of the transformer T 1 produces counter electromotive force, which is returned to the DC power source Vdc through the diode DR and reset winding nR, to greatly reduce a voltage applied to the switch Sa 2 .

Operation of the resonant power conversion apparatus of Embodiment 1 in states C and D will be explained in detail with reference to operating waveforms illustrated in FIG. 7 and a mode transition diagram illustrated in FIG. 8 . In FIG. 8 , only operating parts of the apparatus are illustrated with solid lines.

In mode M 0 of state C, the switch S 2 is OFF and the switch S 1 is ON. The diode D 1 passes the current is 1 that is the same as the current passing through the load RL. The load current ix has a constant value.

In mode M 1 - 1 of state C, the gate signal Sa 2 g turns on the switch Sa 2 to pass the current iLr through the path extending along Lr, n 3 , Da 2 , and Sa 2 . The current ix is divided into the currents iLr and is 2 . At this time, a current passes counterclockwise through the path extending along n 1 , Vdc, Dn 1 , and n 1 . The primary winding n 1 receives the DC power source voltage Vdc. The primary, secondary, and tertiary windings n 1 , n 2 , and n 3 have a turn ratio of 2:1:1, and therefore, one half of the DC power source voltage Vdc is applied to the connection point “a” of the secondary and tertiary windings n 2 and n 3 . In mode M 1 - 2 of state C, the current iLr becomes larger than the current ix and the current is 1 (positive current), which is obtained by subtracting the current ix from the current iLr, passes through the switch S 1 .

In mode M 2 of state C, the gate signal S 1 g turns off the switch S 1 and the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 1 is charged and the capacitor Cr 2 is discharged. This results in increasing the voltage Vs 1 across the switch S 1 and decreasing the voltage Vs 2 across the switch S 2 .

In mode M 3 - 1 of state C, the voltage Vcr 1 across the capacitor Cr 1 becomes larger than the power source voltage Vdc and the diode D 2 turns on to pass the current is 2 through the diode D 2 . At this time, the gate signal S 2 g turns on the switch S 2 , thereby realizing the zero-voltage switching of the switch S 2 . In mode M 3 - 2 of state C, the current iLr becomes smaller than the current ix and the current is 2 , which is obtained by subtracting the current iLr from the current ix, passes through the switch S 2 .

In mode M 0 of state D, the current iLr becomes zero and the gate signal Sa 2 g turns off the switch Sa 2 , so that only the current is 2 passes through the load RL. When the switch Sa 2 turns off, the exciting inductance of the transformer T 1 produces counter electromotive force, which is returned to the DC power source Vdc through the diode DR and reset winding nR, to greatly reduce a voltage applied to the switch Sa 2 .

In mode M 1 of state D, the gate signal Sa 1 g turns on the switch Sa 1 and the current iLr passes through a path extending along Sa 1 , Da 1 , n 2 , Lr, and S 2 , to increase the current is 2 of the switch S 2 . At this time, a current passes counterclockwise through the path extending along n 1 , Vdc, Dn 1 , and n 1 . The primary winding n 1 receives the DC power source voltage Vdc. The primary, secondary, and tertiary windings n 1 , n 2 , and n 3 have a turn ratio of 2:1:1, and therefore, one half of the voltage of the DC power source Vdc is applied to the connection point “a” of the secondary and tertiary windings n 2 and n 3 .

In mode M 2 of state D, the gate signal S 2 g turns off the switch S 2 and the reactor Lr and capacitors Cr 1 and Cr 2 resonate. At this time, the capacitor Cr 2 is charged and the capacitor Cr 1 is discharged. This results in increasing the voltage Vs 2 across the switch S 2 and decreasing the voltage Vs 1 across the switch S 1 .

In mode M 3 of state D, the voltage Vcr 2 across the capacitor Cr 2 becomes larger than the voltage Vdc and the diode D 1 turns on to pass the current is 1 . The gate signal S 1 g turns on the switch S 1 , to pass the current is 1 . In mode M 0 of state C, the current iLr becomes zero. The gate signal Sa 1 g turns off the switch Sa 1 and only the current is 1 passes through the load RL. When the switch Sa 1 turns off, the exciting inductance of the transformer T 1 produces counter electromotive force, which is returned to the DC power source Vdc through the diode DR and reset winding nR, to greatly reduce a voltage applied to the switch Sa 1 .

As mentioned above, the resonant power conversion apparatus according to Embodiment 1 employs the transformer T 1 instead of the two voltage dividing capacitors C 1 and C 2 of the related art illustrated in FIG. 1 , to achieve performance equivalent to that achieved with the capacitors of the related art. According to Embodiment 1, the primary, secondary, and tertiary windings n 1 , n 2 , and n 3 of the transformer T 1 have a turn ratio of 2:1:1, and therefore, one half of the DC voltage Vdc is applied to the connection point “a” of the secondary and tertiary windings n 2 and n 3 .

When one of the switches Sa 1 and Sa 2 is OFF, the exciting inductance of the transformer T 1 produces counter electromotive force, which is returned to the DC power source Vdc through the diode DR and reset winding nR, to greatly reduce a voltage applied to the switches Sa 1 and Sa 2 . This prevents the switches Sa 1 and Sa 2 from deteriorating or breaking, elongates the service life of the switches Sa 1 and Sa 2 , and improves the reliability of the resonant power conversion apparatus.

If there is no diode DR nor reset winding nR, the counter electromotive force produced by the exciting inductance of the transformer T 1 during an OFF period of any one of the switches Sa 1 and Sa 2 causes the secondary and tertiary windings n 2 and n 3 to generate voltages. These voltages are surge voltages applied to the switches Sa 1 and Sa 2 , to break the switches Sa 1 and Sa 2 . This problem never occurs in the resonant power conversion apparatus of Embodiment 1.

FIG. 9 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 2 of the present invention. This is a half-bridge resonant power conversion apparatus. Compared with Embodiment 1 illustrated in FIG. 4 , Embodiment 2 additionally employs capacitors C 10 and C 20 , a reactor Lf, and a capacitor Cf.

Both ends of a DC power source Vdc are connected to a series circuit including the capacitors C 10 and C 20 . Connected between a connection point of the capacitors C 10 and C 20 and a connection point “b′” of switches S 1 and S 2 is a series circuit (a harmonics removing filter circuit) including the reactor Lf and capacitor Cf. Both ends of the capacitor Cf are connected to a load RL, to provide the load RL with single-phase sinusoidal AC power. Effect of Embodiment 2 is similar to that of Embodiment 1.

FIG. 10 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 3 of the present invention. Embodiment 3 employs a single-phase, full-bridge configuration with respect to Embodiment 2 that employs a half-bridge configuration as illustrated in FIG. 9 .

Namely, the resonant power conversion apparatus of Embodiment 3 includes a transformer T 2 provided with a primary winding n 1 , secondary windings n 2 and n 4 , tertiary windings n 3 and n 5 , and a reset winding nR and diodes DR and Dn 1 . In addition, the apparatus of Embodiment 3 employs, for U-phase, switches SU 1 , SU 2 , Sa 1 and Sa 2 , diodes Da 1 and Da 2 , capacitors Cr 1 and Cr 2 , and a reactor Lr 1 , and for V-phase, switches SV 1 , SV 2 , Sa 3 , and Sa 4 , diodes Da 3 and Da 4 , capacitors Cr 3 and Cr 4 , and a reactor Lr 2 .

A controller 10 b has the same function as the controller 10 of Embodiment 1 and controls ON/OFF of the switches SU 1 , SU 2 , Sa 1 and Sa 2 and the switches SV 1 , SV 2 , Sa 3 , and Sa 4 .

Effect of Embodiment 3 is similar to that of Embodiment 1.

FIG. 11 is a circuit diagram illustrating a resonant power conversion apparatus according to Embodiment 4 of the present invention. Embodiment 4 employs a three-phase, full-bridge configuration with respect to Embodiment 1 that employs a one-phase configuration as illustrated in FIG. 4 .

Namely, the resonant power conversion apparatus of Embodiment 4 includes a transformer T 3 provided with a primary winding n 1 , secondary windings n 2 , n 4 , and n 6 , tertiary windings n 3 , n 5 , and n 7 , and a reset winding nR and diodes DR and Dn 1 . In addition, the apparatus of Embodiment 4 includes, for U-phase, switches SU 1 , SU 2 , Sa 1 , and Sa 2 , diodes Da 1 and Da 2 , capacitors Cr 1 and Cr 2 , and a reactor Lr 1 , for V-phase, switches SV 1 , SV 2 , Sa 3 , and Sa 4 , diodes Da 3 and Da 4 , capacitors Cr 3 and Cr 4 , and a reactor Lr 2 , and for W-phase, switches SW 1 , SW 2 , Sa 5 , and Sa 6 , diodes Da 5 and Da 6 , capacitors Cr 5 and Cr 6 , and a reactor Lr 3 .

A controller 10 c has the same function as the controller 10 of Embodiment 1 and controls ON/OFF of the switches SU 1 , SU 2 , Sa 1 , and Sa 2 , switches SV 1 , SV 2 , Sa 3 , and Sa 4 , and switches SW 1 , SW 2 , Sa 5 , and Sa 6 .

Effect of Embodiment 4 is similar to that of Embodiment 1.

The present invention is not limited to the resonant power conversion apparatuses of Embodiments 1 to 4. According to Embodiment 1, the gate signal Sa 1 g to the switch Sa 1 is ON in modes M 1 , M 2 , and M 3 of state A and modes M 1 , M 2 , and M 3 of state D and the gate signal Sa 2 g to the switch Sa 2 is ON in modes M 1 , M 2 , and M 3 of state B and modes M 1 , M 2 , and M 3 of state C. Instead, the gate signal Sa 1 g to the switch Sa 1 may be ON in modes M 1 , M 2 , and M 3 of state A, mode M 0 of state B, modes M 1 , M 2 , and M 3 of state D, and mode M 0 of state C and the gate signal Sa 2 g to the switch Sa 2 is ON in modes M 1 , M 2 , M 3 of state B, mode M 0 of state A, modes M 1 , M 2 , and M 3 of state C, and mode M 0 of state D. This configuration may provides an effect similar to that of Embodiment 1.

As mentioned above, the resonant power conversion apparatus according to the present invention employs a transformer instead of two voltage dividing capacitors, to realize performance equivalent to that realized by the capacitors. Even if the exciting inductance of the transformer produces counter electromotive force during an OFF period of the first or second auxiliary switch, the present invention returns the counter electromotive force to a DC power source through the fourth diode and reset winding, to greatly reduce a voltage applied to the first and second auxiliary switches. The resonant power conversion apparatus according to the present invention, therefore, has long service life and high reliability.

The present invention is applicable to, for example, inverters for electric vehicles.

This application claims benefit of priority under 35 USC §119 to Japanese Patent Application No. 2009-199505, filed on Aug. 31, 2009, the entire contents of which are incorporated by reference herein. Although the invention has been described above by reference to certain embodiments of the invention, the invention is not limited to the embodiments described above. Modifications and variations of the embodiments described above will occur to those skilled in the art, in light of the teachings. The scope of the invention is defined with reference to the following claims.