Phase lock loop (PLL) with gain control

The present invention relates generally to a structure for a voltage controlled oscillator (VCO) in a phase lock loop (PLL), and more particularly to a structure for a VCO with gain control.

A phase lock loop (PLL) is an electronic system that locks in phase and frequency of an output signal to the phase and frequency of an input signal. A PLL is widely employed in applications in communication systems, such as FM demodulators, stereo demodulators, tone detectors, and frequency synthesizers. A PLL is also commonly employed in digital applications that require a high-frequency periodic signal to synchronize the events between high-performance digital circuits. A PLL is particularly desirable to be implemented by advanced very-large-scale-integrated-circuit (VLSI) manufacturing technologies, and integrated with semiconductor integrated circuits (IC) for the various applications fields.

FIG. 1A is a system diagram illustrating the building blocks of an existing PLL, which generally is a nonlinear feedback circuit. The input signal F in is typically a periodic clock signal generated off chip from a reference clock source, such as a crystal oscillator. F in is compared with a local clock signal F local , which is typically a divided version of the output signal F out . The phase detector determines the relative phase difference between the two signals and outputs a signal that is proportional to this phase difference. The output signal from the phase detector is subsequently fed into a charge pump that converts the signal into an analog voltage V c . This analog voltage is typically used as the VCO control signal. When there is a phase difference between the input signal F in and the local signal F local , the value of this analog voltage may increase or decrease to speed up or slow down the VCO, which causes the local signal F local to catch up with the input signal F in or to eliminate the lead of the local signal F local . When an automatic follow-up between the input signal F in and the local signal F local is achieved, the output signal F out is said to be locked on the input signal F in . This behavior makes PLLs particularly useful in applications where an input signal contains desired information, whereas its frequency varies in time. In practice, the analog voltage V c generated by the charge pump first passes a loop filter, typically a low-pass filter, where the high-frequency components are removed from the VCO control signal. The dc component of V c is then fed into the VCO in order to reduce undesirable jitter in the output signal F out .

FIG. 1B is a graph of an output frequency vs. control voltage of the VCO illustrated in FIG. 1A . The linear relationship between the frequency of the output signal F out and the magnitude of the VCO control signal V c may be expressed as the following equation:

F out =K vco ·V c   (1)

where K vco represents the slope of the linear output frequency vs. control voltage relationship, and is the constant VCO frequency gain. F out is the frequency change in the output signal in response to a VCO control signal V c . As an example, a VCO in a PLL fabricated through a 0.25 μm processing technology has a constant frequency gain of about 0.25 MHz/mV, where one millivolt swing in the VCO control voltage translates into a quarter of one MHz frequency shift in the output signal F out .

There are applications where a VCO with fine tuning precision of output frequency is desirable. A PLL having a VCO frequency gain such as that described above provides too coarse a tuning precision to meet the requirements of these applications. Thus, a PLL have a significantly reduced VCO frequency gain is also desirable.

These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by preferred embodiments of the present invention which provide a Phase Lock Loop (PLL) with gain control. The PLL has a dual-path configuration, where a first and a second VCO control voltage are generated in response to a phase or frequency difference between a PLL input signal and an output signal. The PLL comprises a dynamic voltage gain control (DVGC) unit and a voltage-to-current (V2I) unit, where the DVGC creates a baseline reference current in response to the first VCO control voltage and the V2I provides a substantially linear current in response to the second VCO control voltage. The currents from the DVGC and V2I are combined and fed into a current-controlled oscillator, which generates a PLL output frequency signal. Frequency gain of the VCO is substantially reduced, thus providing a PLL with improved tuning precision.

In accordance with a preferred embodiment of the present invention, a phase lock loop (PLL) circuit comprises a phase detector configured to generate a control signal representing a frequency difference between a PLL input signal and an output signal. The phase lock loop circuit also comprises a loop filter configured to generate first and second voltage signals in response to the control signal. The phase lock loop circuit further comprises a voltage controlled oscillator (VCO) configured to modify a frequency of the output signal in response to the first and second voltage signals.

In accordance with another preferred embodiment of the present invention, a voltage controlled oscillator (VCO) comprises a dynamic voltage gain control (DVGC) unit configured to generate a first current, and a voltage-to-current unit configured to generate a second current. The first current varies insignificantly in response to a first voltage signal, while the second current varies substantially linearly in response to a second voltage signal.

In accordance with a further preferred embodiment of the present invention, a phase lock loop (PLL) comprises a loop filter configured to provide a first and a second voltage signal in response to an input control voltage, and a voltage controlled oscillator (VCO) configured to generate a PLL output frequency signal in response to the first voltage signal and the second voltage signal. The input control voltage varies in response to a relative phase of a PLL input frequency signal and the PLL output frequency signal.

For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawing, in which:

FIGS. 1A and 1B illustrate a block diagram and frequency gain of a prior art PLL;

FIGS. 2A and 2B illustrate a block diagram and frequency gain of a prior art PLL;

FIG. 3 is a block diagram of a PLL of an illustrative embodiment;

FIGS. 4A and 4B illustrate a schematic diagram of a dynamic voltage gain control (DVGC) of an illustrative embodiment;

FIGS. 4C and 4D illustrate the operating characteristics of DVGC of an illustrative embodiment;

FIG. 5A illustrates a detailed schematic diagram of an voltage-controlled oscillator (VCO) of an illustrative embodiment;

FIG. 5B illustrates the operating characteristics of the VCO illustrated in FIG. 5A ;

FIG. 6A compares the tuning precision of a prior art PLL and PLL of an illustrative embodiment;

FIG. 6B compares the noise characteristics of a prior art PLL and PLL of an illustrative embodiment;

FIG. 7 illustrates the PLL setting behavior of an illustrative embodiment;

FIG. 8 illustrates a detailed schematic diagram of an voltage-controlled oscillator (VCO) of an illustrative embodiment;

FIG. 9 illustrates the PLL setting behavior of an illustrative embodiment; and

FIG. 10 illustrates a detailed schematic diagram of a DVGC of an illustrative embodiment.

The making and using of the presently preferred embodiments are discussed in detail below. It should be appreciated, however, that the present invention provides many applicable inventive concepts that can be embodied in a wide variety of specific contexts. The specific embodiments discussed are merely illustrative of specific ways to make and use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferred embodiments in a specific context, namely phase lock loops (PLLs) with a voltage controlled oscillator (VCO) that has a reduced VCO frequency gain, providing an improved tuning precision in the VCO output signal frequency in correspondence with a VCO control voltage within a broad tuning range. Embodiments of the present invention may also be applied, however, to VCO structures used in other applications where improved tuning precision in the VCO output signal frequency is desired. While CMOS processing technology is employed to fabricate the VCOs and PLLs in the preferred embodiments, other suitable processing technologies, such as bipolar and BiCMOS, may be also used to implement PLLs in the preferred embodiments.

FIG. 2A shows a block diagram of a prior art PLL where VCO 10 is configured to provide a reduced VCO gain (e.g., K vco ), thus enabling improved output signal F out frequency tuning precision. VCO 10 comprises a voltage-to-current converter 15 . Voltage-to-current converter 15 is configured to generate a dc current I c in response to the VCO control signal V c that is typically generated from the loop filter. VCO 10 also comprises current controlled oscillator (CCO) 25 , which is configured to generate the output signal F out in response to an input dc control current. As an attempt to reduce the frequency gain of VCO 10 , an independent current source, such as a known constant G m current source 20 as shown, is implemented to provide a constant dc current I ref to VCO 10 . I c and I ref are combined in VCO 10 , and the combined current I cco is subsequently fed into CCO 25 . Under this prior art VCO configuration, the frequency of output signal F out varies in response to I cco . Even if a null control signal V cont is provided to VCO 10 , the VCO still outputs an offset frequency signal that corresponds to the constant dc current I ref generated from current source 20 .

FIG. 2B shows a linear frequency gain of VCO 10 illustrated in FIG. 2A . The linear relationship between the output frequency signal F out and the magnitude of the VCO control signal V c may be expressed as the following equation:

F out =K vco ·V c +F o   (2)

where F o is the VCO output signal frequency when no VCO control signal is provided to VCO 10 . K vco represents the slope of the linear output frequency F out vs. VCO control voltage V c . K vco is a constant VCO frequency gain under the current prior art VCO configuration, which may be significantly smaller than that of a counterpart VCO without output signal frequency offset F o .

Although prior art VCO 10 provides reduced VCO frequency gain, it has drawbacks in at least the following aspects. Firstly, implementing constant G m current source 20 requires an increased PLL die area, which is typically undesirable in advanced technology. Secondly, a PLL is an analog circuit that is inherently sensitive to noise and interface. Implementing constant G m current source 20 may introduce noise into VCO 10 that may result in jitter in the resulting output frequency signal. Moreover, it is difficult to remedy the noise thus introduced into the VCO because current source 20 is an open loop system, independent from the PLL feedback loop. The automatic remedy mechanism typically provided by a feedback loop is not viable for this prior art VCO configuration. Thirdly, implementing constant G m current source 20 into VCO 10 may have a negative impact on the operational stability of a pre-fine-tuned feedback system of a PLL. Frequency compensation involving addition of circuit elements is typically required to avoid the stability problem stemmed from adding constant G m current source 20 into a pre-designed PLL.

FIG. 3 illustrates a block diagram of PLL 100 in accordance with a preferred embodiment of the present invention. PLL 100 comprises phase detector 110 , charge pump 120 , loop filter 130 , VCO 140 , and feedback divider 190 . Loop filter 130 is preferably a low-pass filter that extracts the dc component of the input analog voltage signal V coin from charge pump 120 , applying an output voltage V mid to VCO 140 . In one preferred embodiment, a low-pass filter 130 comprises a circuit between the input voltage signal V coin and the ground GND, and the circuit includes a first capacitance C 1 , a resistance R 2 and a second capacitance C 2 in a “π” configuration. R 2 has a resistance of about 10 k; C 1 has a capacitance of about 7 pF; C 2 has a capacitance of about 120 pF, significantly great than C 1 . VCO 140 comprises a voltage-to-current converter 145 . Voltage-to-current converter (V2I) 145 is configured to generate a dc current I v2i in response to a first VCO control signal V coin that is generated from charge pump 120 and coupled to VCO 140 via loop filter 130 . It is noted that V coin is substantially free of ac components after passing through C 1 of loop filter 130 . VCO 140 also comprises dynamic voltage gain control (DVGC) unit 150 that is implemented to provide a dc current I offset in response to a second VCO control signal V mid that is the output voltage from loop filter 130 . Similarly, V mid is substantially free of ac components due to low-pass filter 130 . The dc current I v2i and I offset are combined at node 155 in VCO 140 , and the combined current I total is fed into CCO 160 that converts current I total into an output frequency signal F out . In short, VCO 140 in preferred embodiments is configured to respond to control signals from two signal paths (denoted as path ( 1 ) and ( 2 ) in FIG. 3 ), where path ( 1 ) is configured to take VCO control signal V coin , and path ( 2 ) is configured to take loop filter output signal V mid . CCO 160 is configured to accept the combined control current I total from V2I 145 and DVGC 150 . PLL 100 is therefore configured to generate the frequency signal F out in response to the control signals V coin and V mid . PLL 100 thus has a dual path configuration, in contrast with a prior art PLL with single path configuration as shown, for example, in FIG. 1A . This is also in contrast with the prior art PLL shown in FIG. 2A where, although an additional current path I ref is fed into the VCO, this current path is outside of the feedback loop of the PLL.

FIG. 4A illustrates one exemplary schematic diagram of DVGC 150 of PLL 100 . DVGC 150 is a tunable current source that provides a variable dc current I offset in response to control signal V mid . Preferably, DVGC 150 exhibits at least the following characteristics: the output current I offset being substantially independent of the supply voltage, the output current I offset being substantially independent of variations of manufacturing process and operational temperature, the transconductance Gm of the transistors in DVGC 150 being substantially independent of variations of supply voltage, manufacturing process, and operational temperature. In the current embodiment, DVGC 150 is implemented through a complementary metal-oxide-semiconductor (CMOS) process technology, and comprises a pair of NMOSFETs M 1 and M 2 , a pair of PMOSFETs M 3 and M 4 , a variable resistor Rcv 152 , and an additional PMOSFET M 5 . The sources of M 1 and M 2 are connected to ground GND. The drains of M 1 and M 2 are connected to the drains of M 3 and M 4 , respectively. The gate and the drain of M 1 , and the gate of M 2 are tied together. One terminal of variable resistor Rcv is connected to supply voltage V DD , and another terminal 153 of Rcv is connected to the source of M 3 . The sources of M 4 and M 5 are connected to voltage supply V DD . The gates of M 3 , M 4 , and M 5 , and the drain of M 4 are tied together. In the current embodiment of PLL 100 , V DD is equal to 2.5 V, although other suitable voltage supplies may be also used.

FIG. 4B illustrates a further detailed schematic diagram of variable resistor Rcv 152 . Rcv 152 is implemented between supply voltage V DD and the source 153 of PMOSFET M 3 . Rcv 152 comprises a first branch with resistor R main in parallel with a second branch with PMOSFET M 6 and resistor R base . One terminal of R main and the source of M 6 are connected to the supply voltage V DD . The drain of M 6 is connected to one terminal of R base . The other terminals of R main and R base are connected the source 153 of M 3 . The gate of M 6 is coupled to the output node of low pass filter 130 , where signal V mid is presented. The resistance of Rcv 152 is the resistance of the first branch in parallel with the resistance of the second branch, and can be expressed as the following:

Rcv=R main ∥( R base +R mos )  (3)

where R main and R base have an intermediate resistance of about 3 KΩ and 2 KΩ, respectively. R mos is the resistance of PMOSFET M 6 that varies in response to the gate-to-source voltage V GS of M 6 , which is also referred to as DVGC 150 control signal V cont . V cont represents the difference between V DD and V mid (i.e., V cont =V DD −V mid ). V cont varies in response to V mid presented on the gate of M 6 , and in turn, V mid varies in respond to the variation of signal V coin from charge pump 120 . M 6 operates in different operating regions in response to the magnitude of V mid . When V cont is below the threshold voltage of M 6 , M 6 is turned off and operates in the cutoff region. M 6 is turned on and operates in the saturation region when V cont increases and becomes greater than the threshold voltage of M 6 . M 6 enters triode (linear) operating region when V cont continues to increase. When M 6 operates across the aforementioned operating regions, R mos is reduced from an upper bound large resistance to a lower bound small resistance. Consequently, Rcv 152 varies in response to the variation of R mos , and as a result, output dc current I offset on PMOSFET M 5 of DVGC 150 varies in response to the variation of Rcv. In the current DVGC configuration, control voltage V cont is referred to as being V mid referenced to the supply voltage V DD .

FIG. 4C illustrates the relationship between V cont and variable resistor R cv under the current R cv configuration. It is shown that R cv remains at a substantially constant, upper bound value of about 3.05 KΩ, for example, when V cont is below the threshold voltage 0.8V of M 6 . This corresponds to an upper bound VCO control signal V mid close to V DD . R cv is reduced linearly when V cont is greater than 0.8 V. R cv reaches a lower bound value of about 2.25 KΩ, for example, when V cont reaches its upper limit, which corresponds to a lower bound control signal V mid of about 1.2 V.

FIG. 4D is a graph illustrating the operating characteristics of DVGC 150 in the current embodiment. The output current I offset of DVGC 150 is plotted versus control signal V cont . It is shown that, when V cont is below the threshold voltage 0.8V of M 6 , I offset remains at a lower bound, baseline current offset base of about 40 μA, for example. I offset is increased linearly when V cont is greater than 0.8 V. I offset reaches an upper bound value of about 170 μA, for example, when V cont reaches its upper limit.

FIG. 5A illustrates a detailed schematic diagram of loop filter 130 and VCO 140 implemented with, among others, V2I 145 , DVGC 150 , and CCO 160 . V2I 145 comprises PMOSFET M 7 that is implemented to convert the first VCO control signal V coin into a dc current I v2i . The source of M 7 is connected to supply voltage V DD ; the gate of M 7 is coupled to the first VCO control signal V coin . Within the desired operating range of V coin , between 0 V and 2.5 V, for example, M 7 is configured to produce a continuous and substantially linear current I v2i proportional to V coin . The current I v2i is coupled to node 155 through the drain of M 7 . Similarly, the current I offset generated on PMOSFET M 5 of DVGC 150 is also coupled to node 155 . Current I v2i from V2I 145 and I offset from DVGC 150 are combined at node 150 to form CCO control current I total . I total is subsequently fed into CCO 160 and converted into output frequency signal F out of PLL 100 .

In the current embodiment, CCO 160 is implemented using a CMOS process technology, and comprises a CMOS ring oscillator 162 and a “level up” unit 164 . Ring oscillator 162 comprises a chain of five CMOS inverters INV 1 through INV 5 . The sources of the PMOSFETs of inverters INV 1 through INV 5 are coupled to node 155 . The drains of the NMOSFETs of inverters INV 1 through INV 5 are coupled to GND. The voltage at the output stage of INV 5 is fed back to the input stage of INV 1 . “Level up” unit 164 comprises two cascaded differential pairs D 1 and D 2 . The output signal V 3 from INV 3 and the output signal V 4 are coupled to the differential pairs D 1 and D 2 , respectively. “Level up” unit 164 is configured to convert the voltage difference between V 3 and V 4 into an output frequency signal F out at the output node of differential pair D 2 . Although a CMOS process technology is used to implement CCO 160 in the above example, it is noted, however, that other suitable process technologies, such as a bipolar process, and other CCO configurations may be also used to convert I total into output frequency signal F out .

FIG. 5B is a graph illustrating a comparison between the CCO control current T total (e.g., trace 210 ) as generated by PLL 100 and the COO control current (e.g., trace 200 ) of a prior art PLL having a single path configuration, such as that shown in FIG. 1A . Also shown in FIG. 5 B is the output current I offset (e.g., trace 212 ) generated from DVGC 150 of PLL 100 . Through the comparison, distinct features of PLL 100 are revealed and advantages may be readily appreciated as the following. First, compared with the significant increase of I total versus V coin , I offset 212 output from DVGC 150 provides a current reference that remains nearly unchanged in the desired operating range of V coin due to the large capacitance of C 2 of loop filter 130 . Moreover, I offset 212 provides a baseline reference current I offset base even when no VCO control signal V coin is presented. Second, in order to achieve a predetermined PLL frequency gain (equivalent to a corresponding current gain of CCO, e.g., slope of trace 200 ), the current gain of V2I 145 (e.g., slope of trace 210 ) can be configured substantially smaller than that of a V2I in a prior art VCO without a DVGC unit. This may significantly increase the tuning precision of a PLL.

FIG. 6A compares the tuning precision (PLL output frequency change versus VCO control voltage change, measured in MHz per Volt) of PLL 100 (e.g., trace 230 ) with a dual path configuration with that (e.g., trace 220 ) of a prior art PLL having a single path configuration (e.g., without VCO offset current). The tuning precision is typically frequency-dependent. In the current embodiment, an average improvement of about 12.19% is achieved in the PLL operating range of from about 2800 MHz to about 3100 MHz. In another preferred embodiment, a 50% improvement on PLL tuning precision is obtained.

FIG. 6B compares the output frequency jitter (output frequency uncertainty versus VCO control voltage, measured in MHz per Volt) of PLL 100 (e.g., trace 250 ) with a dual path configuration with that (e.g., trace 240 ) of a prior art PLL having a single path configuration, such as the PLL illustrated with respect to FIG. 2A . The output frequency jitter is typically frequency-dependent. In the current embodiment, an average improvement of about 12.19% is achieved in the PLL operating range of from about 2800 MHz to about 3100 MHz. In another preferred embodiment, a 50% improvement on PLL output frequency jitter is obtained.

As a further advantageous feature, the reduced VCO gain of PLL 100 may lead to a PLL implemented with on a significantly reduced die size. The operating bandwidth of PLL 100 can be approximated as the following:

BW ∝ K VCO * 1 C 2 ( 4 )

where BW is the −3 dB bandwidth of PLL 100 ; K VCO represents the gain of VCO 140 ; C 2 is the capacitance of C 2 in low-pass filter 130 . It is noted from the above equation that, when K VCO is reduced as in preferred embodiments, BW tends to be reduced by a same proportion. To maintain a predetermined PLL 100 bandwidth, C 2 is required to be reduced by a same proportion, resulting in a reduced PLL die size.

As an additional advantageous feature, PLL 100 may be constructed by merging a DVGC unit with a pre-fined-tuned single path PLL. The operational stability of the feedback system of the pre-fine-tuned of a PLL may not be affected. Although an extra zero and pole are created due to the addition of DVGC 150 into an existing PLL system, whereas, the added zero and pole may cancel out each other, leading to a PLL transfer function that is virtually unchanged. As a result, extra circuit elements typically used for frequency compensation may be avoided.

FIG. 7 is a graph illustrating the setting behavior (e.g., trace 270 ) of PLL 100 , where DVGC 150 control signal V mid is plotted versus the tracking and lock-in time. When a phase step is presented at phase detector 110 , V mid is initiated at an upper bound limit of about 2.5V. It takes about 6 μs (microsecond) tracking time for the PLL output frequency signal F out to be locked with an input signal, while V mid is settled at a lower bound limit of about 1.2V.

FIG. 8 illustrates a detailed schematic diagram of VCO 140 ′ implemented in another preferred embodiment of the current invention. The configuration of VCO 140 ′ is similar to VCO 140 illustrated with respect to FIG. 5A , but is different in the following. DVGC 150 ′ in the current embodiment comprises variable Rcv that is implemented between the source of NMOSFET M 2 and ground. DVGC 150 ′ control signal V cont is the difference between low pass filter 130 output signal V mid and ground (i.e., V cont =V mid ). I offset generated from DVGC 150 ′ varies in response to V mid presented on the gate of M 6 , which, in turn, varies in respond to VCO control signal V coin from charge pump 120 . In the current DVGC configuration, control voltage V coin is also referred to as being V mid referenced to ground. Also, V2I 145 ′ is implemented to convert the first VCO control signal V coin into a dc current I v2i , and comprises an NMOSFET M 7 and PMOSFETs M 8 and M 9 . The source of M 7 is connected to ground; the gate of M 7 is coupled to the first VCO control signal V coin . The sources of M 8 and M 9 are connect the supply voltage V DD ; the drain of M 8 is connected to the drain of M 7 ; the gate and the drain of M 8 and the gate of M 9 are tied together; the drain of M 9 is connected to node 155 ; and V2I 145 ′ is configured to produce a continuous and substantially linear current I v2i proportional to V coin within the desired operating range of V coin , between 0 V and 2.5 V, for example. Similarly, the current I offset generated from DVGC 150 ′ is also coupled to node 155 . I v2i and I offset are combined at node 150 to form CCO control current I total . I total is subsequently fed into CCO 160 and converted into PLL output frequency signal F out .

FIG. 9 is a graph illustrating the setting behavior (e.g., trace 280 ) of an embodied PLL comprising VCO 140 ′ illustrated in FIG. 8 . DVGC 150 ′ control signal V mid is plotted versus the tracking and lock-in time. When a phase step is presented at phase detector 110 , V mid is initiated at a lower bound limit of about 0 V. It takes about 7 μs (microsecond) tracking time for the PLL output frequency signal F out to be locked with an input signal, then V mid is settled at an upper bound limit of about 1.5V.

FIG. 10 illustrates a detailed schematic diagram of an enhanced version of a DVGC, namely, DVGC_EN 250 implemented in an additional or alternative embodiment of the present invention. DVGC_EN 250 comprises an active controlled current reference 253 and a current mirror 255 in tandem, each implemented with operational amplifier 260 (OP AMP), among other devices as shown. In the current DVGC_EN configuration, control voltage V cont is referenced to the supply voltage V DD . The configuration of DVGC_EN 250 supports a constant—Gm characteristic of the MOS transistors being substantially independent of variations of supply voltage, manufacturing process, and operational temperature. Thus, DVGC_EN 250 provides an improved power noise rejection capability (e.g., measured as power-supply noise rejection ratio, or PSRR) and more stable output reference current I offset . The implementation of DVGC_EN 250 is compatible with a CMOS process technology.

Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims. As an example, the inventive feature involving a dynamic voltage gain control (DVGC) unit and dual path configuration disclosed in the preferred embodiments may be implemented in applications other than a PLL, where reduced VCO gain and improved VCO tuning precision are desired. As another example, it will be readily understood by those skilled in the art that materials, process steps, process parameters in forming the preferred embodiments may be varied while remaining within the scope of the present invention.

Moreover, the scope of the present application is not intended to be limited to the particular embodiments of the process, machine, manufacture, composition of matter, means, methods and steps described in the specification. As one of ordinary skill in the art will readily appreciate from the disclosure of the present invention, processes, machines, manufacture, compositions of matter, means, methods, or steps, presently existing or later to be developed, that perform substantially the same function or achieve substantially the same result as the corresponding embodiments described herein may be utilized according to the present invention. Accordingly, the appended claims are intended to include within their scope such processes, machines, manufacture, compositions of matter, means, methods, or steps.