Current converter, frequency mixer, radiofrequency transmission system and method for frequency mixing

This application claims the benefit of the priority date of German application DE 10 2006 043 902.3, filed on Sep. 19, 2006, the contents of which are herein incorporated by reference in their entirety.

The invention relates to a current converter. The invention also relates to a frequency mixer having a current converter and to a radiofrequency transmission system. The invention furthermore relates to a method for frequency mixing.

The demands on modern communication standards and on the signal quality of transmission devices are rising with the growing need for high data rates and increasing mobility. Mobile radio standards which have now become customary, such as Universal Mobile Telecommunications System (UMTS), Wideband Code Division Multiple Access (WCDMA), Global System for Mobile Communications (GSM), Enhanced Data Rates for GSM Evolution (EDGE), Wireless Local Area Network (WLAN) or Medium Rate Bluetooth, use bandwidth-efficient modulation types for transmitting high data rates both from a base station to a mobile appliance and from a mobile appliance to a base station. Examples of these modulation types are Quadrature Phase Shift Keying (QPSK), 8-Phase Shift Keying (8-PSK) or Quadrature Amplitude Modulation (QAM). With these types of modulation, what is known as a carrier signal has both its phase and its amplitude modulated in order to transmit the data.

In this context, a popular transmission device comprises a unit for baseband signal processing and a unit for radiofrequency signal processing. In this arrangement, the data for transmission are preprocessed in the baseband unit such that they can be modulated onto the carrier signal and amplified in the radiofrequency unit so as finally to be broadcast via an antenna. To modulate the data which is to be transmitted, what are known as mixers or frequency mixers are used.

FIG. 5 shows an exemplary embodiment of a conventional mixer operating on the basis of the double balanced principle. Such a mixer is also called a Gilbert mixer. The frequency mixer comprises two signal paths comprising transistors T 1 , T 2 whose control connections form a signal input SIN 1 , SIN 2 of the frequency mixer. The two signal paths are coupled to one another via a coupling impedance element IM 1 , which comprises an ohmic resistance, for example. The signal input SIN 1 , SIN 2 can be used to supply a signal which comprises the data for transmission. In addition, a transistor T 7 coupled to the transistor T 1 and a transistor T 8 coupled to the transistor T 2 are provided which respectively form a current source, in the form of a current mirror, together with a transistor T 9 . A reference input IB coupled to the transistor T 9 can be used to supply a reference current.

The frequency mixer also has two transistor pairs T 3 , T 4 and T 5 , T 6 whose control connections are coupled to an oscillator input LO 1 , LO 2 . The transistor pair T 3 , T 4 has a first connection coupled to the transistor T 1 , while the transistor pair T 5 , T 6 has a first connection coupled to the transistor T 2 . The transistors T 3 , T 5 have a second signal connection connected to a first signal output connection SOT 1 . Similarly, second signal connections of the transistors T 5 , T 6 are connected to a second signal output connection SOT 2 . The connections SOT 1 , SOT 2 form a signal output of the frequency mixer.

In the frequency mixer shown, the signal input SIN 1 , SIN 2 is used to supply data for mixing which control a current in the signal paths via the transistors T 1 , T 2 . The oscillator input LO 1 , LO 2 is used to supply an oscillator signal, which is usually a radiofrequency square-wave signal. This alternately turns on the transistors T 3 and T 4 and T 5 and T 6 , which routes the current through the transistors T 1 , T 2 alternately to the signal output connections SOT 1 and SOT 2 . At the signal output SOT 1 , SOT 2 , it is thus possible to tap off a mixed, differential output signal. On the basis of the principle illustrated, the input signal and the oscillator signal are multiplied.

Such mixers are used in a vector modulator, for example. FIG. 6 shows an exemplary embodiment of a conventional vector modulator. This comprises two frequency mixers MIX 1 , MIX 2 which are supplied with data for transmission as vector data I, Q. In this arrangement, the data component I usually has a phase shift of 90° relative to the data component Q. The vector modulator also comprises an oscillator for producing an oscillator signal. This signal is supplied to a frequency divider which halves the frequency of the oscillator signal and at the same time derives two oscillator signals with a 90° shift relative to one another, which are output to the frequency mixers MIX 1 , MIX 2 . The output signals from the mixers MIX 1 , MIX 2 are added and are provided at a common signal output RFOUT.

Frequency mixers may also be used in a polar modulator, as shown in FIG. 7 , for example. In a polar modulator, the data for transmission are described as vectors with an amplitude component R and a phase component φ. In this exemplary embodiment, a signal processor DSP routes the phase component φ to a phase locked loop ΣΔ-PLL with a ΣΔ modulator which controls an oscillator CO. The oscillator CO may be a voltage controlled oscillator (VCO) or a digitally controlled oscillator (DCO). The oscillator produces a carrier signal which comprises the phase information for the phase component φ. The carrier signal is routed as an oscillator signal to a mixer MIX 3 which is in the form of the Gilbert mixer shown in FIG. 5 , for example.

The signal processor DSP also outputs the amplitude component R to an amplitude modulator AM whose output is coupled to the signal input of the mixer MIX 3 . The output of the mixer MIX 3 is coupled to the signal output RFOUT for outputting the mixed radiofrequency signal.

With reference to FIG. 5 , the mixers MIX 1 , MIX 2 , MIX 3 in the exemplary embodiments shown in FIGS. 6 and 7 convert an input voltage at the signal input SIN 1 , SIN 2 into a current via the transistors T 1 , T 2 and the current sources formed by the transistors T 7 , T 8 , T 9 . The transistors T 1 , T 2 are therefore a voltage/current converter. Since the transistors T 1 , T 2 usually have a nonlinear characteristic, the voltage/current conversion produces nonlinear distortions in the input signal, for example in the form of harmonics.

When the converted current is actually mixed with the radiofrequency oscillator signal by means of the transistors T 3 , T 4 , T 5 , T 6 , the harmonics in the current signal may result in intermodulation products in the mixed radiofrequency signal. Particularly the third harmonic can widen the modulation spectrum and disturb the adjacent frequency channels. Hence, the signal quality of the mixed signal can be impaired, which is manifested by an increased bit error rate or an impaired error vector magnitude (EVM), for example.

In the exemplary embodiment shown in FIG. 5 for a frequency mixer, the distortions can be reduced by resistive negative feedback, the provision of a large operating current and lowering of the amplitude at the input of the frequency mixer, for example. However, the resistive negative feedback increases the thermal noise in the frequency mixer, while the gain falls. A larger operating current in the frequency mixer usually increases shot noise in a bipolar transistor or channel noise in a field effect transistor on account of the necessary increase in the size of the transistor channel. Lowering the input amplitude in turn regularly results in impairment of the efficiency of the frequency mixer.

This is problematical particularly because the thermal noise in the frequency mixer can dominate the overall noise in a transmitter.

The following presents a simplified summary in order to provide a basic understanding of one or more aspects of the invention. This summary is not an extensive overview of the invention, and is neither intended to identify key or critical elements of the invention, nor to delineate the scope thereof. Rather, the primary purpose of the summary is to present one or more concepts of the invention in a simplified form as a prelude to the more detailed description that is presented later.

In one exemplary embodiment, a current converter comprises a differential signal input, a differential current output and a first and a second differential amplifier. The first differential amplifier has a first current path comprising a first transistor and a second current path comprising a second transistor. In the case of the first and second transistors, control connections form a differential input of the first differential amplifier. The first and second transistors have respective first connections, which are coupled to the differential current output, and respective second connections, which are coupled to one another via a coupling impedance element.

In one embodiment the second differential amplifier has an output which is coupled to the differential input of the first differential amplifier. In addition, the second differential amplifier has a first and a second input which are coupled to the differential signal input of the current converter. Furthermore, the first input is coupled to the second connection of the second transistor via a first impedance element and the second input is coupled to the second connection of the first transistor via a second impedance element. The first and second impedance elements advantageously have at least one ohmic component.

In one embodiment, current conversion is effected by actuating the first and second transistors with the output signals from the second differential amplifier. Resistance-based or impedance-based feedback of signals on the first and second transistors to the inputs of the second differential amplifier linearizes the current conversion, that is to say that a differential output current at the differential current output of the current converter is linearly dependent on a differential input signal at the differential signal input. Harmonics in the differential output current are therefore reduced.

In other embodiments, the first differential amplifier may comprise a current source which is coupled to the transistors and which is formed by a current mirror, for example. In addition, the second differential amplifier may be formed by an operational amplifier. The differential input signal may be supplied as a differential voltage via impedance elements at the first and second inputs of the second differential amplifier. In this arrangement, the impedance elements may be switchable and designed to have an adjustable impedance value.

One illustrative embodiment of a frequency mixer comprises not only a current converter in accordance with one of the embodiments described, but also a mixer cell which has a differential current input which is coupled to the differential current output of the current converter. In addition, the mixer cell has a differential oscillator input for supplying an oscillator signal and a differential signal output for outputting a mixed output signal. The mixer cell may be in the form of a double balanced mixer, in one embodiment.

The use of a current converter with reduced harmonics in the differential output current in one embodiment of the frequency mixer allows frequency mixing to be achieved in which intermodulation products in the mixed radiofrequency output signal are reduced in comparison with a conventional frequency mixer.

In another embodiment of the frequency mixer, a second or further mixer cell may be provided which has its current input connected to the differential current output of the current converter. By way of example, the further mixer cell is designed for a different frequency band than the first mixer cell. In this case, the mixer cells are supplied with oscillator signals at different oscillator frequencies. The use of a common current converter for a plurality of mixer cells allows the space requirement on the chip to be reduced when integrating the frequency mixer in an integrated circuit.

In another embodiment of the invention, a frequency mixer comprises a first amplification device which outputs a differential current depending on a differential control signal and derives a differential feedback signal from the differential current. In addition, a second amplification device is provided which amplifies a differential input signal at its input depending on the differential feedback signal to form the differential control signal. The frequency mixer also comprises a mixer cell which produces a differential output signal by multiplying the differential current and an oscillator signal.

In one embodiment, producing the feedback signal in the first amplification device and supplying the feedback signal to the second amplification device allows linearization of a differential current in the first amplification device with respect to the differential input signal. This in turn results in reduced intermodulation products in the differential output signal from the mixer cell.

In another embodiment, a differential input signal is amplified depending on a differential feedback signal to form a differential control signal. A differential current is controlled depending on the differential control signal and a differential feedback signal is derived from the differential current. The differential current is mixed with an oscillator signal. The mixing can be done by means of double balanced mixing, in one embodiment.

To the accomplishment of the foregoing and related ends, the invention comprises the features hereinafter fully described and particularly pointed out in the claims. The following description and the annexed drawings set forth in detail certain illustrative aspects and implementations of the invention. These are indicative, however, of but a few of the various ways in which the principles of the invention may be employed. Other objects, advantages and novel features of the invention will become apparent from the following detailed description of the invention when considered in conjunction with the drawings.

The invention is explained in more detail below using a plurality of exemplary embodiments with reference to the drawings in which:

FIG. 1 shows one embodiment of a current converter according to the invention,

FIG. 2 shows one embodiment of a frequency mixer according to the invention,

FIG. 3 shows one embodiment of a switchable resistance according to the invention,

FIG. 4 shows another embodiment of a frequency mixer according to the invention,

FIG. 5 shows a conventional frequency mixer,

FIG. 6 shows a vector modulator, and

FIG. 7 shows a polar modulator.

In the following description further aspects and embodiments of the present invention are disclosed. In addition, reference is made to the accompanying drawings, which form a part hereof, and in which is shown by way of illustration, in which the invention may be practiced. The embodiments of the drawings present a disclosure in order to provide a better understanding of one or more aspects of the present invention. This disclosure is not intended to limit the features or key-elements of the invention to a specific embodiment. Rather, the different elements, aspects and features disclosed in the embodiments can be combined in different ways by a person skilled in the art to achieve one or more advantages of the present invention. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. The elements of the drawing are not necessarily to scale relative to each other. Like reference numerals designate corresponding similar parts.

FIG. 1 shows one embodiment of a current converter 10 . The current converter 10 comprises a first differential amplifier DIF 1 having a first transistor T 1 , a second transistor T 2 and a current source IS. The first transistor T 1 is arranged in a current path CP 1 between a reference potential connection VSS and a first connection IOT 1 of a differential current output IOT 1 , IOT 2 which also comprises the current source IS. Similarly, the transistor T 2 is arranged in a second current path CP 2 between a second connection IOT 2 of the differential current output IOT 1 , IOT 2 and the reference potential connection VSS. In this arrangement, first connections 11 , 21 of the first and second transistors T 1 , T 2 are connected to the differential current output IOT 1 , IOT 2 . Second connections 12 , 22 of the transistors T 1 , T 2 are coupled to one another via coupling impedance elements IM 1 , which usually have the same impedance value. A connecting node of the coupling impedance elements IM 1 is a mid-potential connection to which the current source IS is connected in this case.

The current converter 10 also comprises a second differential amplifier DIF 2 , which has a differential signal input with a first input connection IN+ and a second input connection IN−. The first input IN+ is coupled to the second connection 22 of the second transistor T 2 via a first impedance element Z 1 and to a connection SIN 1 via a further impedance element Z 3 . The second connection IN− of the second differential amplifier DIF 2 is connected to the second connection 12 of the transistor T 1 via a second impedance element Z 2 . In addition, the connection IN− is coupled to a second input connection SIN 2 via a further impedance element Z 4 . The connections SIN 1 , SIN 2 form a differential signal input of the current converter 10 .

Control connections 13 , 23 of the first and second transistors T 1 , T 2 form a differential input of the first differential amplifier DIF 1 and are coupled to output connections OT+, OT− of the second differential amplifier DIF 2 .

In the embodiment of FIG. 1 , the second differential amplifier DIF 2 is formed by an operational amplifier. In this embodiment the differential amplifiers DIF 1 , DIF 2 each perform the function of an amplification device. Each of the first and the second differential amplifiers DIF 1 , DIF 2 comprises a separate current source which are independent of each other. Therefore the differential amplifiers DIF 1 , DIF 2 can be operated independently.

In one embodiment, the first and second transistors T 1 , T 2 are in the form of n-channel field effect transistors. In an alternative embodiment, the transistors T 1 , T 2 may also be in the form of bipolar transistors.

The differential signal input SIN 1 , SIN 2 is used to supply a differential input signal, which may be a voltage signal, to the inputs IN+, IN− of the second differential amplifier DIF 2 . The differential input signal is amplified to form a differential control signal, which is output at the outputs OT+, OT− of the second differential amplifier DIF 2 . The control signal for the second differential amplifier DIF 2 controls a differential current in the first and second current paths CP 1 , CP 2 via the transistors T 1 , T 2 .

The impedance elements Z 1 , Z 2 take the respective currents in the first and second current paths CP 1 , CP 2 and derive a feedback signal which is routed to the inputs IN+, IN− of the second differential amplifier DIF 2 . This involves negative feedback, since the current in the first transistor T 1 , which is actuated via the first connection OT+, which is a noninverting output of the second differential amplifier DIF 2 , is fed back to the second input IN−, which is an inverting input of the second differential amplifier DIF 2 . Similarly, the current in the second transistor T 2 , which is actuated via the second output OT−, which is an inverting output, is fed back to the noninverting input IN+ of the second differential amplifier DIF 2 . By way of example, the impedance elements Z 1 , Z 2 , Z 3 , Z 4 may be in the form of complex resistances with a purely or predominantly ohmic component.

The feedback actuation of the transistors T 1 , T 2 can improve the linearity response of the transistors T 1 , T 2 . This is because the transistors T 1 , T 2 are actuated such that a current flowing through them is linearly dependent on the differential input signal by virtue of a ratio of the impedance values of the impedance elements Z 1 , Z 2 , Z 3 , Z 4 . In other words, the feedback makes it possible to ensure a linear dependency for a differential output current at the differential current output IOT 1 , IOT 2 on a differential input signal at the signal input SIN 1 , SIN 2 . Using the proposed principle, it is possible to reduce not only odd-numbered harmonics but also even-numbered harmonics in the output current.

By virtue of a suitable choice of impedance values for the impedance elements Z 1 , Z 2 , Z 3 , Z 4 , it is possible to set a gain factor for amplifying the differential input signal into the differential output current. If the differential input signal is supposed to be supplied as an impressed current signal, the input impedance elements Z 3 , Z 4 may even be omitted. In this embodiment, the current converter 10 is a linearized current/current converter.

In one embodiment, the impedance elements Z 1 , Z 2 , Z 3 , Z 4 may also be in the form of complex resistances with a capacitive or inductive component. In this embodiment, the impedance elements Z 1 , Z 2 , Z 3 , Z 4 have filter properties for filtering the differential input signal or the differential current, for example in line with low pass filtering.

The differential output current from the current converter 10 is also influenced by the coupling impedance elements IM 1 . In one embodiment, the coupling impedance elements IM 1 may be formed by ohmic resistances. In this case, there is no frequency-based influencing of the differential output current. In another embodiment, the coupling impedance elements IM 1 may have an inductive or capacitive component. Hence, with a suitable choice of impedance value for the coupling impedance elements IM 1 , it is possible to achieve filter properties for the coupling impedance elements IM 1 and for the differential current to be filtered, for example in line with low pass filtering.

FIG. 2 shows an embodiment of a frequency mixer having a current converter 10 and a mixer cell 20 . The design and operation of the current converter 10 essentially correspond to those of the embodiment shown in FIG. 1 . The second connections 12 , 22 of the first and second transistors T 1 , T 2 are coupled via a coupling impedance element IM 1 , which may again have resistive, capacitive or inductive properties.

In addition, two current sources IS 1 , IS 2 are provided. The first current source IS 1 , which is in the form of a current mirror comprising a transistor T 7 and a transistor T 9 , is connected to the second connection 12 of the first transistor T 1 . Accordingly, the second current source IS 2 , which is in the form of a current mirror comprising the transistor T 9 and a transistor T 8 , is connected to the second connection 22 of the second transistor T 2 . A reference current is supplied to the transistor T 9 via a reference connection IB.

The mixer cell 20 comprises a third, a fourth, a fifth and a sixth transistor T 3 , T 4 , T 5 , T 6 having a control connection 33 , 43 , 53 , 63 and a first signal connection 31 , 41 , 51 , 61 and also a second signal connection 32 , 42 , 52 , 62 , respectively. In this case, the control connections 33 , 63 of the third and sixth transistors T 3 , T 6 are coupled to a first connection LO 1 of an oscillator input LO 1 , LO 2 , and control connections 43 , 53 of the fourth and fifth transistors T 4 , T 5 are coupled to a second connection LO 2 of the oscillator input LO 1 , LO 2 .

The mixer cell 20 has a differential current input INT 1 , INT 2 which is coupled to the differential current output IOT 1 , IOT 2 of the current converter 10 . In addition, the mixer cell 20 comprises a differential oscillator input LO 1 , LO 2 for supplying an oscillator signal LO and a differential signal output SOT 1 , SOT 2 for outputting a mixed output signal.

The first signal connections 31 , 41 of the third and fourth transistors T 3 , T 4 are coupled to a first connection INT 1 and the first signal connections 51 , 61 of the fifth and sixth transistors T 5 , T 6 are coupled to a second connection INT 2 of the current input INT 1 , INT 2 . The second signal connections 32 , 52 of the third and fifth transistors T 3 , T 5 are connected to a first connection SOT 1 and the second signal connections 42 , 62 of the fourth and sixth transistors T 4 , T 6 are connected to a second connection SOT 2 of the signal output SOT 1 , SOT 2 .

As described previously for FIG. 1 , the current converter 10 performs a linear conversion of a differential input voltage at the signal input SIN 1 , SIN 2 into a differential output current at the current output IOT 1 , IOT 2 , which in this embodiment simultaneously forms the differential current input INT 1 , INT 2 of the mixer cell 20 . In this embodiment, the impedance elements Z 1 , Z 2 , Z 3 , Z 4 and the coupling impedance element IM 1 are in the form of switchable or otherwise variable resistances.

FIG. 3 shows one embodiment of a switchable resistance which can be inserted into the current converter 10 in FIG. 2 . In this case, the resistance Z 1 comprises a plurality of parallel-connected series circuits which each comprise a resistor R 1 , R 12 , R 1 N and an associated switching transistor TS 1 , TS 2 , TSN. The transistors TS 1 , TS 2 , TSN are used as switches in one embodiment in order to activate or deactivate the resistors R 11 , R 12 , R 1 N. This can be done by means of appropriate control signals on the control inputs CT 1 , CT 2 , CTN of the transistors TS 1 , TS 2 , TSN. The resistance values of the resistors R 11 , R 12 , R 1 N may be the same or may have a binary or other type weighting relative to one another. Accordingly, a resistance value for the switchable resistance Z 1 can be set by means of appropriate control signals.

With reference to FIG. 2 , by setting the resistance values for the impedance elements Z 1 , Z 2 , Z 3 , Z 4 and for the coupling impedance element IM 1 it is possible to set the gain factor of the current converter 10 . In other words, the amplification in the current converter 10 is produced using an adjustable gain factor in one embodiment. In this case, the amplification does not have a negative influence on the linearity of the current converter 10 on account of the unchanged linear properties of the impedance elements Z 1 , Z 2 , Z 3 , Z 4 and of the coupling impedance element IM 1 .

The differential output current from the current converter 10 is supplied to the mixer cell 20 comprising the transistors T 3 , T 4 , T 5 , T 6 , which are also referred to as a mixer quartet. The transistors T 3 , T 4 , T 5 , T 6 are actuated by means of the oscillator signal, which is usually in square-wave form, which multiplies the differential current and the oscillator signal. In this context, it is again also possible to refer to double balanced mixing. Since the harmonics are suppressed in the differential current by the current converter, intermodulation products in the mixed output signal at the signal output SOT 1 , SOT 2 will end up being small. Since the controlled differential current in the current converter 10 is supplied directly to the mixer quartet T 3 , T 4 , T 5 , T 6 in the mixer cell 20 , the power consumption is increased only insignificantly in comparison with the conventional frequency mixer, depending on the currents in the second differential amplifier DIF 2 .

In one embodiment the frequency mixer may be used in a radiofrequency transmission system. By way of example, the mixer is used in a transmission/reception device for a WLAN system. This involves the application of Orthogonal Frequency Division Multiplexing (OFDM) modulation, for example 64-QAM-OFDM modulation. The modulator used may be a vector modulator, as is shown in FIG. 6 , for example. In contrast to a vector modulator with a conventional mixer, in the case of a vector modulator with a linearized mixer based on one of the embodiments described, the power consumption of the transmitter may be halved or the output power may be increased.

A frequency mixer according to one of the embodiments described may also be used in a polar modulator, as is shown in FIG. 7 , for example. In this case too, the use of a linearized frequency mixer can reduce the power consumption of the modulator arrangement in comparison with conventional embodiments. Besides WLAN systems, the proposed embodiments can also be applied to frequency mixers for other transmission systems and standards, such as UMTS or GSM/EDGE.

FIG. 4 shows another embodiment of a frequency mixer. As an extension to the embodiment shown in FIG. 2 , the frequency mixer comprises a further, second mixer cell 20 a , which has a design generally corresponding to that of the first mixer cell 20 . The further mixer cell 20 a thus comprises a further mixer quartet with the transistors T 3 a , T 4 a , T 5 a , T 6 a , whose control connections are coupled to a further differential oscillator input LO 1 a , LO 2 a . In addition, the further mixer cell 20 a has a further differential current input INT 1 a , INT 2 a , which is coupled to the differential current output IOT 1 , IOT 2 of the current converter 10 . A mixed output signal can be output at a further differential signal output SOT 1 a , SOT 2 a.

In one embodiment, the first mixer cell 20 and the further mixer cell 20 a are designed for different frequency bands, that is to say for different frequencies of an oscillator signal. By way of example, in one embodiment the first mixer cell 20 is designed for a frequency of approximately 2.4 GHz, while the further mixer cell 20 a is designed for a frequency of approximately 5 GHz. However, the production of an input signal for modulation which is supplied to the current converter 10 may be identical for both resultant frequency ranges. The input signal can accordingly be converted by means of the current converter 10 into a differential current which, by way of example, is supplied either to the first mixer cell 20 or to the further mixer cell 20 a , or both, in order to obtain an output signal mixed with the respective oscillator signal.

In one embodiment the first mixer cell 20 is supplied with an oscillator signal at a first oscillator frequency. To deactivate the second mixer cell 20 a , it is possible to apply to the further oscillator input LO 1 a , LO 2 a , for example, a signal which puts the transistors T 3 a , T 4 a , T 5 a , T 6 a into an off state, for example a ground signal. This prevents a flow of current through the transistors T 3 a , T 4 a , T 5 a , T 6 a in the further mixer cell 20 a , and mixing is performed only by means of the first mixer cell 20 .

Accordingly, for modulation with an oscillator signal at a second oscillator frequency, this oscillator signal can be supplied to the further oscillator input LO 1 a , LO 2 a , while the oscillator input LO 1 , LO 2 of the first mixer cell 20 is supplied with a ground signal. In this case, modulation is effected only by means of the further mixer cell 20 a . A flow of current through the first mixer cell 20 is prevented by the transistors T 3 , T 4 , T 5 , T 6 , which are off.

With the embodiment shown in FIG. 4 , it is possible to save the space for a further current converter when the frequency mixer is integrated in an integrated circuit, resulting in a reduction in manufacturing costs. The proposed principle is particularly suitable for using radio systems using linear modulation methods.

Even though use in a transmission device is assumed in the embodiments shown, with the frequency of the oscillator signal being higher than a signal frequency of the differential input signal, the proposed embodiments may also be used in a down-conversion mixer in a reception system. In the case of a down-conversion mixer, the frequency of the oscillator signal is usually less than or equal to the frequency of the input signal.

Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art, that any arrangement which is calculated to achieve the same purpose may be substituted for the specific embodiments shown. It is to be understood, that the above description is intended to be illustrative and not restrictive. This application is intended to cover any adaptations or variations of the invention. Combinations of the above embodiments and many other embodiments will be apparent to those of skill in the art upon reading and understanding the above description. The scope of the invention includes any other embodiments and applications in which the above structures and methods may be used. The scope of the invention should, therefore, be determined with reference to the appended claims along with the scope of equivalents to which such claims are entitled.

It is emphasized that the Abstract is provided to comply with 37 C.F.R. section 1.72( b ) requiring an abstract that will allow the reader to quickly ascertain the nature and gist of the technical disclosure. It is submitted with the understanding, that it will not be used to interpret or limit the scope or meaning of the claims.