A bidirectional DC-DC converter 10 has smoothing capacitors Cs 1 and Cs 2 , a smoothing reactor Ls, resonant reactors Lr 1 and Lr 2 , and a resonant capacitor Cr, as well as insulated gate bipolar transistors (IGBTs) Q 1 , Q 2 , and Q 3 to which buffer capacitors C 1 , C 2 , and C 3 are connected in parallel and diodes D 1 , D 2 , and D 3 are connected in a back-to-back configuration. After part of the energy stored in the smoothing reactor Ls and/or resonant reactors Lr 1 and Lr 2 draws charges in the buffer capacitor C 3 and is stored in the resonant capacitor Cr, the energy is stored in the resonant reactors Lr 1 and Lr 2 . The energy is used to draw charges in the buffer capacitors C 1 and C 2 , achieving soft switching of IGBTs Q 1 , Q 2 , and Q 3.

Description

16 parts

›CLAIM OF PRIORITY

The present application claims priority from Japanese application serial no. 2006-111842, filed on Apr. 14, 2006, the content of which is hereby incorporated by reference into this application.

›FIELD OF THE INVENTION

The present invention relates to a bidirectional DC-DC converter having a soft switching function and to a power supply apparatus with the bidirectional DC-DC converter.

›BACKGROUND OF THE INVENTION

Hybrid vehicles with high efficiency have recently become widespread in response to a growing concern about global environmental conservation. The efficiency of a hybrid vehicle is increased by storing energy in an electricity storage unit and using the stored energy as necessary. A bidirectional DC-DC converter is used to control the charging and discharging of the electricity storage unit.

In conventional bidirectional DC-DC converters, hard switching is dominant in which a switchover is made while a voltage is applied to a switching device or current is flowing in it. The action by which a switching device is turned on from the turned-off state or turned off from the turned-on state is called switching. The efficiency of hard switching is low because the energy loss involved in switching is large.

A resonant bidirectional DC-DC converter that reduces the loss involved in switching to increase the efficiency is disclosed in Japanese Patent Laid-open No. 2003-33013 (Patent Document 1). This resonant bidirectional DC-DC converter performs boost and buck operations between 12-V and 42-V power supplies, so a pair of main metal oxide semiconductor field effect transistors (MOSFETs) and a main smoothing reactor are used as a basic main circuit. In addition to this basic main circuit, a pair of auxiliary MOSFETs and a resonant reactor are used as an auxiliary resonant circuit so as to implement soft switching. The pair of main MOSFETs are connected in series between both buses of a second power supply. An auxiliary smoothing reactor is connected between the series connection points and the positive bus of a first power supply. A snubber (buffer) capacitor is connected in parallel to each of the pair of main MOSFETs. The auxiliary resonant circuit is connected between both ends of the main reactor; the auxiliary resonant circuit comprises a pair of anti-series auxiliary MOSFETs and a series auxiliary (resonant) reactor.

The auxiliary resonant circuit controls the switching of the auxiliary MOSFET pair in both the boost and buck operations so that the charging and discharging of the snubber capacitors connected to the main MOSFETs are controlled by resonant current flowing in the resonant reactor, thereby achieving soft switching.

A resonant bidirectional DC-DC converter is disclosed in Japanese Patent Laid-open No. 2005-176540 (Patent Document 2), in which four main transistors are connected in a bridge form, a main rector is connected among their bridging points, and an auxiliary resonant circuit is connected in parallel to the main reactor. This auxiliary resonant circuit also comprises a pair of anti-series auxiliary transistors and a series auxiliary (resonant) reactor as in (Patent Document 1).

In Japanese Patent Laid-open No. 2005-184965 (Patent Document 3), a bidirectional DC-DC converter using many devices is disclosed.

Japanese Patent Laid-open No. 2004-343923 (Patent Documents 4), 2001-37214 (Patent Documents 5), 2003-153527 (Patent Documents 6), 2003-189602 (Patent Documents 7), and 2004-201373 (Patent Documents 8) disclose technologies for increasing the efficiency of a unidirectional DC-DC converter by reducing a switching loss.

›SUMMARY OF THE INVENTION

The conventional resonant bidirectional DC-DC converters disclosed in Patent Documents 1 and 2 use a reactor with a large current rating as the main (smoothing) reactor, resulting in an increase in size and weight. This is because the main reactor with a large current rating is needed to allow current to flow into the auxiliary reactor in the direction opposite to the direction in which current flows in the main reactor so that charges stored in the snubber capacitors connected to main switching devices can be drawn.

A pair of auxiliary switching devices are also needed to achieve soft switching of the main switching devices. An increase in the number of switching devices makes the structure complex and raises the cost.

Just complementary switching is insufficient in control of the main switching devices and auxiliary switching devices; timing is also important, resulting in complexity in control. From the viewpoint of this, the complex structure and increased cost cannot be avoided.

In addition, the auxiliary switching devices are turned on based on zero-current switching (ZCS), generating a switching loss larger than in zero-voltage switching (ZVS).

The technology disclosed in Patent Document 3 also uses many devices, and an increase in the number of devices makes the structure complex and raises the cost.

As described above, the conventional bidirectional DC-DC converters are problematic in that they have many factors that sacrifice their sizes, weights, and costs in order to achieve soft switching of the main switching devices.

An object of the present invention is to provide a bidirectional DC-DC converter that uses a reactor with a relative small current rating as the smoothing reactor to achieve soft switching.

Another object of the present invention is to provide a bidirectional DC-DC converter that achieves soft switching with a relatively small number of switching devices.

A yet another object of the present invention is to provide a bidirectional DC-DC converter that achieves soft switching by performing relatively simple switching control for the switching devices.

A still another object of the present invention is to provide a bidirectional DC-DC converter that achieves zero-voltage switching (ZVS) even when the auxiliary switching device is turned on and off.

In a bidirectional DC-DC converter having a first smoothing capacitor connected in parallel to a first power supply, a second smoothing capacitor connected in parallel to a second power supply, first and second switching devices, a smoothing reactor, and a control means for repeatedly storing energy in and releasing energy from the smoothing reactor and turning on and off the first and second switching devices so that electric power is transferred and received between the first and second smoothing capacitors, the bidirectional DC-DC converter in one aspect of the present invention comprises a resonant reactor, a storage circuit for storing part of the energy released from the smoothing reactor and/or resonant reactor in a period in which the smoothing reactor releases the energy it stored, and a third switching device by which part of the energy stored in the storage circuit is released into the resonant reactor; the control means uses the energy stored in the resonant reactor to turn on and off the first to third switching devices so that charges stored in buffer capacitors for the first and second switching devices are drawn immediately before the first and second switching devices are turned on.

In a preferred mode of the present invention, the resonant reactor stores energy by using current for storing energy in and/or releasing energy from the smoothing reactor; the energy stored in the resonant reactor and/or the energy of the smoothing reactor is transferred to a capacitor in the storage circuit; the energy stored in the capacitor is stored in the resonant reactor through the third switching device; the energy stored in the resonant reactor is used to draw charges stored in the snubber (buffer) capacitors (or output capacitors) of the first and second switching devices.

According to a preferred mode of the present invention, current flowing in the smoothing reactor may be nearly equal to current in the case in which soft switching is not used, so a bidirectional DC-DC converter can be provided that uses a reactor with a relative small current rating as the smoothing reactor to achieve soft switching.

According to another preferred mode of the present invention, a bidirectional DC-DC converter can be provided that achieve soft switching with a relatively small number of switching devices.

According to yet another preferred mode of the present invention, a bidirectional DC-DC converter can be provided that achieves soft switching by relatively simple switching control of main (first and second) switching devices and an auxiliary (third) switching device.

According to still another preferred mode of the present invention, a bidirectional DC-DC converter can be provided that can achieve zero-voltage switching (ZVS) even when the auxiliary (third) switching device is turned on and off.

Other objects and features of the present invention will be clarified in embodiments given below.

›BRIEF DESCRIPTION OF THE DRAWINGS · 1 of 2

FIG. 1 is the circuit configuration of a bidirectional DC-DC converter according to a first embodiment of the present invention.

FIGS. 2A to 2F are circuit diagrams, each of which illustrates a boost operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIGS. 3G to 3L are also circuit diagrams, each of which illustrates a boost operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIG. 4 shows voltage and current waveforms for illustrating the boost operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIGS. 5A to 5F are circuit diagrams, each of which illustrates a buck operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIGS. 6G to 6L are also circuit diagrams, each of which illustrates a buck operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIG. 7 shows voltage and current waveforms for illustrating the buck operation of the bidirectional DC-DC converter in the first embodiment of the present invention.

FIG. 8 shows a main circuit diagram of a bidirectional DC-DC converter according to a second embodiment of the present invention.

FIG. 9 shows a main circuit diagram of a bidirectional DC-DC converter according to a third embodiment of the present invention.

FIG. 10 shows a main circuit diagram of a bidirectional DC-DC converter according to a fourth embodiment of the present invention.

FIGS. 11A to 11F are circuit diagrams, each of which illustrates a boost operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention.

FIGS. 12G to 12L are also circuit diagrams, each of which illustrates a boost operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention.

FIGS. 13A to 13F are circuit diagrams, each of which illustrates a buck operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention.

FIGS. 14G to 14L are also circuit diagrams, each of which illustrates a buck operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention.

FIG. 15 shows a main circuit diagram of a bidirectional DC-DC converter according to a fifth embodiment of the present invention.

FIG. 16 shows a main circuit diagram of a bidirectional DC-DC converter according to a sixth embodiment of the present invention.

FIG. 17 shows a main circuit diagram of a bidirectional DC-DC converter according to a seventh embodiment of the present invention.

FIG. 18 shows a main circuit diagram of a bidirectional DC-DC converter according to an eighth embodiment of the present invention.

FIG. 19 shows a main circuit diagram of a bidirectional DC-DC converter according to a ninth embodiment of the present invention.

FIG. 20 shows a main circuit diagram of a bidirectional DC-DC converter according to a tenth embodiment of the present invention.

FIG. 21 shows a main circuit diagram of a bidirectional DC-DC converter according to an eleventh embodiment of the present invention.

FIG. 22 shows a main circuit diagram of a bidirectional DC-DC converter according to a twelfth embodiment of the present invention.

FIGS. 23A to 23F are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 .

FIGS. 24G to 24L are also circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 .

FIGS. 25A to 25F are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 .

FIGS. 26G to 26L are also circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 .

FIG. 27 shows a main circuit diagram of a bidirectional DC-DC converter according to a thirteenth embodiment of the present invention.

FIG. 28 shows a main circuit diagram of a bidirectional DC-DC converter according to a fourteenth embodiment of the present invention.

FIG. 29 shows a main circuit diagram of a bidirectional DC-DC converter according to a fifteenth embodiment of the present invention.

FIGS. 30A to 30F are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 .

FIGS. 31G to 31L are also circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 .

FIGS. 32A to 32F are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 .

FIGS. 33G to 33L are also circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 .

FIG. 34 shows a main circuit diagram of a bidirectional DC-DC converter according to a sixteenth embodiment of the present invention.

FIG. 35 shows a main circuit diagram of a bidirectional DC-DC converter according to a seventeenth embodiment of the present invention.

FIG. 36 shows a main circuit diagram of a bidirectional DC-DC converter according to an eighteenth embodiment of the present invention.

FIG. 37 shows a main circuit diagram of a bidirectional DC-DC converter according to a nineteenth embodiment of the present invention.

›BRIEF DESCRIPTION OF THE DRAWINGS · 2 of 2

FIG. 38 shows a main circuit diagram of a bidirectional DC-DC converter according to a twentieth embodiment of the present invention.

FIG. 39 shows a main circuit diagram of a bidirectional DC-DC converter according to a twenty-first embodiment of the present invention.

FIG. 40 shows the general structure of a power supply apparatus in a hybrid vehicle, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus.

FIG. 41 shows the general structure of a power supply apparatus in a vehicle, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus.

FIG. 47 shows the general structure of an uninterruptible power supply in which a bidirectional DC-DC converter according to the present invention is used.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 1 of 10

Embodiments of the present invention will be described in detail with reference to the drawings. The same or equivalent elements in the drawings are assigned the same reference numerals to eliminate duplicate description. In the description below, an insulated gate bipolar transistor (IGBT) is used as an exemplary switching device, and a voltage nearly equal to or below the forward drop voltage of a switching device or diode in the on state will be referred to as the zero voltage.

First Embodiment

FIG. 1 is the circuit configuration of a bidirectional DC-DC converter according to a first embodiment of the present invention. The bidirectional DC-DC converter 10 has first and second switching devices (IGBTs) Q 1 and Q 2 as well as a third switching device (IGBT) Q 3 ; the switching of these switching devices is controlled by a control means 2 . The control means 2 receives outputs of a current sensor 3 and voltage sensors 4 and 5 . Other circuit elements are smoothing capacitors Cs 1 and Cs 2 , a smoothing reactor Ls, resonant reactors Lr 1 and Lr 2 , a resonant capacitor Cr, diodes D 1 to D 3 , and buffer capacitors C 1 to C 3 .

The bidirectional DC-DC converter 10 is connected between a power supply V 1 and another power supply V 2 , the voltage of which is higher than the voltage of power supply V 1 , and performs voltage conversion between power supply V 1 and power supply V 2 . A load R 1 is connected to power supply V 1 , and another load R 2 is connected to power supply V 2 .

Smoothing capacitor Cs 1 is connected between a negative bus LN 0 and a positive bus LN 1 of power supply V 1 , and resonant capacitor Cs 2 is connected between negative bus LN 0 and a positive bus LN 2 of power supply V 2 . The emitter of IGBT Q 1 is connected to negative bus LN 0 , and the collector of IGBT Q 2 is connected to positive bus LN 2 . One end of the smoothing reactor Ls is connected to positive bus LN 1 , and the other end of the smoothing reactor Ls is connected to a node N 0 .

Resonant reactor Lr 1 is connected between the collector of IGBT Q 1 and node N 0 , and resonant reactor Lr 2 is connected between the emitter of IGBT Q 2 and node N 0 . These resonant reactors Lr 1 and Lr 2 are magnetically coupled to use the same core, thereby increasing the usage efficiency of the core and making the core compact. The point at which IGBT Q 1 and resonant reactor Lr 1 are interconnected is called a node N 1 , and the point at which IGBT Q 2 and resonant reactor Lr 2 are interconnected is called a node N 2 .

IBGT Q 3 is connected in series to the resonant capacitor Cr to form an active resonant circuit. Another active resonant circuit is formed between node N 1 and node N 2 in such a way that the emitter of IGBT Q 3 faces node N 1 .

Diodes D 1 to D 3 are connected between the collectors and emitters of IGBTs Q 1 to Q 3 , respectively, so that current flows from the emitter to the collector. Buffer capacitors C 1 to C 3 are connected between the collectors and emitters of IGBTs Q 1 to Q 3 , respectively.

Symbols representing voltages and currents in the circuit diagram in FIG. 1 will be defined before operation is described in detail. Voltages VQ 1 to VQ 3 between the collectors and emitters of IGBTs Q 1 to Q 3 are positive on the collector side. Composite currents flowing in IGBTs Q 1 to Q 3 and in diodes D 1 to D 3 respectively connected in parallel to IGBTs Q 1 to Q 3 are positive when they flow from the collectors to the emitters in IGBTs Q 1 to Q 3 ; the composite currents are denoted IQD 1 to IQD 3 .

The direction of current ILs flowing into the smoothing reactor Ls is defined as follows: in a boost operation in which energy is sent from power supply V 1 to power supply V 2 , the current is positive when it flows from positive bus LN 1 to node N 0 ; in a buck operation in which energy is sent from power supply V 2 to power supply V 1 , the current is positive when it flows from node N 0 to positive bus LN 1 . Current ILr 1 flowing into resonant reactor Lr 1 is positive when it flows from node N 1 to node N 0 . Current ILr 2 flowing into resonant reactor Lr 2 is positive when it flows from node N 0 to node N 2 .

Current ICr flowing into the resonant capacitor Cr is positive when it flows in the direction in which forward current flows in diode D 3 .

(From V 1 to V 2 : Boost Operation)

FIGS. 2A to 2F and FIGS. 3G to 3L are circuit diagrams, each of which illustrates a boost operation in the first embodiment of the present invention. FIG. 4 shows voltage and current waveforms for illustrating the boost operation in the first embodiment of the present invention.

The boost operation in the first embodiment of the present invention will be described in detail with reference to FIGS. 2A to 2F , FIGS. 3G to 3L , and FIG. 4 . In the description below, the boost operation is defined as the operation to transfer energy in power supply V 1 to power supply V 2 . FIGS. 2A to 2F and FIGS. 3G to 3L correspond to modes A to L in FIG. 4 .

(Mode A)

In mode A, IGBT Q 1 is in the on state, and IGBTs Q 2 and Q 3 are in the off state. The current ILs in the smoothing reactor Ls gradually increases, the value of current ILr 1 of resonant reactor Lr 1 gradually increases in the negative direction, and the energy in power supply V 1 is stored in the smoothing reactor Ls and resonant reactor Lr 1 . Buffer capacitors C 2 and C 3 and the resonant capacitor Cr are charged with the polarity shown in FIG. 2A at this time.

(Mode B)

IBGT Q 1 is then turned off at time t 1 . Voltage VQ 1 at buffer capacitor C 1 connected in parallel to IGBT Q 1 gradually increases from zero at this time. Therefore, IGBT Q 1 undergoes zero-voltage switching at time t 1 . As voltage VQ 1 increases, the voltage between the collector of IGBT Q 2 and the collector of IGBT Q 1 decreases. However, currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time. Buffer capacitors C 2 and C 3 release their stored charges, causing voltages VQ 2 and VQ 3 to drop. When VQ 3 reaches a voltage of 0 V, diode D 3 is turned on. FIG. 2B shows the state of the circuit at that time. Specifically, the current ILs passing through the smoothing reactor Ls is divided into two paths; after passing through resonant reactor Lr 1 , the current is divided into current that flows into buffer capacitor C 1 and current that flows into buffer capacitor C 3 (diode D 3 when voltage VQ 3 is 0 V), resonant capacitor Cr, and then buffer capacitor C 2 . From another viewpoint, resonant reactors Lr 1 and Lr 2 function so that the current ILs in the smoothing reactor Ls flows into the resonant capacitor Cr.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 2 of 10

(Mode C)

When voltage VQ 2 then reaches a voltage of 0 V at time t 2 , diode D 2 is turned on, bringing the state in mode C. The current ILs flowing in the smoothing reactor Ls flows into resonant reactor Lr 1 , diode D 3 , the resonant capacitor Cr, and diode D 2 . The voltage of the resonant capacitor Cr is applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , causing currents ILr 1 and ILr 2 to change with time in the positive direction. Specifically, part of the current ILs flowing in the smoothing reactor Ls flows into resonant reactor Lr 2 and is then combined with the current that has passed through the resonant capacitor Cr; the resulting current flows into diode D 2 . At this time, the power supply V 1 's energy stored in the smoothing reactor Ls is supplied to power supply V 2 , and the current ILs gradually decreases. Part of the energy stored in the smoothing reactor Ls and resonant reactor Lr 1 is stored in the resonant capacitor Cr.

(Mode D)

IGBTs Q 2 and Q 3 are then turned on at time t 3 . Incidentally, IGBT Q 2 may not be turned on. IGBT Q 2 may be left turned on or off in a period from time t 3 to time t 9 described below. Since voltages VQ 2 and VQ 3 are zero at time t 3 , IGBTs Q 2 and Q 3 undergo zero-voltage switching, bringing the state in mode D. Current ILs flowing in the smoothing reactor Ls is divided into current that flows into resonant reactor Lr 1 and current that flows into resonant reactor Lr 2 . Current ILr 1 flowing in resonant reactor Lr 1 flows into diode D 3 and resonant capacitor Cr and is then combined with current ILr 2 that has passed through resonant reactor Lr 2 ; the resulting current flows into diode D 2 . Currents ILr 1 and ILr 2 change with time in the positive direction, as in mode C, so the value of current ILr 1 decreases and current ILr 2 increases. At this time, the power supply V 1 's energy stored in the smoothing reactor Ls is supplied to power supply V 2 , and the current ILs gradually decreases. Part of the energy stored in the smoothing reactor Ls and resonant reactor Lr 1 is stored in the resonant capacitor Cr.

(Mode E)

Current ILr 1 then changes from negative to positive at time t 4 , bringing the state in mode E. The current ILs flowing in the smoothing reactor Ls flows into resonant reactor Lr 2 and diode D 2 , the power supply V 1 's energy stored in the smoothing reactor Ls is supplied to power supply V 2 , and the current ILs gradually decreases. Since IGBT Q 3 is in the on state, the voltage in the resonant capacitor Cr continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 . The resonant capacitor Cr releases charges in a path extending through IGBT Q 3 , resonant reactor Lr 1 , resonant reactor Lr 2 , and the resonant capacitor Cr in that order; the energy in the resonant capacitor Cr is stored in resonant reactors Lr 1 and Lr 2 .

(Mode F)

IGBTs Q 2 and Q 3 are then turned off at time t 5 , bringing the state in mode F. IGBT Q 2 may be tuned off before time t 9 described below arrives. Voltage VQ 3 at buffer capacitor C 3 connected in parallel to IGBT Q 3 gradually increases from zero at this time. Therefore, IGBT Q 3 undergoes zero-voltage switching at time t 5 . Since voltage VQ 1 decreases as voltage VQ 3 increases, buffer capacitor C 1 releases its stored charges. Currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time. Accordingly, the discharge current in the buffer capacitor C 1 decreases the discharge current in the resonant capacitor Cr and increases the conducting current of diode D 2 . The current ILs flowing in the smoothing reactor Ls continues to flow into resonant reactor Lr 2 and diode D 2 , the power supply V 1 's energy stored in the smoothing reactor Ls is supplied to power supply V 2 , and the current ILs gradually decreases.

(Mode G)

When voltage VQ 1 then reaches a voltage of 0 V at time t 6 , diode D 1 is turned on, bringing the state in mode G. The charging of buffer capacitor C 3 is completed, and the current that has flowed in the resonant capacitor Cr in mode F flows into diode D 2 . The current flowing in diode D 1 passes through resonant reactor Lr 1 and is then combined with the current ILs that has passed through the smoothing reactor Ls; the resulting current flows into resonant reactor Lr 2 and diode D 2 . The voltage of power supply V 2 is applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , the energy stored in the resonant reactors Lr 1 and Lr 2 is supplied to power supply V 2 , and currents ILr 1 and ILr 2 gradually decrease. The conducting currents in diodes D 1 and D 2 also decrease accordingly.

(Mode H)

IGBT Q 1 is then turned on at time t 7 . Since voltage VQ 1 at time t 7 is 0 V, IGBT Q 1 undergoes zero-voltage switching, bringing the state in mode H. The state of the circuit is the same as in mode G, as shown in FIGS. 3G and 3H ; the energy stored in the resonant reactors Lr 1 and Lr 2 is supplied to power supply V 2 , and currents ILr 1 and ILr 2 and the conducting currents in diodes D 1 and D 2 gradually decrease.

(Mode I)

Current ILr 1 then changes from positive to negative at time t 8 , bringing the state in mode I. Since IGBT Q 1 is in the on state, the voltage of power supply V 2 continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 ; the value of current ILr 1 increases in the negative direction and current ILr 2 decreases. Accordingly, current ILs flowing in the smoothing reactor Ls is divided into current that flows into resonant reactor Lr 2 and then diode D 2 and current that flows into resonant reactor Lr 1 and then IGBT Q 1 . If IGBT Q 2 is in the on state, it is tuned off before the period of mode I expires.

(Mode J)

Current ILr 2 then changes from positive to negative at time t 9 , bringing the state in mode J. IGBT Q 2 is in the off state. The J mode assures a period (reverse recovery time) required by diode D 2 to change (reversely recover) from the on state to the off state. The voltage of power supply V 2 continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 ; the values of currents ILr 1 and ILr 2 increase in the negative direction. The reverse current in diode D 2 flows into resonant reactor Lr 2 and is then combined with the current ILs that has passed through the smoothing reactor Ls; the resulting current passes through resonant reactor Lr 1 and flows into IGBT Q 1 .

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 3 of 10

(Mode K)

Diode D 2 reversely recovers at time t 10 , bringing the state in mode K. Voltage VQ 2 at buffer capacitor C 2 connected in parallel to IGBT Q 2 gradually increases from zero at this time. Since voltage VQ 3 decreases as voltage VQ 2 increases, buffer capacitor C 3 releases its stored charges. The discharge current of buffer capacitor C 3 passes through the resonant capacitor Cr and is then combined with the current that has charged buffer capacitor C 2 ; the resulting current flows into resonant reactor Lr 2 and is then combined with the current ILs that has passed through the smoothing reactor Ls; the resulting current flows into resonant reactor Lr 1 . Currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time. Accordingly, the discharge current in the buffer capacitor C 3 decreases the current in IGBT Q 1 .

(Mode L)

When voltage VQ 3 then reaches a voltage of 0 V at time t 11 , diode D 3 is turned on, bringing the state in mode L. The charging of buffer capacitor C 2 from power supply V 2 is completed, reducing the current in IGBT Q 1 . Since currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time, the current for charging the resonant capacitor Cr increases. The power supply V 1 's energy is stored in the smoothing reactor Ls, and the current ILs gradually increases. The voltage of the resonant capacitor Cr is applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , and currents ILr 1 and ILr 2 change with time in the positive direction. The energy stored in resonant reactors Lr 1 and Lr 2 is stored in the resonant capacitor Cr. The energy stored in resonant reactors Lr 1 and Lr 2 within the reserve recovery time of diode D 2 can be thus recovered in the resonant capacitor Cr without causing a surge.

When the current in resonant reactor Lr 2 then reaches zero at time t 12 , the state in mode A is brought again. Resonant current flows in resonant reactors Lr 1 and Lr 2 , the resonant capacitor Cr, and buffer capacitor C 3 at this time, which may cause waveforms to fluctuate.

(From V 2 to V 1 : Buck Operation)

FIGS. 5A to 5F and FIGS. 6G to 6L are circuit diagrams, each of which illustrates a buck operation in the first embodiment of the present invention. FIG. 7 shows voltage and current waveforms for illustrating the buck operation in the first embodiment of the present invention.

The buck operation in the first embodiment of the present invention will be described in detail with reference to FIGS. 5A to 5F , FIGS. 6G to 6L , and FIG. 7 . In the description below, the buck operation is defined as the operation to transfer energy in power supply V 2 to power supply V 1 . FIGS. 5A to 5F and FIGS. 6G to 6L correspond to modes A to L in FIG. 7 .

(Mode A)

In mode A, IGBT Q 2 is in the on state, and IGBTs Q 1 and Q 3 are in the off state. Current ILs in the smoothing reactor Ls gradually increases, the value of current ILr 2 of resonant reactor Lr 2 gradually increases in the negative direction, and the energy in power supply V 2 is stored in the smoothing reactor Ls and resonant reactor Lr 2 . Buffer capacitors C 1 and C 3 and the resonant capacitor Cr are charged with the polarity shown in FIG. 5A at this time.

(Mode B)

IBGT Q 2 is then turned off at time t 1 . Voltage VQ 2 at buffer capacitor C 2 connected in parallel to IGBT Q 2 gradually increases from zero at this time. Therefore, IGBT Q 2 undergoes zero-voltage switching at time t 1 . As voltage VQ 2 increases, the voltage between the emitter of IGBT Q 2 and the emitter of IGBT Q 1 decreases. However, currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time. Buffer capacitors C 1 and C 3 release their stored charges, causing voltages VQ 1 and VQ 3 to drop. When VQ 3 reaches a voltage of 0 V, diode D 3 is turned on. FIG. 5B shows the state of the circuit at that time. Specifically, the current flowing in buffer capacitor C 1 flows into buffer capacitor C 3 (diode D 3 when voltage VQ 3 is 0 V) and the resonant capacitor Cr, and is then combined with the current that has passed through buffer capacitor C 2 ; the resulting current flows into resonant reactor Lr 2 and the smoothing reactor Ls. From another viewpoint, resonant reactors Lr 1 and Lr 2 function so that the current ILs in the smoothing reactor Ls flows into the resonant capacitor Cr.

(Mode C)

When voltage VQ 1 then reaches a voltage of 0 V at time t 2 , diode D 1 is turned on, bringing the state in mode C. The current flowing in diode D 1 flows into diode D 3 , the resonant capacitor Cr, and resonant reactor Lr 2 , and then flows into the smoothing reactor Ls. The voltage in the resonant capacitor Cr is then applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , causing currents ILr 1 and ILr 2 to change with time in the positive direction. Specifically, part of the current flowing in diode D 1 flows into resonant reactor Lr 1 and is then combined with the current that has passed through resonant reactor Lr 2 ; the resulting current flows into the smoothing reactor Ls. At this time, the power supply V 2 's energy stored in the smoothing reactor Ls is supplied to power supply V 1 , and the current ILs gradually decreases. Part of the energy stored in the smoothing reactor Ls and resonant reactor Lr 2 is stored in the resonant capacitor Cr.

(Mode D)

IGBTs Q 1 and Q 3 are then turned on at time t 3 . Incidentally, IGBT Q 1 may not be turned on. IGBT Q 1 may be left turned on or off in a period from time t 3 to time t 9 described below. Since voltages VQ 1 and VQ 3 are zero at time t 3 , IGBTs Q 1 and Q 3 undergo zero-voltage switching, bringing the state in mode D. The current flowing diode D 1 is divided into current that flows into resonant reactor Lr 1 and current that flows into diode D 3 . The current flowing in diode D 3 flows into resonant capacitor Cr and resonant reactor Lr 2 and is then combined with the current that has passed through resonant reactor Lr 1 ; the resulting current flows into the smoothing reactor Ls. Currents ILr 1 and ILr 2 change with time in the positive direction, as in mode C, so the value of current ILr 2 decreases and current ILr 1 increases. At this time, the power supply V 2 's energy stored in the smoothing reactor Ls is supplied to power supply V 1 , and the current ILs gradually decreases. Part of the energy stored in the smoothing reactor Ls and resonant reactor Lr 2 is stored in the resonant capacitor Cr.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 4 of 10

(Mode E)

Current ILr 2 then changes from negative to positive at time t 4 , the state in mode E. The current flowing in diode D 1 flows into resonant reactor Lr 1 and the smoothing reactor Ls, the power supply V 2 's energy stored in the smoothing reactor Ls is supplied to power supply V 1 , and the current ILs gradually decreases. Since IGBT Q 3 is in the on state, the voltage in the resonant capacitor Cr continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 . The resonant capacitor Cr releases charges in a path extending through IGBT Q 3 , resonant reactor Lr 1 , resonant reactor Lr 2 , and the resonant capacitor Cr in that order; the energy in the resonant capacitor Cr is stored in resonant reactors Lr 1 and Lr 2 .

(Mode F)

IGBTs Q 1 and Q 3 are then turned off at time t 5 , bringing the state in mode F. IGBT Q 1 may be tuned off before time t 9 described below arrives. Voltage VQ 3 at buffer capacitor C 3 connected in parallel to IGBT Q 3 gradually increases from zero at this time. Therefore, IGBT Q 3 undergoes zero-voltage switching at time t 5 . Since voltage VQ 2 decreases as voltage VQ 3 increases, buffer capacitor C 2 releases its stored charges. Currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time. Accordingly, the discharge current in the buffer capacitor C 2 decreases the discharge current in the resonant capacitor Cr and increases the conducting current of diode D 1 . The current flowing in diode D 1 continues to flow into resonant reactor Lr 1 and the smoothing reactor Ls, the power supply V 2 's energy stored in the smoothing reactor Ls is supplied to power supply V 1 , and the current ILs gradually decreases.

(Mode G)

When voltage VQ 2 then reaches a voltage of 0 V at time t 6 , diode D 2 is turned on, bringing the state in mode G shown in FIG. 6G . The charging of buffer capacitor C 3 is completed, and the current that has flowed in the resonant capacitor Cr in mode F flows into diode D 1 . The current flowing in diode D 1 passes through resonant reactor Lr 1 and is divided into current that flows into the smoothing reactor Ls and current that flows into resonant reactor Lr 2 . The current flowing in resonant reactor Lr 2 flows into diode D 2 . The voltage of power supply V 2 is applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , the energy stored in the resonant reactors Lr 1 and Lr 2 is recovered into power supply V 2 , and currents ILr 1 and ILr 2 gradually decrease. The conducting currents in diodes D 1 and D 2 also decrease accordingly.

(Mode H)

IGBT Q 2 is then turned on at time t 7 . Since voltage VQ 2 at time t 7 is 0 V, IGBT Q 2 undergoes zero-voltage switching, bringing the state in mode H. The state of the circuit is the same as in mode G; the energy stored in the resonant reactors Lr 1 and Lr 2 is recovered into power supply V 2 , and currents ILr 1 and ILr 2 and the conducting currents in diodes D 1 and D 2 gradually decrease.

(Mode I)

Current ILr 2 then changes from positive to negative at time t 8 , bringing the state in mode I. Since IGBT Q 2 is in the on state, the voltage of power supply V 2 continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 ; the value of current ILr 2 increases in the negative direction and current ILr 1 decreases. The current flowing in diode D 1 and resonant reactor Lr 1 and the current flowing IGBT Q 2 and resonant reactor Lr 2 are combined; the resulting current flows into the smoothing reactor Ls. If IGBT Q 1 is in the on state, it is tuned off before the period of mode I expires.

(Mode J)

Current ILr 1 then changes from positive to negative at time t 9 , bringing the state in mode J. IGBT Q 1 is in the off state. The J mode assures a period (reverse recovery time) required by diode D 1 to change (reversely recover) from the on state to the off state. The voltage of power supply V 2 continues to be applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 ; the values of currents ILr 1 and ILr 2 increase in the negative direction. The current passing through IGBT Q 2 flows into resonant reactor Lr 2 and is then divided into current that flows into smoothing reactor Ls and current that flows into resonant reactor Lr 1 . The current that has passed through resonant reactor Lr 1 is reverse current in diode D 1 .

(Mode K)

Diode D 1 then reversely recovers at time t 10 , bringing the state in mode K. Voltage VQ 1 at buffer capacitor C 1 connected in parallel to IGBT Q 1 gradually increases from zero at this time. Since voltage VQ 3 decreases as voltage VQ 1 increases, buffer capacitor C 3 releases its stored charges. The discharge current of buffer capacitor C 3 passes through the resonant capacitor Cr and is then combined with the current that has passed through IGBT Q 2 ; the resulting current flows into resonant reactor Lr 2 and is divided into current that flows into the smoothing reactor Ls and current that flows into resonant reactor Lr 1 . The current passing through resonant reactor Lr 1 is further divided into current that flows into buffer capacitor C 3 and current that flows into buffer capacitor C 1 . Accordingly, the discharge current in the buffer capacitor C 3 decreases the current in IGBT Q 2 .

(Mode L)

When voltage VQ 3 then reaches a voltage of 0 V at time t 11 , diode D 3 is turned on, bringing the state in mode L. The charging of buffer capacitor C 1 from power supply V 2 is completed, reducing the current in IGBT Q 2 . Since currents ILr 1 and ILr 2 in the resonant reactors Lr 1 and Lr 2 do not change rapidly with time, the current for charging the resonant capacitor Cr increases. The power supply V 2 's energy is stored in the smoothing reactor Ls, and the current ILs gradually increases. The voltage of the resonant capacitor Cr is applied to the series circuit comprising resonant reactors Lr 1 and Lr 2 , and currents ILr 1 and ILr 2 change with time in the positive direction. The energy stored in resonant reactors Lr 1 and Lr 2 is stored in the resonant capacitor Cr. The energy stored in resonant reactors Lr 1 and Lr 2 within the reserve recovery time of diode D 1 can be thus recovered in the resonant capacitor Cr without causing a surge.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 5 of 10

When the current in resonant reactor Lr 1 then reaches zero at time t 12 , the state in mode A is brought again. Resonant current flows in resonant reactors Lr 1 and Lr 2 , the resonant capacitor Cr, and buffer capacitor C 3 at this time, which may cause waveforms to fluctuate.

Operation common to the boost operation and buck operation of the bidirectional DC-DC converter 10 according to the first embodiment of the present invention will be described below. First, in a period in which the smoothing reactor Ls releases the energy it stored, resonant reactors Lr 1 and Lr 2 bring part of the energy released by the smoothing reactor Ls to the resonant capacitor Cr. Diode D 3 is turned on at this time; when IGBT Q 3 is turned on, it undergoes zero-voltage switching. Next, the charges of the resonant capacitor Cr are released through IGBT Q 3 , and the energy in the resonant capacitor Cr is stored in resonant reactors Lr 1 and Lr 2 . When IGBT Q 3 is turned off, the current stored in resonant reactors Lr 1 and Lr 2 causes diodes D 1 and D 2 to be turned on, thereby causing IGBTs Q 1 and Q 2 to under go zero-voltage switching. The voltage stored in the resonant capacitor Cr and the current stored in resonant reactors Lr 1 and Lr 2 have the same polarity in the boost operation and buck operation. That is, the same principle is used to achieve soft switching, independently of the operation direction.

As described above, the bidirectional DC-DC converter 10 achieves zero-voltage soft switching for all elements without changing circuit connections in bidirectional operation. The energy stored in resonant reactors Lr 1 and Lr 2 within the reserve recovery times of diodes D 1 and D 2 can be thus recovered in the resonant capacitor Cr without causing a surge.

The bidirectional DC-DC converter 10 can thus accomplish easy control by use of a control means 2 that provides a period in which both IGBTs Q 1 and Q 3 are turned off together in the boost operation and another period in which both IGBTs Q 2 and Q 3 are turned off together in the buck operation and turns these IGBTs on and off alternately. If the direction in which the current flows in the smoothing reactor Ls is detected, it can be determined whether the operation is the boost operation or the buck operation. Therefore, a current sensor 3 for sensing the current flowing in the smoothing reactor Ls is provided. When the sensor 3 senses that the current flowing in the smoothing reactor Ls is directed from the smoothing reactor Ls to node N 0 , the control means 2 provides a period in which both IGBTs Q 1 and Q 3 are turned off together, and turns on and off them alternately. When the sensor 3 senses that the current flowing in the smoothing reactor Ls is directed from node N 0 to the smoothing reactor Ls, the control means 2 provides a period in which both IGBTs Q 2 and Q 3 are turned off together, and turns on and off them alternately. This facilitates control.

Although resonant reactors Lr 1 and Lr 2 are magnetically coupled in the first embodiment, they do not necessarily have to be magnetically coupled. Either of resonant reactors Lr 1 and Lr 2 can be eliminated for compactness, as described in second and third embodiments below.

Second Embodiment

FIG. 8 shows a main circuit diagram of a bidirectional DC-DC converter according to a second embodiment of the present invention. The bidirectional DC-DC converter 11 in the second embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 10 in the first embodiment shown in FIG. 1 and restoring the connection.

The circuit operations and control method are the same as with the bidirectional DC-DC converter 10 in the first embodiment. The waveforms in the circuits are the same as in FIGS. 4 and 7 .

Third Embodiment

FIG. 9 shows a main circuit diagram of a bidirectional DC-DC converter according to a third embodiment of the present invention. The bidirectional DC-DC converter 12 in the third embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 10 in the first embodiment shown in FIG. 1 and restoring the connection.

The circuit operations and control method are the same as with the bidirectional DC-DC converter 10 in the first embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Fourth Embodiment

In the bidirectional DC-DC converter 10 in the first embodiment, IGBTs Q 1 and Q 2 are not interconnected, so an IGBT module in which a collector and emitter of two IGBTs are interconnected cannot be used. In the bidirectional DC-DC converter 13 in a fourth embodiment, therefore, its circuit connection is such that an IGBT module can be used.

FIG. 10 shows a main circuit diagram of a bidirectional DC-DC converter according to the fourth embodiment of the present invention. In the main circuit of the bidirectional DC-DC converter 13 in the fourth embodiment, first IGBT Q 1 and resonant reactor Lr 1 in FIG. 1 are mutually exchanged and second IGBT Q 2 and resonant reactor Lr 2 also in FIG. 1 are mutually exchanged; the circuit elements are the same as in FIG. 1 .

Only differences from FIG. 1 will be described. The collector of IGBT Q 1 and the emitter of IGBT Q 2 are connected to node N 0 . Resonant reactor Lr 1 is connected between the emitter of IGBT Q 1 and negative bus LN 0 , and resonant reactor Lr 2 is connected between the collector of IGBT Q 2 and positive bus LN 2 . The connection point between IGBT Q 1 and resonant reactor Lr 1 is called node N 1 , and the connection point between IGBT Q 2 and resonant reactor Lr 2 is called node N 2 . IGBT Q 3 and a resonant capacitor Cr are interconnected in series to form an active resonant circuit. Another active resonant circuit is formed between node N 1 and node N 2 in such a way that the emitter of IGBT Q 3 faces node N 2 .

Symbols representing the voltages and currents in the circuit diagram in FIG. 10 will be defined. Current ILr 1 flowing in resonant reactor Lr 1 is positive when it flows from negative bus LN 0 to node N 1 , and current ILr 2 flowing in resonant reactor Lr 2 is positive when it flows from node N 2 to positive bus LN 2 . The definitions of the symbols for the other voltages and currents are the same as for the bidirectional DC-DC converter 10 in the first embodiment.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 6 of 10

FIGS. 11A to 11F and FIGS. 12G to 12L are circuit diagrams, each of which illustrates a boost operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention. FIGS. 13A to 13F and FIGS. 14G to 14L are circuit diagrams, each of which illustrates a buck operation of the bidirectional DC-DC converter in the fourth embodiment of the present invention. The control method is the same as for the bidirectional DC-DC converter 10 in the first embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Resonant reactors Lr 1 and Lr 2 incorporated in the bidirectional DC-DC converter 13 in the fourth embodiment may be magnetically coupled (not shown). The use of a common core increases the usage efficiency of the core and makes the core compact. Alternatively, either of resonant reactors Lr 1 and Lr 2 may be eliminated for compactness, as described in a fifth embodiment ( FIG. 15 ) to a tenth embodiment ( FIG. 20 ) below.

Fifth Embodiment

FIG. 15 shows a main circuit diagram of a bidirectional DC-DC converter according to a fifth embodiment of the present invention. The bidirectional DC-DC converter 14 in the fifth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to positive bus line LN 1 .

The circuit operations and control method are the same as with the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 . The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Sixth Embodiment

FIG. 16 shows a main circuit diagram of a bidirectional DC-DC converter according to a sixth embodiment of the present invention. The bidirectional DC-DC converter 15 in the sixth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to positive bus line LN 2 .

Seventh Embodiment

FIG. 17 shows a main circuit diagram of a bidirectional DC-DC converter according to a seventh embodiment of the present invention. The bidirectional DC-DC converter 16 in the seventh embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to negative bus line LN 0 .

Eighth Embodiment

FIG. 18 shows a main circuit diagram of a bidirectional DC-DC converter according to an eighth embodiment of the present invention. The bidirectional DC-DC converter 17 in the eighth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to positive bus line LN 1 .

The circuit operations and control method are the same as with the bidirectional DC-DC converter 13 in the fourth embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Ninth Embodiment

FIG. 19 shows a main circuit diagram of a bidirectional DC-DC converter according to a ninth embodiment of the present invention. The bidirectional DC-DC converter 18 in the ninth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to positive bus line LN 2 .

Tenth Embodiment

FIG. 20 shows a main circuit diagram of a bidirectional DC-DC converter according to a tenth embodiment of the present invention. The bidirectional DC-DC converter 19 in the tenth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 13 in the fourth embodiment shown in FIG. 10 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to negative bus line LN 0 .

Eleventh Embodiment

FIG. 21 shows a main circuit diagram of a bidirectional DC-DC converter according to an eleventh embodiment of the present invention. The bidirectional DC-DC converter 30 in FIG. 21 differs from the first embodiment in FIG. 1 in that the circuit to the right of smoothing reactor Ls in FIG. 1 is also provided to the left of smoothing reactor Ls. Accordingly, the bidirectional DC-DC converter 30 has a first vertical arm formed by interconnecting first and second switching devices Q 11 and Q 12 in series and a second vertical arm formed by interconnecting third and fourth switching devices Q 21 and Q 22 in series. The bidirectional DC-DC converter 30 also has a smoothing reactor Ls, one end of which is connected to the series connection point between first and second switching devices Q 11 and Q 12 and the other end of which is connected to the series connection point between third and fourth switching devices Q 21 and Q 22 . The bidirectional DC-DC converter 30 also includes a first smoothing reactor Cs 1 connected to both ends of the first vertical arm and connected in parallel to a first power supply V 1 as well as a second smoothing reactor Cs 2 connected to both ends of the second vertical arm and connected in parallel to a second power supply V 2 . First resonant reactors Lr 11 and Lr 12 are connected in series to first and second switching devices Q 11 and Q 12 , respectively. Similarly, second resonant reactors Lr 21 and Lr 22 are connected in series to the third and fourth switching devices Q 21 and Q 22 , respectively. As described in the previous embodiments, both resonant reactors Lr 11 and Lr 12 and both resonant reactors Lr 21 and Lr 22 are not necessarily provided; at least either of them is sufficient.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 7 of 10

A first active resonant circuit 6 is formed by connecting a fifth switching device Q 13 to a first resonant capacitor Cr 1 in series. Similarly, a second active resonant circuit 7 is formed by connecting a sixth switching device Q 23 to a second resonant capacitor Cr 2 in series. A bidirectional DC-DC converter 30 enabling soft switching is thereby structured.

This embodiment has boost and buck functions executed in bidirectional operations in which energy in power supply V 1 is transferred to power supply V 2 and energy in power supply V 2 is transferred to power supply V 1 . In this embodiment, the bidirectional DC-DC converter 10 in the first embodiment in FIG. 1 is duplicated, so circuit operations and a control method would be understood as in FIG. 1 .

Twelfth Embodiment

Next, bidirectional DC-DC converters in a twelfth embodiment ( FIG. 22 ) to a twenty-first embodiment ( FIG. 39 ) will be described in which only a pair of main elements are used as the main switching devices and boost and buck functions are used to transfer energy during bidirectional operations between power supply V 1 and power supply V 2 .

FIG. 22 shows a main circuit diagram of a bidirectional DC-DC converter 20 according to the twelfth embodiment of the present invention. A vertical arm is provided with smoothing capacitors Cs 1 and Cs 2 ; power supply V 1 is connected in parallel to Cs 1 and power supply V 2 is connected in parallel to Cs 2 . Voltage conversion is performed between power supply V 1 and power supply V 2 .

FIGS. 23A to 23F and FIGS. 24G to 24L are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 . FIGS. 25A to 25F and FIGS. 26G to 26L are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the twelfth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 . When a period in which both IGBTs Q 1 and Q 3 are turned off together is provided and they are turned on and off alternately, energy in power supply V 1 is transferred to power supply V 2 . When a period in which both IGBTs Q 2 and Q 3 are turned off together is provided and they are turned on and off alternately, energy in power supply V 2 is transferred to power supply V 1 . Other schemes in the control method are the same as in the bidirectional DC-DC converter 10 in the first embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Resonant reactors Lr 1 and Lr 2 incorporated in the bidirectional DC-DC converter 20 in the twelfth embodiment in FIG. 22 may be magnetically coupled (not shown). The use of a common core increases the usage efficiency of the core and makes the core compact. Alternatively, either of resonant reactors Lr 1 and Lr 2 may be eliminated for compactness, as described in thirteenth and fourteenth embodiments below.

Thirteenth Embodiment

FIG. 27 shows a main circuit diagram of a bidirectional DC-DC converter according to a thirteenth embodiment of the present invention. The bidirectional DC-DC converter 21 in the thirteenth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 20 in the twelfth embodiment shown in FIG. 22 and restoring the connection. The circuit operations and control method are the same as with the bidirectional DC-DC converter 20 in the twelfth embodiment shown in FIG. 22 . The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Fourteenth Embodiment

FIG. 28 shows a main circuit diagram of a bidirectional DC-DC converter according to a fourteenth embodiment of the present invention. The bidirectional DC-DC converter 22 in the fourteenth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 20 in the twelfth embodiment shown in FIG. 22 and restoring the connection. The circuit operations and control method are the same as with the bidirectional DC-DC converter 20 in the twelfth embodiment shown in FIG. 22 . The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Fifteenth Embodiment

In the above-mentioned bidirectional DC-DC converter 20 in the twelfth embodiment in FIG. 22 , IGBTs Q 1 and Q 2 are not interconnected, so an IGBT module in which a collector and emitter of two IGBTs are interconnected cannot be used. In the bidirectional DC-DC converter 23 in the fifteenth embodiment, therefore, its circuit connection is such that an IGBT module can be used.

FIG. 29 shows a main circuit diagram of a bidirectional DC-DC converter according to a fifteenth embodiment of the present invention. In the bidirectional DC-DC converter 23 in the fifteenth embodiment, first IGBT Q 1 and resonant reactor Lr 1 in FIG. 22 are mutually exchanged and second IGBT Q 2 and resonant reactor Lr 2 also in FIG. 22 are mutually exchanged; the circuit elements are the same as in FIG. 22 .

Only differences from FIG. 22 will be described. The collector of IGBT Q 1 and the emitter of IGBT Q 2 are connected to node N 0 . Resonant reactor Lr 1 is connected between the emitter of IGBT Q 1 and negative bus LN 0 , and resonant reactor Lr 2 is connected between the collector of IGBT Q 2 and positive bus LN 2 . The connection point between IGBT Q 1 and resonant reactor Lr 1 is called node N 1 , and the connection point between IGBT Q 2 and resonant reactor Lr 2 is called node N 2 . IGBT Q 3 and a resonant capacitor Cr are interconnected in series to form an active resonant circuit. Another active resonant circuit is formed between node N 1 and node N 2 in such a way that the emitter of IGBT Q 3 faces node N 2 .

Symbols representing the voltages and currents in the circuit diagram in FIG. 29 will be defined. Current ILr 1 flowing in resonant reactor Lr 1 is positive when it flows from negative bus LN 0 to node N 1 , and current ILr 2 flowing in resonant reactor Lr 2 is positive when it flows from node N 2 to positive bus LN 2 . The definitions of the symbols for the other voltages and currents are the same as for the bidirectional DC-DC converter 20 in the twelfth embodiment in FIG. 22 .

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 8 of 10

FIGS. 30A to 30F and FIGS. 31G to 31L are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 1 to power supply V 2 . FIGS. 32A to 32F and FIGS. 33G to 33L are circuit diagrams, each of which illustrates an operation in which the bidirectional DC-DC converter in the fifteenth embodiment of the present invention transfers energy in power supply V 2 to power supply V 1 . The control method is the same as in the bidirectional DC-DC converter 20 in the twelfth embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Resonant reactors Lr 1 and Lr 2 incorporated in the bidirectional DC-DC converter 23 in the fifteenth embodiment may be magnetically coupled (not shown). The use of a common core increases the usage efficiency of the core and makes the core compact. Alternatively, either of resonant reactors Lr 1 and Lr 2 may be eliminated for compactness, as described in a sixteenth embodiment ( FIG. 34 ) to a twenty-first embodiment ( FIG. 39 ) below.

Sixteenth Embodiment

FIG. 34 shows a main circuit diagram of a bidirectional DC-DC converter according to a sixteenth embodiment of the present invention.

The bidirectional DC-DC converter 24 in the sixteenth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to positive bus line LN 1 .

The circuit operations and control method are the same as with the bidirectional DC-DC converter 23 in the fifteenth embodiment shown in FIG. 29 . The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Seventeenth Embodiment

FIG. 35 shows a main circuit diagram of a bidirectional DC-DC converter according to a seventeenth embodiment of the present invention. The bidirectional DC-DC converter 25 in the seventeenth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to positive bus line LN 2 .

The circuit operations and control method are the same as with the bidirectional DC-DC converter 23 in the fifteenth embodiment. The waveforms at various parts in the circuit are the same as in FIGS. 4 and 7 .

Eighteenth Embodiment

FIG. 36 shows a main circuit diagram of a bidirectional DC-DC converter according to an eighteenth embodiment of the present invention. The bidirectional DC-DC converter 26 in the eighteenth embodiment is structured by removing resonant reactor Lr 2 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 2 and reconnecting it to negative bus line LN 0 .

Nineteenth Embodiment

FIG. 37 shows a main circuit diagram of a bidirectional DC-DC converter according to a nineteenth embodiment of the present invention. The bidirectional DC-DC converter 27 in the nineteenth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to positive bus line LN 1 .

Twentieth Embodiment

FIG. 38 shows a main circuit diagram of a bidirectional DC-DC converter according to a twentieth embodiment of the present invention. The bidirectional DC-DC converter 28 in the twentieth embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to positive bus line LN 2 .

Twenty-First Embodiment

FIG. 39 shows a main circuit diagram of a bidirectional DC-DC converter according to a twenty-first embodiment of the present invention. The bidirectional DC-DC converter 29 in the twenty-first embodiment is structured by removing resonant reactor Lr 1 from the bidirectional DC-DC converter 23 in the fifteenth embodiment in FIG. 29 and restoring the connection and by disconnecting one end of the active resonant circuit from node N 1 and reconnecting it to negative bus line LN 0 .

So far, bidirectional DC-DC converters in 21 different embodiments of the present invention have been described. Next, exemplary applications, in which any one of a variety of these embodiments can be used, will be described.

FIG. 40 shows the general structure of a power supply apparatus in a hybrid vehicle, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus. One end of the bidirectional DC-DC converter 10 is connected to an inverter 105 connected to a motor 106 , and the other end is connected to a DC power supply 101 . The DC power supply 101 is connected to a DC-DC converter 102 , to which another DC power supply 103 supplying a voltage lower than the voltage of the DC power supply 101 is connected. Energy is transferred and received between the DC power supplies 101 and 103 . The power supply 103 is connected to, for example, an electronic device 104 operating on 14 V.

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 9 of 10

When the inverter 105 powers the motor 106 , the bidirectional DC-DC converter 10 boosts the voltage of the DC power supply 101 and supplies its energy to the inverter 105 . When the inverter 105 causes the motor 106 to regenerate electric power, the bidirectional DC-DC converter 10 drops the voltage of the inverter 105 and supplies regenerative energy to the DC power supply 101 . The DC power supply 101 may be a capacitor.

FIG. 41 shows the general structure of a power supply apparatus in a vehicle, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus. The bidirectional DC-DC converter 10 is connected between a DC power supply 111 for supplying energy to an electronic device 113 operating on 14 V and another DC power supply 112 for supplying electric power to an electronic device 114 operating on 42 V. When energy is supplied from the DC power supply 111 to the DC power supply 112 , a boost operation is performed. When energy is supplied from the DC power supply 112 to the DC power supply 111 , a buck operation is performed.

FIG. 42 shows the general structure of a power supply apparatus in a fuel cell-powered vehicle, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus. One end of the bidirectional DC-DC converter 10 is connected to an inverter 125 connected to a motor 126 , and the other end is connected to a DC power supply 121 . The DC power supply 121 is connected to a DC-DC converter 122 , to which a fuel cell 123 is connected; the DC-DC converter 122 converts the voltage of the fuel cell 123 and supplies its energy to the DC power supply 121 .

This structure enables the energy of the fuel cell 123 to be stored in the DC power supply 121 . When the inverter 125 powers the motor 126 , the bidirectional DC-DC converter 10 boosts the voltage of the DC power supply 121 and supplies its energy to the inverter 125 . When the inverter 125 causes the motor 126 to regenerate electric power, the bidirectional DC-DC converter 10 drops the voltage of the inverter 125 and supplies regenerative energy to the DC power supply 121 . The DC power supply 121 may be a capacitor.

According to the embodiments in FIGS. 40 to 42 , a bidirectional DC-DC converter in the present invention can be used to increase its conversion efficiency and make it lightweight, so an increase in mileage and improvement in running performance can be achieved. Since the DC-DC converter can also be made compact, the degree of freedom in vehicle design increases, enabling various types of vehicle structures to be designed.

FIG. 43 shows the general structure of a power supply apparatus in a mobile unit, a bidirectional DC-DC converter according to the present invention being used in the power supply apparatus, a fuel cell being provided as a power supply. A DC-DC converter 133 converts the voltage of the fuel cell 134 and supplies its energy to a mobile load 132 . One end of the bidirectional DC-DC converter 10 is connected to a capacitor 131 , and the other end is connected to the mobile load 132 . This structure enables the mobile load to receive energy from the fuel cell 134 , the capacitor 131 , or both. It is also possible to store the energy of the fuel cell 134 in the capacitor 131 . A DC power supply may be used instead of the capacitor 131 . A system in which the fuel cell 134 is replaced with a solar cell may also be possible.

According to this embodiment, the use of the inventive bidirectional DC-DC converter increases the conversion efficiency, thereby prolonging a continuous operation time. An increase in conversion efficiency also enables the electricity storage unit to be made compact and lightweight. Since the DC-DC converter can be made compact and lightweight, a compact, lightweight mobile unit can be achieved.

FIG. 44 shows the general structure of a power supply apparatus in a grid-connected photovoltaic generation system, a bidirectional DC-DC converter according to the present invention and an electricity storage unit being used in the power supply apparatus. A DC-DC converter 143 converts the voltage of a solar cell 144 and supplies its energy to the DC side of an inverter 142 . One end of the bidirectional DC-DC converter 10 is connected to a DC power supply 141 , and the other end is connected to the DC side of the inverter 142 . The AC side of the inverter 142 is connected to an AC load 145 , such as a home electric appliance, and an AC power supply system 146 .

This structure enables the AC load 145 and AC power supply system 146 to receive energy from the solar cell 144 , the DC power supply 141 , or both. It is also possible to store the energy of the solar cell 144 in the DC power supply 141 . Furthermore, it is also possible to store energy from the AC power supply system 146 into the DC power supply 141 . A capacitor may be used instead of the DC power supply 141 . A system in which the solar cell 144 is replaced with a fuel cell may be possible.

According to this embodiment, the use of the inventive bidirectional DC-DC converter increases the conversion efficiency, thereby reducing light and fuel expenses, carbon dioxide emission, and primary energy supply as well as making the electricity storage unit compact. Furthermore, an amount to absorb variations in power is assured and noise is reduced, making a contribution to providing quality electric power.

FIG. 45 shows the general structure of a power supply apparatus in a loop controller, a bidirectional DC-DC converter according to the present invention and an electricity storage unit being used in the power supply apparatus. The AC side of an inverter 152 is connected to a power system 154 ; the AC side of an inverter 153 is connected to another power system 155 . The DC sides of the inverters 152 and 153 are mutually connected. The inverters 152 and 153 transfer and receive energy between the power systems 154 and 155 . One end of the bidirectional DC-DC converter 10 is connected to a DC power supply 151 , and the other end is connected to the DC sides of the inverters 152 and 153 .

›DESCRIPTION OF THE PREFERRED EMBODIMENTS · 10 of 10

This structure enables energy to be stored in the DC power supply 151 from the power supply system 154 , the power supply system 155 , or both. The structure also enables the DC power supply 151 to supply its energy to the power supply system 154 , the power supply system 155 , or both. A capacitor may be used instead of the DC power supply 151 . This type of loop controller has functions for controlling power flow of power systems and storing energy, by which the power quality of the power systems is improved and the load ratio is increased.

According to this embodiment, the use of the inventive bidirectional DC-DC converter increases the conversion efficiency, thereby reducing carbon dioxide emission and primary energy supply as well as making the electricity storage unit compact. Furthermore, an amount to deliver electricity and an amount to absorb variations in power are assured and noise is reduced, making a contribution to providing quality power.

FIG. 46 shows the general structure of a power supply apparatus for an elevator, a bidirectional DC-DC converter according to the present invention and an electricity storage unit being used in the power supply apparatus. An AC-DC converter 162 rectifies the voltage of an AC power supply 164 , and supplies its energy to an inverter 163 connected to a motor 165 . One end of the bidirectional DC-DC converter 10 is connected to a capacitor 161 , and the other is connected to the DC side of the AC-DC converter 162 .

This structure enables the inverter 163 to receive energy from the AC power supply 164 , the capacitor 161 , or both so as to power the motor 165 . It is also possible to store in the capacitor 161 energy obtained when the inverter 163 causes the motor 165 to regenerate electric power. A DC power supply may be used instead of the capacitor 161 .

According to this embodiment, the use of the inventive bidirectional DC-DC converter increases the conversion efficiency, thereby reducing carbon dioxide emission and primary energy supply as well as making the electricity storage unit compact. Furthermore, an amount to absorb variations in power is assured and noise is reduced, making a contribution to providing quality power.

FIG. 47 shows the general structure of an uninterruptible power supply in which a bidirectional DC-DC converter according to the present invention is used. The uninterruptible power supply 178 comprises a bidirectional DC-DC converter 10 , a DC power supply 171 , a DC-DC converter 172 , an AC-DC converter 173 , and an inverter 174 . The AC-DC converter 173 rectifies the voltage of an AC power supply 175 and supplies its energy to the DC-DC converter 172 and the inverter 174 . The DC-DC converter 172 supplies energy to a DC load 177 , and the inverter 174 supplies energy to an AC load 176 . One end of the bidirectional DC-DC converter 10 is connected to a DC power supply 171 , and the other end is connected to the DC side of the AC-DC converter 173 .

In this structure, when power of the AC power supply 175 fails or its voltage drops for an instant, the bidirectional DC-DC converter 10 operates, so the energy of the DC power supply 171 can be supplied to the DC-DC converter 172 and the inverter 174 . Energy supply to the DC load 177 and AC load 176 can also be continued. When the AC power supply 175 recovers, the energy of the AC power supply 175 , resulting from voltage rectification by the AC-DC converter 173 , is stored in the DC power supply 171 . A capacitor may be used instead of the DC power supply 171 .

According to this embodiment, the use of the inventive bidirectional DC-DC converter increases the conversion efficiency, thereby making the electricity storage unit compact and prolonging a power supply time during an outage.

Claims as granted

20 claims

Classifications

5 codes

IPC · International Patent Classification

Section G — Physics

G05F1/00

USPC · US Patent Classification

323/222323/282323/271323/232

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Bao Q Vu

art unit 2838 · TC 2800

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