Method for inhibiting thermal run-away

This invention relates, in general, to power converters and, more particularly, to multi-phase power converters.

Power converters are used in a variety of electronic products including automotive, aviation, telecommunications, and consumer electronics. Power converters such as Direct Current to Direct Current (“DC-DC”) converters have become widely used in portable electronic products such as laptop computers, personal digital assistants, pagers, cellular phones, etc., which are typically powered by batteries. DC-DC converters are capable of delivering multiple voltages from a single voltage independent of the load current being drawn from the converter or from any changes in the power supply feeding the converter. One type of DC-DC converter that is used in portable electronic applications is a buck converter. This converter, also referred to as a switched mode power supply, is capable of switching an input voltage from one voltage level to a lower voltage level. A buck converter is typically controlled by a controller that can be configured to be a multi-phase controller having a plurality of output current channels that switch at different times. The output currents flowing in the output current channels are summed and delivered to the load. An advantage of this configuration is that each channel conducts a portion of the total load current. For example, in a 4-phase buck controller, each channel conducts 25% of the output current. This lowers the power dissipated by each output. A drawback with a multi-phase buck controller is that when the currents are not balanced, one of the current channels will conduct more current than the other current channels, which could lead to thermal failure. Another drawback is that a dynamic load coupled to the controller may have the same repetition rate as one of the outputs of the multi-phase buck converter. In this case, the currents in the channels become unbalanced causing the converter to suffer thermal failure.

Hence, a need exists for a multi-phase controller circuit and a method of operating the multi-phase controller circuit that maintains a balanced current at its outputs. In addition, it is desirable for the multi-phase controller circuit to be cost and time efficient to manufacture.

The present invention will be better understood from a reading of the following detailed description, taken in conjunction with the accompanying drawing figures, in which like reference characters designate like elements, and in which:

FIG. 1 is schematic diagram of a multi-phase controller circuit in accordance with an embodiment of the present invention;

FIG. 2 is a schematic diagram of a portion of the multi-phase controller circuit of FIG. 1 ;

FIG. 3 is a timing diagram of the operation of the multi-phase controller circuit of FIG. 1 ;

FIG. 4 is a schematic diagram of a multi-phase controller circuit in accordance with another embodiment of the present invention;

FIG. 5 is a schematic diagram of a portion of the multi-phase controller circuit of FIG. 4 ; and

FIG. 6 is a schematic diagram of a multi-phase controller circuit in accordance with yet another embodiment of the present invention.

Generally, the present invention provides a method for balancing current in a multi-phase power converter at varying load transition rates. The multi-phase power converter comprises an oscillator or other ramp signal generator, a pulse width modulator, and at least one power stage. In accordance with one aspect of the present invention, the current is balanced by dithering an output signal of the oscillator or the ramp signal generator. It should be understood that dithering the output signal is defined as constantly varying the frequency of the oscillator output signal or the ramp signal. Dithering the oscillator output signal or the ramp signal keeps the load step rate and the switching frequency of the multi-phase power controller from matching for a significant period of time. Balancing the current inhibits thermal run-away in the multi-phase power converter.

In accordance with another aspect of the present invention, the current is balanced by suspending the oscillator output signal. This introduces a phase delay in the output signals so that the output signals are not synchronized to the load step rate.

In accordance with yet another aspect of the present invention, the current is balanced by dithering the oscillator output signal or the ramp signal and suspending the oscillator output signal.

FIG. 1 is a block diagram of a multi-phase power converter 10 manufactured in a semiconductor substrate in accordance with an embodiment of the present invention. What is shown in FIG. 1 is a Pulse Width Modulator (“PWM”) circuit 12 having “n” sets of inputs 12 1 , 12 2 , 12 3 , . . . , 12 n , where “n” is an integer. Each set of the “n” sets of inputs comprises an error input 12 nA and an oscillation input 12 nB . It should be noted that the letters “A” and “B” are used in the reference characters to distinguish between error inputs and oscillation inputs, respectively. Thus, input 12 1 comprises an error input 12 1A and an oscillation input 12 1B ; input 12 2 comprises an error input 12 2A and an oscillation input 12 2B ; input 12 3 comprises an error input 12 3A and an oscillation input 12 3B ; and input 12 n comprises an error input 12 nA and an oscillation input 12 nB .

Multi-phase power converter 10 further includes an error amplifier 16 having an output 17 connected to error inputs 12 1A , 12 2A , 12 3A , . . . , 12 nA and an oscillator 18 having an input 32 and a plurality of outputs, wherein the plurality of outputs are connected to corresponding oscillation inputs 12 1B , 12 2B , 12 3B , . . . , 12 nB . In accordance with one embodiment, error amplifier 16 comprises an operational amplifier 20 connected in a negative feedback configuration in which an impedance 22 is coupled between the output of operational amplifier 20 and its inverting input and an impedance 24 is connected to the inverting input of operational amplifier 20 . By way of example, impedance 22 comprises a capacitor 26 coupled in parallel with a series connected resistor 28 and capacitor 30 , and impedance 24 comprises a resistor. The non-inverting input of operational amplifier 20 is coupled for receiving a reference voltage level V REF1 . It should be understood that the feedback configuration of error amplifier 16 is not a limitation of the present invention and that it may be realized using other feedback configurations known to those skilled in the art.

Outputs 14 1 , 14 2 , 14 3 , . . . , 14 n of PWM circuit 12 are connected to corresponding inputs of power stages 34 1 , 34 2 , 34 3 , . . . , 34 n , respectively. One output of power stage 34 1 is connected to an output node 50 . Similarly, outputs of power stages 34 2 , 34 3 , . . . , 34 n are connected to output node 50 . Power stages 34 1 , 34 2 , 34 3 , . . . , 34 n , have current sense modules 35 1 , 35 2 , 35 3 , . . . , 35 n , respectively, that generate feedback currents I FEED1 , I FEED2 , I FEED3 , . . . , I FEEDn that are proportional to the currents flowing through energy storage elements 44 1 , 44 2 , 44 3 , . . . , 44 n . Feedback current signals I FEED1 , I FEED2 , I FEED3 , . . . , I FEEDn , are fed back to PWM circuit 12 through feedback interconnects 37 1 , 37 2 , 37 3 , . . . , 37 n , respectively. Circuit configurations for current sense modules are known to those skilled in the art.

Power stages 34 1 , 34 2 , 34 3 , . . . , 34 n , comprise driver circuits 54 1 , 54 2 , 54 3 , . . . , 54 n , respectively, having inputs that serve as the inputs of power stage 34 1 , 34 2 , 34 3 , . . . , 34 n , high-side driver outputs connected to the gates of the respective switching transistors 56 1 , 56 2 , 56 3 , . . . , 56 n , and low-side driver outputs connected to the gates of the respective switching transistors 58 1 , 58 2 , 58 3 , . . . , 58 n . The drains of high-side switching transistors 56 1 , 56 2 , 56 3 , . . . , 56 n are coupled for receiving a source of operating potential such as, for example, V CC , and the sources of high-side switching transistors 56 1 , 56 2 , 56 3 , . . . , 56 n are connected to the respective drains of low-side switching transistors 58 1 , 58 2 , 58 3 , . . . , 58 n . The sources of low-side switching transistors 58 1 , 58 2 , 58 3 , . . . , 58 n are coupled for receiving a source of operating potential such as, for example, V SS . The commonly connected sources and drains of transistors 56 1 , 56 2 , 56 3 , . . . , 56 n , and transistors 58 1 , 58 2 , 58 3 , . . . , 58 n , respectively, are connected to a terminal of the respective energy storage elements 44 1 , 44 2 , 44 3 , . . . , 44 n . The other terminals of energy storage elements 44 1 , 44 2 , 44 3 , . . . , 44 n serve as outputs of power stages 34 1 , 34 2 , 34 3 , . . . , 34 n . By way of example, energy storage elements 44 1 , 44 2 , 44 3 , . . . , 44 n are inductors. It should be noted that for “n” equal to two, power converter 10 is a 2-phase power converter; for “n” equal to three, power converter 10 is a 3-phase power converter; for “n” equal to four, power converter 10 is a 4-phase power converter, etc.

An oscillator control circuit 60 is coupled to input 32 of oscillator 18 via a resistor 59 . More particularly, oscillator control circuit 60 has an input 61 connected to output 17 of error amplifier 16 for receiving compensation voltage V COMP , an input 63 coupled for receiving a reference voltage V REF2 , and an output 65 connected to input 32 of oscillator 18 . Briefly referring to FIG. 2 , a schematic diagram of oscillator control circuit 60 in accordance with one embodiment of the present invention is illustrated. What is shown in FIG. 2 is a comparator 62 having an inverting input which serves as input 61 of oscillator control circuit 60 , a non-inverting input which serves as input 63 of oscillator control circuit 60 , and an output. Preferably, comparator 62 has hysteresis. The output of comparator 62 is connected to the gate of a Field Effect Transistor (“FET”) 76 . The source of FET 76 is coupled for receiving a source of operating potential such as, for example, V SS , and the drain of FET 76 serves as an open drain output and as output 65 of oscillator control circuit 18 .

A load 80 is coupled between output node 50 and a source of operating potential such as, for example, V SS . An output capacitor 82 is connected in parallel with load 80 . Output node 50 is connected in a feedback configuration to impedance 24 .

In accordance with one embodiment, current imbalance and, therefore, thermal runaway is inhibited by adjusting the frequency of power converter 10 so that a load step period and the on-time of multi-phase power converter 10 are in a temporal relationship. It should be understood that the on-time of multi-phase power converter 10 is the time during which one or more of high side switching transistors 56 1 - 56 n is on. The temporal relationship is such that the load step period and the on-time of multi-phase power converter 10 is not coincident, the same, or similar with the load step current for an extended period of time. This is accomplished by programming oscillator 18 to generate a plurality of oscillator output signals having predetermined frequency and phase relationships. In accordance with one embodiment, power converter 10 is a 4-phase power converter, i.e., variable “n” is equal to 4, and oscillator 18 generates four triangular waveforms that are separated by 90 angular degrees. Oscillator 18 may be programmed by coupling a resistor 84 between input 32 of oscillator 18 and a source of operating potential, such as, for example, V SS . The structure for controlling the output frequency of oscillator 18 is not limited to being a resistor. Other circuit networks can be coupled to input 32 . For example, a resistor divider network may be coupled to input 32 . Oscillator 18 transmits the oscillator output signals to inputs 12 1B , 12 2B , 12 3B , . . . , 12 nB of PWM circuit 12 . It should be noted that when power converter 10 is a 2-phase power converter, oscillator 18 generates two triangular waveforms that are separated by 180 degrees; when power converter 10 is a 3-phase power converter, oscillator 18 generates three triangular waveforms that are separated by 120 degrees, when power converter 10 is an n-phase power converter, oscillator 18 generates “n” triangular waveforms that are separated by 360/n degrees. As discussed hereinbefore, power converter 10 can be a 2-phase power converter, a 3-phase power converter, a 4-phase power converter, a 5-phase power converter, etc. It should be further noted that the oscillator output signal is also referred to as a ramp signal.

In addition, error amplifier 16 transmits a compensation signal, V COMP , to inputs 12 1A , 12 2A , 12 3A , . . . , 12 nA of PWM circuit 12 . Compensation signal V COMP is also referred to as an error signal V ERROR and appears at output 17 of error amplifier 16 .

Referring now to FIG. 3 , a timing diagram illustrating the triangular waveforms or ramp signals generated by oscillator 18 for a 4-phase power converter is illustrated. What is shown in FIG. 3 is a triangular waveform 90 having an amplitude ranging from voltage level V L90 to voltage level V H90 , a triangular waveform 92 having an amplitude ranging from voltage level V L92 to voltage level V H92 , a triangular waveform 94 having an amplitude ranging from voltage level V L94 to voltage level V H94 , and a triangular waveform 96 having an amplitude ranging from voltage level V L96 to voltage level V H96 . Triangular waveforms 90 and 92 have phase angles that are separated by 90 angular degrees; triangular waveforms 92 and 94 have phase angles that are separated by 90 angular degrees; triangular waveforms 94 and 96 have phase angles that are separated by 90 angular degrees; and triangular waveforms 96 and 90 have phase angles that are separated by 90 angular degrees. During times t 0 to t 8 , waveform 90 leads waveform 92 by 90 degrees, waveform 90 leads waveform 94 by 180 degrees, waveform 90 leads waveform 96 by 270 degrees, and compensation voltage V COMP has a substantially constant voltage value of V COMP1 . It should be noted that waveforms 90 - 96 have been shown as separate plots for the sake of clarity and that voltage level V COMP1 is the same voltage level for each of the plots for waveforms 90 - 96 .

When waveform 90 has a voltage value greater than voltage V COMP , signal 100 appearing output 14 1 of PWM 12 has a logic low voltage level, i.e., a logic 0 level. When waveform 90 has a voltage value less than voltage V COMP , signal 100 has a logic high voltage level, i.e., a logic 1 level. Similarly, when waveforms 92 - 96 have voltage values greater than voltage V COMP , signals 102 - 106 appearing at outputs 14 2 - 14 n of PWM 12 , respectively, have logic low voltage levels, i.e., logic 0 levels, and when waveforms 92 - 96 have voltage values less than voltage V COMP , signals 102 - 106 appearing at outputs 14 2 - 14 n of PWM 12 , respectively, have logic high voltage levels, i.e., logic 1 levels. Thus, signals 102 - 106 are generated by comparing compensation signal V COMP with waveforms 90 - 96 , respectively.

At time t 8 , load current I LOAD , decreases which increases voltage V OUT and causes voltage V COMP to decrease from a voltage level V COMP1 to a voltage level V COMP2 . PWM outputs 14 1 - 14 n are held low, i.e., the corresponding pulse width modulator circuits of PWM 12 are off when voltage signal V COMP is at voltage level V COMP2 . Because the corresponding pulse width modulators are off, waveforms 90 - 96 become non-time varying and have voltage levels V S90 , V S92 , V S94 , and V S96 , respectively. Therefore oscillator output signals 90 - 96 are suspended. Thus, a phase shift angle is introduced into waveforms 90 - 96 . In other words, the time during which they are suspended merely introduces a delay into waveforms 90 - 96 . Thus, at time t 8 waveform 90 begins to decrease from voltage level V H90 . However, at time t 9 , waveform 90 is suspended at a voltage level V S90 and remains at this voltage level until time t 10 at which time it continues decreasing to voltage level V L90 . Similarly, at time t 9 , waveform 92 is suspended at a voltage level V S92 and remains at this voltage level until time t 10 at which time it continues increasing to voltage level V H92 ; waveform 94 is suspended at a voltage level V S94 and remains at this voltage level until time t 10 at which time it continues increasing to voltage level V H94 ; and waveform 96 is suspended at a voltage level V S96 and remains at this voltage level until time t 10 at which time it continues decreasing to voltage level V L96 . While waveforms 90 - 96 are suspended, PWM signals 100 - 106 have a zero duty cycle, i.e., they are at logic low or logic 0 voltage levels.

It should be noted that like voltage level V COMP1 , voltage level V COMP2 is the same for each waveform 90 - 96 . Voltage levels V H90 , V H92 , V H94 , and V H96 may be the same and voltage levels V L90 , V L92 , V L94 , and V L96 are the same.

At time t 10 , output voltage V OUT begins to recover causing compensation voltage V COMP appearing at output 17 of error amplifier 16 to increase. At time t 11 , output voltage level V OUT has recovered and compensation voltage V COMP appearing at output 17 of error amplifier 16 is at voltage level V COMP1 . Thus, waveforms 90 - 96 continue from where they were suspended. During the time period between times t 10 and t 15 , signals 100 - 106 appearing at outputs 14 1 - 14 n of PWM 12 are at a logic low voltage level when waveforms 90 - 96 have voltage values greater than compensation V COMP and they are at a logic high voltage level when waveforms 90 - 96 have voltage values less than compensation voltage V COMP .

FIG. 4 is a block diagram of a multi-phase power converter 150 in accordance with another embodiment of the present invention. What is shown in FIG. 4 is PWM circuit 12 , error amplifier 16 , oscillator 18 , power stages 34 1 - 34 n , load 80 , and load capacitor 82 , which have been described with reference to FIG. 2 . Multi-phase power converter 150 further includes a dither network 152 having an output 153 coupled to input 32 of oscillator 18 via resistor 59 .

FIG. 5 is a block diagram of dither network 152 in accordance with an embodiment of the present invention. Dither network 152 may comprise operational amplifiers 154 and 162 , wherein each operational amplifier has an inverting input, a non-inverting input, and an output. A resistor 156 is connected between the output of operational amplifier 154 and its inverting input and a resistor 158 is connected between the output of operational amplifier 154 and its non-inverting input. In addition, a resistor 160 is connected between the non-inverting input of operational amplifier 154 and a source of operating potential such as, for example, V SS , and a capacitor 163 is connected between the inverting input of operational amplifier 154 and a source of operating potential such as, for example, V SS . The non-inverting input of operational amplifier 162 is connected to the inverting input of operational amplifier 154 and the output of operational amplifier 162 is connected to its inverting input in a unity gain configuration. The output of operational amplifier 162 is also connected to output 153 through a resistor 164 . Output 153 is coupled for receiving a source of operating potential such as, for example, V SS , through a resistor 168 .

In operation, dither circuit 152 changes the switching frequency of power stages 34 1 - 34 n to inhibit the load step rate and the switching frequency of switches 34 1 - 34 n , from matching for a significant period of time. This prevents a build-up of an imbalance of current in the channels.

FIG. 6 is a block diagram of a multi-phase power converter 200 in accordance with another embodiment of the present invention. What is shown in FIG. 6 is PWM circuit 12 , error amplifier 16 , oscillator 18 , power stages 34 1 - 34 n , load 80 , and load capacitor 82 , which have been described with reference to FIGS. 2 and 4 . Multi-phase power converter 200 further includes a oscillator control and dither network 202 having an output 203 coupled to input 32 of oscillator 18 through resistor 59 .

In operation, oscillator control and dither network 202 changes the switching frequency of power stages 34 1 - 34 n and introduces a phase delay to inhibit the load step rate and the switching frequency of switches 34 1 - 34 n , from matching for a significant period of time. This prevents a build-up of an imbalance of current in the channels, thereby inhibiting thermal runaway and thus thermal failure of multi-phase power converter 200 .

By now it should be appreciated that a method of balancing current in a multi-phase power converter for inhibiting thermal run-away at varying load transition rates has been provided. In accordance with an embodiment of the present invention, thermal run-away is inhibited by adjusting the frequency or period of multi-phase power converter and a load step period such that they are not coincident, the same, or similar for an extended period of time. In accordance with another embodiment, thermal run-away is inhibited by dithering the switching frequency of power stages so that the load step rate and the switching frequency of the switches do not match for a significant period of time. An advantage of the present invention is that it is cost efficient to implement.

Although certain preferred embodiments and methods have been disclosed herein, it will be apparent from the foregoing disclosure to those skilled in the art that variations and modifications of such embodiments and methods may be made without departing from the spirit and scope of the invention. For example, the method can be implemented using a digital technique. It should be noted that the word “when” is taken to mean at the time an event occurs and while the event is occurring unless otherwise stated. It is intended that the invention shall be limited only to the extent required by the appended claims and the rules and principles of applicable law.