Frequency stabilization of chopper-stabilized amplifiers

1. Field of the Invention

The present invention relates to the field of chopper stabilized amplifiers.

2. Prior Art

Frequency compensation for precision OpAmps (operational amplifiers) has a large history. FIG. 1 illustrates the nested Miller compensation scheme of U.S. Pat. No. 4,559,502 for compensation of a 3-stage amplifier (multipath hybrid nested Miller compensation is shown in U.S. Pat. No. 5,486,790). The single output stage does not need frequency compensation. The compensation capacitors C M11 and C M12 compensate for the extra gain of amplifier G 2 , while C M21 and C M22 compensate for the extra gain of amplifier G 3 .

For obtaining a low offset, choppers can be placed around the input amplifier G 3 . This is shown in FIG. 2 . The offset of amplifier G 3 will appear as a residual square wave at the output as fed back to the input of the amplifier. The average offset is nearly zero.

The chopper stage amplifier G 3 requires compensation capacitors C M21 and C M22 similar to the version without choppers of FIG. 1 . The chopper amplifier of FIG. 2 has a frequency response corresponding to the lower amplitude characteristic of FIG. 13 , but does not have chopper stabilization. The frequency response has a horizontal part with high gain at low frequencies, and a straight 6 dB/oct roll off at higher frequencies. This straight roll off is desired for providing stability with a large choice of gain settings by the feedback network. The slope of 6 dB/oct normally results in a phase margin of 90 degrees.

The situation changes with a chopper-stabilized OpAmp as in FIG. 3 . Chopper stabilization is used to compensate for the offset of the OpAmp. A chopper Ch 2 is used to sense the offset of the OpAmp and convert this into a square wave. This square-wave signal is converted into a current by voltage to current converter G 7 and redirected into a DC current by chopper Ch 4 . Next, the redirected current is fed into an integrator G 6 and then converted again in a current by voltage to current converter G 5 and subtracted from the output current of the offset producing input stage G 3 , so that its offset gets compensated. The choppers and integrator are needed to reduce the influence of the offset of the sense amplifier G 7 . This offset merely causes a small triangular voltage at the output of integrator G 6 . A reduced triangular voltage residue is present at the input of the amplifier. The larger the ratio τ=C M61 , 62 /GM 7 , where GM 7 is the transconductance of the voltage to current converter G 7 , the smaller the triangular voltage.

The frequency characteristic of the chopper stabilized amplifier of FIG. 3 differs from that of FIG. 2 in that a bubble arises in the amplitude characteristic at low frequencies, as depicted in the upper curve of FIG. 13 . The bubble is the result of the extra low-frequency gain caused by the multipath integrator G 6 . From a certain frequency f 3 back to the lower frequency f 1 , the multipath integrator G 6 has more gain than the main path, and its extra pole shows as a 12 dB/oct slope. From frequency f 1 , where the gain of the main amplifier reaches its flat maximum, there is a 6 dB/oct roll off Back to frequency f 2 . Finally, when at very low frequencies the leakage of the integration ends the integration function, the frequency characteristic becomes horizontal. If the amplifier is used in high-gain feedback settings, the 12 dB/oct part may cause undesired near-unstable behavior. Therefore, ways to straighten out this bubble are needed.

If the chopper chopper-stabilized architecture of FIG. 4 is used, (U.S. Pat. No. 6,734,723) there is the same frequency bubble as the circuit of FIG. 3 . In FIG. 4 , the chopper Ch 2 has two functions: firstly as input chopper for the main amplifier G 3 with output chopper Ch 1 , and secondly, chopper Ch 2 converts the input offset of transconductance amplifier G 3 into a square wave referred to the input. Now the offset of amplifier G 3 can be sensed by transconductance amplifier G 7 , independent of the offset of amplifier G 7 , redirected by chopper Ch 4 , and integrated by integrator G 6 and corrected for by G 5 . This results in a similar bubble of the frequency characteristic as that of FIG. 4 ,: which is shown in FIG. 13 . Therefore, there is a need to straighten the bubble out in this case also.

FIG. 5 illustrates a prior art multipath operational amplifier structure having four stages in the gain path. This gain path is hybrid nested compensated. This means that it has a backward nest C M11 and C M12 around amplifier G 1 at the output, and a forward nest C M61 and C M62 around amplifier G 6 , both encompassed by an outward nest C M51 and C M52 around amplifier G 5 . It also has a multipath amplifier G 3 in order to increase the bandwidth. This results in straight frequency roll off with 6 dB/oct if the dimensioning is done well.

FIG. 1 illustrates the prior art nested Miller compensation scheme of U.S. Pat. No. 4,559,502 for compensation of a 3-stage operational amplifier.

FIG. 2 illustrates the prior art amplifier of FIG. 1 with choppers placed around the input amplifier.

FIG. 3 illustrates another prior art chopper-stabilized operational amplifier.

FIG. 4 illustrates a prior art chopper chopper-stabilized operational amplifier architecture.

FIG. 5 illustrates a prior art multipath operational amplifier structure having four stages in the gain path.

FIG. 6 illustrates a multipath operational amplifier structure having five stages in the gain path with multipath hybrid double-nested Miller compensation.

FIG. 7 illustrates the multipath operational amplifier structure of FIG. 6 with chopper-stabilization added.

FIG. 8 illustrates a chopper chopper-stabilized amplifier with multipath hybrid double-nested Miller compensation.

FIG. 9 illustrates a chopper-stabilized current-feedback instrumentation amplifier.

FIG. 10 illustrates the chopper-stabilized current-feedback instrumentation amplifier of FIG. 9 with multipath hybrid double-nested Miller compensation.

FIG. 11 illustrates a chopper chopper-stabilized current-feedback instrumentation amplifier.

FIG. 12 illustrates the chopper chopper-stabilized current-feedback instrumentation amplifier of FIG. 11 with multipath hybrid double-nested Miller compensation added.

FIG. 13 presents curves illustrating a desired 6 dB/oct roll off in the frequency response of an amplifier and an undesired low frequency bubble in the frequency response in some amplifiers.

FIG. 14 presents curves illustrating a desired 6 dB/oct roll off in the frequency response of the amplifiers of FIGS. 7 , 8 , 10 and 12 .

FIGS. 15 through 18 illustrate circuits similar to those of FIGS. 7 , 8 , 10 and 12 in, respectively, though dispensing with the use of amplifier G 2 and Miller compensation capacitors CM 21 and CM 22 .

To straighten out the bubble in the frequency response described before, chopper-stabilized methods may be combined with the prior art multipath hybrid nested Miller compensation in two steps.

The circuit of FIG. 5 may be extended with another amplifier G 2 in the gain path. To make it stable, a double nested structure is added at the output by C M21 and C M22 . The result is the multipath hybrid double-nested Miller compensation structure of FIG. 6 . It has the desired straight 6 dB/oct roll off of the frequency characteristic.

Now chopper-stabilization can be added in the amplifier of FIG. 6 . This results in the chopper stabilized OpAmp with multipath hybrid double-nested Miller compensation of FIG. 7 . The feed forward nested capacitors C M61 and C M62 can now be chosen to fit the requirements of an integrator of the signal from amplifier G 7 in the chopper-stabilized amplifier. At a right choice of all amplifier and capacitor values, the desired straight roll off of the frequency characteristic is obtained as shown in FIG. 14 .

As an alternate embodiment, the chopper chopper-stabilized amplifier may be combined with the multipath hybrid double-nested Miller compensation as shown in FIG. 8 . This also provides the straight 6 dB roll off shown in FIG. 14 . It is interesting to see how the choppers chop the amplifiers G 3 , G 6 , and G 5 completely including the integrator feed forward nest C M61 and C M62 , without mixing up the outer nest C M51 and C M52 together with the double nest C M11 and C M12 , C M21 and C M22 at the output. This is because the outer nest C M51 and C M52 are connected before chopper Ch 4 , while in FIG. 7 , this connection was behind chopper Ch 4 .

The principles of this invention can also be applied to chopper-stabilized instrumentation amplifiers. This type of amplifier itself is the subject of a separate application for patent entitled “Chopper Chopper-Stabilized Instrumentation and Operational Amplifiers” filed on Feb. 8, 2005 as application Ser. No. 11/055123, the disclosure of which is hereby incorporated by reference.

A chopper-stabilized current-feedback instrumentation amplifier in accordance with the above referenced application is shown in FIG. 9 . Resistors R 1 and R 2 form an output attenuator for feedback of the attenuated voltage V s+ and V s− . The multipath hybrid double-nested Miller compensation structure can be combined with the amplifier of FIG. 9 by just adding C M51 and C M52 as shown in FIG. 10 . It has the desired straight 6 dB/oct roll off, as shown in the upper part of FIG. 14 .

Also the chopper chopper-stabilized current-feedback instrumentation amplifier of FIG. 11 is disclosed in the foregoing application. As disclosed therein, the voltage-to-current converter G 3 senses the input signal V in =V in+ −V in− , while the voltage-to-current converter G 4 takes the sense feedback output signal V s =V s+ −V s− . If G 3 =G 4 , the high loop gain of the whole amplifier forces the feedback sense voltage V s to be equal and opposite to the input voltage V in .

The choppers Ch 1 Ch 2 and Ch 3 chop the offset voltage of the amplifiers G 3 and G 4 . The chopped offset can be regarded as a square-wave interference voltage referred to the input voltage of amplifiers G 3 and G 4 . The input voltage V in is determined by an external source, and while generally may be a varying signal, it does not contain the square-wave signal. The high loop gain of the whole amplifier forces the feedback-sense voltage V s to compensate the chopped input offset voltage. Therefore, this square-wave chopped input offset will be superimposed on the desired feedback sense voltage V s .

In the circuit of FIG. 11 , amplifiers G 7 and G 8 (voltage to current converters or transconductance amplifiers) are used to obtain a gauge to control the offset of amplifiers G 3 and G 4 . More specifically, with respect to DC levels, the closed loop circuit settles with V in and V s being equal and opposite voltages. However the sense voltage V s has the offset caused square wave on it while V in does not. Consequently the output current of amplifier G 7 plus the output current of amplifier G 8 will simply be the square wave caused by the offset of amplifier G 3 . Thus the resulting current represents the square-wave chopped input offset voltage component from V s , and largely suppresses the desired input and feedback sense voltages.

Next the combined output currents of amplifiers G 7 and G 8 are rectified into a DC current by the chopper Ch 4 . This DC current represents the input offset voltage. Next this DC current is integrated by an integrator amplifier G 6 , with the integrator output voltage being converted into a current by G 5 and added to the output currents of amplifiers G 3 and G 4 in order to gradually cancel the input offset voltage of these amplifiers. Since the offsets are at most very slowly varying, such as by temperature or time variations, in general the response of this offset control loop need not be particularly fast, and generally is intentionally given a time constant much longer than the chopper frequency period so as to be a substantially fixed offset compensation during each chopper period. Note that the integrator has the effect of integrating the rectified square wave on the sense voltage V s , no matter how small, so that, neglecting other sources of error, the offset control loop settles when the offset is eliminated, and is not limited to the gain within the control loop.

The amplifier of FIG. 11 can be combined with the multipath hybrid double-nested Miller compensation structure to obtain a straight 6 dB/oct roll off. To that purpose two capacitors C M51 and C M52 have been added, as is shown in FIG. 12 .

In the case of either opamps or instrumentation amplifiers, autozeroing and offset correction loops may be added in accordance with the copending application set forth above. In that regard, it will be noted that the instrumentation amplifier of FIG. 10 is an extension of the opamp of FIG. 7 , and that the instrumentation amplifier of FIG. 12 is an extension of the opamp of FIG. 8 .

The embodiments of the invention disclosed herein show that it is possible to combine chopper-stabilized and chopper chopper-stabilized OpAmps and instrumentation amplifiers with multipath hybrid nested Miller compensation to provide a good high frequency compensation structure to obtain a desired straight roll off of the frequency characteristic with 6 dB/oct. The exemplary embodiments are described with respect to differential amplifiers, though may be realized as single ended amplifiers also, that is, as single input, single output amplifiers. Also in the embodiments disclosed, two output stages are shown, though in some cases, such as in the case of amplifiers that are lightly loaded, a single stage may be used, dispensing with the use of amplifier G 2 and Miller compensation capacitors CM 21 and CM 22 . This is illustrated for the circuits of FIGS. 7 , 8 , 10 and 12 in FIGS. 15 through 18 , respectively. Also amplifier G 5 may be an attenuator, either an amplifier with a gain of less than one, or simply resistors for converting the integrator output to a current for input to the current summing point or for attenuation. Additional Miller compensated, nested amplifiers may also be incorporated as desired. Thus while certain preferred embodiments of the present invention have been disclosed and described herein for purposes of illustration and not for purposes of limitation, it will be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention.