Amplitude control device for electrical oscillator and electrical oscillator comprising such a device

The present invention relates to an amplitude control device for an electrical oscillator, and an electrical oscillator comprising such a device.

The present invention is applicable to radio communication systems, and more particularly, to mobile telephone systems, such as GSM, DCS, PCS, UMTS, etc.

An electrical oscillator is a circuit that produces an electrical signal at a frequency defined by a time constant specific to it. FIG. 1 illustrates a principle diagram for an electrical oscillator that includes an inductance L, a capacitor C and a holding amplifier A. The oscillator time constant is equal to √{square root over (LC)} and the oscillation frequency is equal to ½π√{square root over (LC)}.

The amplitude of voltage oscillations measured at the terminals of the LC circuit is determined by non-linearities of the holding amplifier. It is important to check the amplitude of the voltage oscillations to achieve correct interfacing between the oscillator and the oscillator load circuits that receive the oscillation voltage on their inputs.

If the oscillator is integrated on silicon, manufacturing parameter dispersions affect the value of the amplitude of oscillations. For example, this is the case for the resistivity of metals for which the dispersions (+/−10%) modify the quality coefficient of the inductances and the capacitors, or the resistivity of poly-crystalline silicon for which the dispersions (+/−20%) modify the oscillator biasing current.

One known way of reducing these dispersions is to make amplitude slaving circuits that apply a retroaction on the oscillator biasing current. For example, this type of slaving circuit is described in the article titled “A 2V 2.5 GHz-104 dB/Hz At 100 kHz Fully Integrated VCO Wideband Low Noise Automatic Amplitude Control” (Alfio Zanchi et al, IEEE JSSC, VOL 36, No. 4, pp. 611-619, April 2001), and in the article titled “A Low Noise Low Power VCO With Automatic Amplitude Control For Wireless Application” (M. A. Margarit et al., IEEE JSSC, VOL 34, No. 6, pp. 761-771, June 1999).

An example of an oscillator with its slaving circuit according to known art is shown in FIG. 2 . The circuit in FIG. 2 comprises an oscillator 1 , a rectification stage 2 and a differential amplifier 3 . The oscillator 1 comprises two transistors Q 1 , Q 2 , two resistances R 1 , R 2 , two inductances L 1 , L 2 , three capacitors C 1 , C 2 and C 3 , and a current generator G. Three biasing voltages VBB, VCC, VEE power the oscillator 1 . The output voltage from the oscillator 1 is taken from the terminals of capacitor C 1 .

The rectification stage 2 receives the output voltage from the oscillator on its input and detects the peak level of the oscillation voltage, for example, by double alternation rectification. The output voltage from the rectification stage is applied to a first input to the differential amplifier 3 . A reference voltage Vref is applied to the second input of the differential amplifier 3 . The output signal from the differential amplifier 3 controls the amplitude of the biasing current that passes through the current generator G. Thus, the amplitude of the output voltage from the oscillator 1 is controlled by the level of the reference voltage Vref.

This type of circuit has the disadvantage that it copies all the noise of the reference voltage Vref for frequencies within the slaving passband, and that there is no noise control outside this passband. This copying affects the phase noise in addition to the amplitude noise. Ideally, amplitude regulation should only generate an amplitude modulation. However, the non-linear nature and parametric effects of the oscillator cause the amplitude noise to be converted to phase noise, thus deteriorating the spectral quality of the oscillation signal.

The invention relates to an amplitude control device for a signal output by an oscillator. The amplitude control device comprises slaving means comprising rectification means to rectify the signal output by the oscillator, and differential amplification means to form a control signal for the oscillator biasing current starting from the signal taken at the output from the rectification means and a reference voltage.

The slaving means further includes a divider bridge to form a first fraction of the reference voltage and a second fraction of the reference voltage starting from the reference voltage. The first fraction of the reference voltage is applied to a first input of the differential amplification means. An adder adds the signal output by the rectification means and the second fraction of the reference voltage. The signal output by the adder is applied to a second input to the differential amplification means.

Advantageously, the invention considerably reduces the contribution of the reference voltage noise to the oscillator phase noise. Only a fraction of the reference voltage is used to define the amplitude of the oscillations. Consequently, for frequencies less than the amplitude slaving passband, the noise injected into the oscillator is proportional to the product of this fraction by the reference voltage noise.

According to one improvement to the invention, the device controlling the amplitude of the signal output by an oscillator comprises additional slaving means to control the oscillator noise level for frequencies greater than frequencies within the passband of the slaving means.

The invention also relates to an electrical oscillator characterized in that it comprises a signal amplitude control device according to the invention.

Other characteristics and advantages of the invention will become clear after reading a preferred embodiment of the invention with reference to the attached figures among which:

FIG. 1 is a schematic diagram of an electrical oscillator according to the prior art;

FIG. 2 is a schematic diagram of an electrical oscillator with an oscillation amplitude slaving circuit according to the prior art;

FIG. 3 is a schematic diagram of an electrical oscillator with an oscillation amplitude slaving circuit according to the present invention;

FIG. 4 is a more detailed schematic diagram of the electrical oscillator shown in FIG. 3 ;

FIG. 5 is a schematic diagram of an electrical oscillator with an improvement to the circuit shown in FIG. 3 ; and

FIG. 6 is a more detailed schematic diagram of the electrical oscillator shown in FIG. 5 .

The same references denote the same elements on all figures. FIGS. 1 and 2 have already been described, therefore these figures will not be described again. FIG. 3 shows an electrical oscillator with an amplitude slaving circuit according to the invention. The circuit in FIG. 3 comprises an oscillator 1 and an amplitude slaving circuit 4 . The oscillator 1 is identical to the oscillator shown in FIG. 2 . The amplitude slaving circuit 4 comprises a voltage divider bridge 5 , a rectification stage 6 , an adder 7 and a differential amplifier 8 .

The input terminals to the amplitude slaving circuit 4 are the input terminals to the rectification stage 6 . The amplitude slaving circuit 4 output is the output from the differential amplifier 8 . A first capacitor C 4 is between the first output terminal of the oscillator 1 and a first input terminal to the rectification stage 6 . Similarly, a second capacitor C 5 is between a second output terminal of the oscillator 1 and a second output terminal from oscillator 1 .

The voltage divider bridge 5 comprises three resistances R 8 , R 9 , R 10 mounted in series and a decoupling capacitor C 9 is mounted in parallel with resistances R 9 and R 10 . The reference voltage Vref is applied to a first terminal of the resistance R 8 , the second terminal of which is connected to a first input (+) of the differential amplifier 8 and to the first terminal of the resistance R 9 The second terminal of the resistance R 9 is connected to a first input of adder 7 and to a first terminal of resistance R 10 . The second terminal of resistance R 10 is connected to a power supply voltage VEE.

The output from the rectification stage 6 is connected to a second input of an adder 7 , the output of which is connected to the second input (−) of the differential amplifier 8 . The output signal from the differential amplifier 8 forms the amplitude control signal for biasing the current that passes through the current generator G.

FIG. 4 illustrates an example electrical oscillator like that shown in FIG. 3 . Apart from the divider bridge 5 , the amplitude slaving circuit 4 comprises two P-type MOS (Metal Oxide Semiconductor) transistors MP 1 and MP 2 , three bipolar transistors Q 4 , Q 5 and Q 6 , two decoupling capacitors C 7 , C 8 and three resistances R 5 , R 6 and R 7 . The transistors Q 4 and Q 5 perform the rectification. The differential amplifier 8 is composed of transistors MP 1 , MP 2 , Q 4 , Q 5 , Q 6 and the resistance R 7 . The participation of transistors Q 4 and Q 5 in the rectification and differential amplification functions has the advantage of limiting the number of stages in the amplitude slaving circuit 4 , and consequently, obtaining better noise performances of this circuit.

Transistors Q 4 and Q 5 are mounted on a common emitter. The common emitter of transistors Q 4 and Q 5 is connected to a first terminal of the resistance R 7 , the second terminal of which is connected to a power supply voltage VEE. The decoupling capacitor C 8 is parallel with the resistance R 7 .

The bases of transistors Q 4 and Q 5 form the inputs to the rectification circuit 6 and are connected to a first terminal of the resistance R 5 , and to a first terminal of resistance R 6 . The second terminals of the resistances R 5 and R 6 are both connected to a first terminal of capacitor C 7 . The second terminal of the capacitor C 7 is connected to the power supply voltage VEE.

The collectors of transistors Q 4 and Q 5 are connected to each other and form the output from the differential amplifier. The collectors of Q 4 and Q 5 are connected to the source of the transistor MP 1 , and the drain of this transistor is connected to the biasing voltage VCC. The collector of transistor Q 6 is connected to the source of transistor MP 2 , the drain of which is connected to the biasing voltage VCC. The gates of transistors MP 1 and MP 2 are connected to each other and to the source of transistor MP 2 . The emitter of transistor Q 6 is connected to the emitters of transistors Q 4 and Q 5 . The base of transistor Q 6 is connected to the second terminal of resistance R 8 . The second terminal of resistance R 9 is connected to the second terminals of resistances R 5 and R 6 .

The current generator G includes a resistance R 4 in series with a N-type MOS transistor MN 1 . The source terminal of the transistor MN 1 is connected to the DC voltage VEE. The output voltage from the slaving circuit, which is picked up on the collector common to transistors Q 4 and Q 5 , is applied to the gate of transistor MN 1 . A capacitor C 6 is between the grid of transistor MN 1 and the voltage VEE.

In the remainder of the description, the voltages U 1 , U 2 , U 3 , U 4 , U 5 and U 6 represent the following potential differences: the potential difference taken between the node common to resistances R 8 and R 9 and the circuit ground; the potential difference taken between the node common to resistances R 9 and R 10 and the circuit ground; the potential difference taken between the emitters common to transistors Q 4 , Q 5 and Q 6 and the circuit ground; the potential difference taken between the first input to the slaving circuit 4 and the circuit ground; the potential difference taken between the second input to the slaving circuit 4 and the circuit ground; and the potential difference taken between the output from the slaving circuit 4 and the circuit ground.

When the oscillator has not yet started, the voltages U 4 and U 5 are equal to U 2 and the differential amplifier composed of Q 4 , Q 5 , Q 6 , R 7 , MP 1 and MP 2 is unbalanced since the voltage U 1 is greater than U 4 and U 5 . Consequently, the voltage U 6 is close to VCC. The transistor MN 1 then has a minimum resistance Ron (MN 1 ) which creates a maximum biasing current for the oscillator so that oscillations can start.

The amplitude of the oscillations then increases gradually, thus superposing two alternating voltages with opposite phases onto the DC component of the voltages U 4 and U 5 such that when a positive half-alternation of the voltage U 4 takes place, and transistor Q 5 is blocked, transistor Q 4 outputs current Ie(Q 4 ) such that:

Ie ⁡ ( Q4 ) = Is · ⅇ · U4 - U3 P

where P is the thermodynamic potential.

The situation is reversed during a positive half-alternation of U 5 . The currents that pass through the emitters of transistors Q 4 and Q 5 are summed and filtered by the capacitor C 8 , and the voltage U 3 is an image of the average value of these currents. When this average value of the current is equal to the DC current output by the transistor Q 6 , the following is true:

< Ie ⁡ ( Q4 ) >= 2 T ⁢ ∫ 0 T 2 ⁢ Is · ⅇ ⁢ U4 - U3 P ⁢   ⁢ ⅆ t , Ie ⁡ ( Q6 ) = Is · ⅇ ⁢ U1 - U3 P ,

where p is the thermodynamic potential, <Ie(Q 4 )> is the average value of the current Ie(Q 4 ), and T is the period of the oscillation signal.

U4 + U2 + Um · sin ⁡ ( 2 · π · t T ) ,

where Um is the amplitude of the signal present on one of the oscillator output nodes. If the filtering done by the capacitor C 8 is sufficiently efficient, it can be assumed that the voltage U 3 is constant and we then have:

2 T ⁢ ∫ 0 T 2 ⁢ Is · ⅇ U2 + Um · sin ⁡ ( 2 · π · t T ) P ⁢   ⁢ ⅆ t = Is · ⅇ · U1 P namely :   ⁢ 2 T ⁢ ∫ 0 T 2 ⁢ ⅇ Um · sin ⁡ ( 2 · π · t T ) P ⁢ ⅆ t = ⅇ U1 - U2 P

if we make a variable substitution

x = 2 · π · t T , we ⁢   ⁢ obtain :   ⁢ ln ⁡ [ ∫ 0 π ⁢ ⅇ Um · sin ⁡ ( x ) P ⁢   ⁢ ⅆ x ] = U1 - U2 P [ 3 ]

Equation 3 shows that the amplitude Um only depends on the difference (U 1 −U 2 ) that is a fraction of the reference voltage Vref. We have:

( U1 - U2 = R9 R8 + R9 + R10 ⁢   ⁢ VREF )

Thus, the noise of the reference voltage Vref injected into the slaving is also multiplied by only this fraction of the reference voltage, which reduces its contribution to the oscillator phase noise.

It may be noted that the circuit according to the invention can be used to adapt the noise quantity injected by the reference voltage to the amplitude of the oscillations. Equation [3] shows that a reduction in the reference voltage Vref causes a reduction in the quantity U 1 −U 2 , and consequently a reduction in the amplitude Um of the oscillations. The oscillators thus made are only slightly sensitive to the reference voltage noise.

However, for frequencies greater than the frequencies of the passband of the slaving circuit 4 , there is some phase noise that returns from the oscillator. This noise return phenomenon is particularly visible if the transistor MN 1 is operating in its saturated zone. It is then necessary to guarantee that MN 1 operates in its pure resistance zone. This condition is satisfied by the use of an additional slaving circuit.

FIG. 5 shows a slaving device comprising such an additional slaving circuit. FIG. 6 shows an example embodiment of the slaving device shown in FIG. 5 . With reference to FIG. 5 , the additional slaving circuit 9 controls the oscillator noise level for frequencies greater than the frequencies within the passband of the slaving circuit 4 .

The additional slaving circuit 9 is composed of a differential amplifier, a first input of which is connected to the emitters of transistors Q 1 and Q 2 , and a second input of which is connected to a DC voltage VXX. The output of the differential amplifier is connected to the terminal common to resistances R 1 and R 2 of the oscillator 1 , which is therefore no longer connected to a DC voltage VBB as is the case according to the prior art.

According to the example shown in FIG. 6 , the circuit 9 comprises a resistance R 3 and a transistor Q 3 . The first input, the second input and the output from the differential amplifier respectively correspond to the base, the emitter and the collector of transistor Q 3 . The emitter of transistor Q 3 is connected to the power supply voltage VEE (and then VXX=VEE), and the collector of transistor Q 3 is connected to a first terminal of the resistance R 3 , the second terminal of which is connected to the power supply voltage VCC. The resistance R 3 fixes the biasing current of the second slaving. The resistance R 4 of the current generator is chosen such that the transistor MN 1 operates in its pure resistance zone. The following relation must then be satisfied:

Vbe(Q 3 )−R 4 .Iosc>Vdss(MN 1 ), where

Vbe(Q 3 ), Iosc and Vdss(MN 1 ) respectively represent the voltage on the base of transistor Q 3 , the oscillator biasing current, and the drain/source saturation voltage of transistor MN 1 .

It may also be noted that the additional slaving circuit contributes to reducing the phase noise of R 1 , R 2 , Q 1 and Q 2 for frequencies less than its passband. This statement is particularly relevant if a CMOS type technology is used (HCMOS9 type), if transistors Q 1 and Q 2 are replaced by NMOS transistors, and if Q 3 is a native NPN transistor based on this technology. Under these conditions, the contribution of low frequency noise from NMOS transistors to the oscillator phase noise is almost entirely canceled.

This invention is applicable to oscillators in general, and particularly to voltage controlled oscillators (VCO).