Synchronous-rectified DC to DC converter with improved current sensing

This application claims the benefit of U.S. Provisional Patent Application Ser. No. 60/151,826, filed Sep. 1, 1999. Notice: More than one reissue application has been filed for the reissue of Pat. No. 6,246,220. The reissue applications are reissue application No. 10/044,506 ( the parent reissue ) and reissue application No. 10 / 282 , 753 ( the present, continuation reissue application ). Both Reissue applications are reissues of the same U.S. Pat. No. 6,246,220.

This reissue Application claims the benefit of U.S. Provisional Patent Application Ser. No. 60 / 151 , 826 , filed Sep. 1 , 1999 .

A synchronous buck DC to DC converter typically employs a pair of switches arranged to connect one end of an inductor to either an input supply voltage or to ground. The second end of the inductor is attached to a load. It is well known to use field effect transistors (FET's) as these switches. Load current flows from the supply through the upper FET and the inductor while the FET is on, and from ground through the lower FET and the inductor while that FET is on.

It is desirable to sense the value of the load current to perform various functions such as, for example, to deliberately decrease the output voltage as load current increases (i.e., output voltage “droop”), to provide for current limiting or over-current trip to protect the load and the converter components, and in order to balance the output current being sourced by each channel in a multi-channel or multi-phase converter. The load current can be sensed through determining the DC resistor of the inductor and sensing the voltage drop across that DC resistance, or by sensing the voltage drop across an added series sense resistor. The load current can also be detected by sensing the voltage drop caused by the load current flowing through the upper FET switch. However, each of these methods has their disadvantages. Sensing the load current by using the DC resistance of the inductor requires adding an R-C filter across the inductor to remove the AC component of the current. Thus, additional components are required and extra cost incurred. Adding a series sense resistor also requires an extra component, increase cost, and reduces system efficiency. Furthermore, sensing the voltage drop across the drain-to-source resistance of the upper FET when it is conducting has often proven to be impractical, since the “on” time of that switch is typically very short.

Therefore, what is needed in the art is a DC/DC converter with improved current sensing. Furthermore, what is needed in the art is an apparatus and method which enables the sensing of load current in a DC/DC converter by sensing the voltage drop across the drain-to-source resistance of a switching FET.

Moreover, what is needed in the art is an apparatus and method which enables sensing and detection of overcurrent in a DC/DC converter.

The present invention provides a power supply with improved current sensing.

The invention comprises, in one form thereof, a DC to DC buck pulse width modulator converter circuit having an input, a high side output and a low side output. A high side switch is electrically connected between a common output node and a voltage supply, and controls a flow of current therethrough dependent upon the high side output. A low side switch is electrically connected between the common output node and ground, and controls a flow of current therethrough dependent upon the low side output. A virtual ground amplifier includes a second input electrically connected to ground. A current feedback resistor is electrically connected intermediate the common output node and a first input of the virtual ground amplifier. A variable impedance component is electrically connected to an output of the virtual ground amplifier and to the first input of the virtual ground amplifier. The impedance of the variable impedance component is varied dependent upon the output of the virtual ground amplifier. A sample and hold circuit is electrically connected intermediate the input of the pulse width modulator converter circuit and the variable impedance component. The sample and hold circuit sources a virtual ground current through the variable impedance component, and samples the virtual ground current.

An advantage of the DC/DC converter or the present invention is that it provides an improved method and apparatus to measure the voltage drop across the drain-to-source resistance of a FET having a very brief “on” time.

Another advantage of the DC/DC converter of the present invention is that the amount of droop in the output voltage in response to a change in load current is easily manipulated and scaled by selecting an appropriate value for the voltage feedback resistor.

Yet another advantage of the DC/DC converter of the present invention is that the sensitivity or magnitude of the current limiting or trip is easily manipulated or scaled by selecting an appropriate value for the voltage feedback resistor.

A still further advantage of the DC/DC converter of the present invention is that a broad range of load current and component values is accommodated by selecting an appropriate value for the current feedback resistor.

The above-mentioned and other features and advantages of this invention, and the manner of attaining them, will become apparent and be better understood by reference to the following description of embodiments of the invention in conjunction with the accompanying drawings, wherein:

FIG. 1 is a high-level schematic and functional drawing of one embodiment of the DC/DC converter of the present invention;

FIG. 2 is a detailed schematic and functional diagram of the DC/DC converter of FIG. 1 ;

FIG. 3a and 3b are timing diagrams illustrating the operation of the DC/DC converter of FIG. 2 ;

FIG. 4 is a diagram of a node voltage versus load current of the DC/DC converter of FIG. 2 ;

FIG. 5 is a schematic of a negative current source for use with the DC/DC converter of FIG. 2 ; and

FIG. 6 is a detailed schematic of a second embodiment of a DC/DC converter of the present invention.

Corresponding reference characters indicate corresponding parts throughout the several views. The exemplifications set out herein illustrate one preferred embodiment of the invention, in one form, and such exemplifications are not to be construed as limiting the scope of the invention in any manner.

Referring now to the drawings, and particularly to FIG. 1 , there is shown one embodiment of a DC/DC converter of the present invention. DC/DC converter 10 includes low side field effect transistor (FET) 12 and high side FET 14 . The drain of low side FET 12 is electrically connected to the source of high side FET 14 at common output node 16 . The drain of high side FET 14 is connected to power supply 18 . The source of low side FET 12 is electrically connected to ground. Each gate terminal of low side FET 12 and high side FET 14 is electrically connected to a respective output (not referenced) of buck converter circuit 20 . Inductor 24 and current feedback resistor 26 are each electrically connected to common output node 16 , and thus to the drain of FET 12 and source of FET 14 . More particularly, inductor 24 is electrically connected between common output node 16 and load reservoir capacitor 28 , and current feedback resistor 26 is electrically interconnected between common output node 16 and virtual ground circuit node 30 . A load 32 , schematically represented as a resistor, is electrically connected in parallel with load reservoir capacitor 28 .

Virtual ground amplifier 34 has its inverting input 34 a electrically connected to virtual ground circuit node 30 and its non-inverting input 34 b connected to ground. Output 34 c of virtual ground amplifier 34 is electrically connected to and drives the gate of FET 36 . The source of FET 36 is electrically connected to virtual ground circuit node 30 . The drain of FET 36 is electrically connected to sample and hold circuit 38 . Thus, as will be apparent to one skilled in the art, virtual ground amplifier 34 and FET 36 are configured to continuously drive virtual ground circuit node 30 toward ground potential. With virtual ground circuit node 30 being continuously driven towards ground potential, the end of current feedback resistor 26 that is connected to circuit node 30 will be at ground potential and the end connected to common output node 16 will have a negative voltage. This negative voltage at the end of current feedback resistor 26 that is connected to common output node 16 will be equal to the product of output current I OUT and the on-state resistance that exists between the drain and source (RDS ON ) of low side FET 12 . Current I SENSE flows through current feedback resistor 26 and has a magnitude determined by the ratio of RDS ON of low side FET 12 to the value of current feedback resistor 26 . Thus, I SENSE is the product of output current I OUT and the ratio of RDS ON of low side FET 12 to current feedback resistor 26 , and as such is representative of output current I OUT . Load current I L is the current flowing through inductor 24 and is substantially equal to output current I OUT minus I SENSE . Typically, since the ratio of RDS ON to the value of current feedback resistor 26 is relatively small, I SENSE is substantially smaller than output current I OUT . Therefore, output current I OUT and load current I L will be of substantially similar magnitudes and thus I SENSE will also be representative of load current I L .

The value of current feedback resistor 26 is selected to provide a convenient value of current flow for the values of load current I L and/or the value of RDS ON of low side FET 12 . Thus, the sensitivity or magnitude of, for example, the voltage droop, current limiting or trip, and current balancing incorporated into DC/DC converter 10 is scaled by selecting the value of current feedback resistor 26 relative to the value of RDS ON of low side FET 12 . Furthermore, the voltage drop across RDS ON of low side FET 12 , which is usually negative, is accommodated in DC/DC converter 10 without the need for a negative voltage supply.

Referring now to FIG. 2 , system control circuit 40 is electrically connected to sample and hold circuit 38 . As stated hereinabove, the drain of FET 36 connects the sample and hold circuit 38 . The current supplied by the source of FET 36 flows from sample and hold circuit 38 into the drain of FET 36 , out of the source of FET 36 , and into virtual ground circuit node 30 . Also flowing into virtual ground circuit node 30 , from the opposite direction, is I SENSE which, as stated above, is representative of load current I L . In order to hold virtual ground circuit node 30 at ground potential, virtual ground amplifier 34 , via output 34 c, adjusts the current flowing through FET 36 and into virtual ground circuit node 30 to be substantially equal to I L SENSE . Since I SENSE is representative of the load current I L , the current flowing through FET 36 and into virtual ground circuit node 30 , as controlled by virtual ground amplifier 34 and FET 36 , is also representative of load current I L . System control circuit 40 periodically issues control signal 40 a to sample and hold circuit 38 . Control signal 40 a is issued when FET 36 is in the on or conducting condition. In response to control signal 40 a, sample and hold circuit 38 samples the current flowing through FET 36 when FET 36 is in the on condition and holds the sampled value. Thus, the sample value acquired by sample and hold circuit 38 is also representative of load current I L . Sample and hold circuit 38 issues sample signal 38 a which is representative of the sampled value of current flowing through FET 36 .

DC/DC converter 10 monitors the voltage V OUT across load 32 through voltage feedback resistor 44 . Voltage feedback resistor 44 is connected at one end to load 32 and at the other end to inverting input 46 a of error amplifier 46 . V FB is the voltage across voltage feedback resistor 44 . The non-inverting input 46 b of error amplifier 46 is electrically connected to reference voltage supply 48 , which provides a predetermined voltage that is substantially equal to the desired output voltage of DC/DC converter 10 . Error amplifier 46 regulates the voltage of inverting input 46 a to be substantially equal to the voltage from reference voltage supply 48 . Since the voltage at inverting input 46 a is substantially equal to the sum of V OUT and V FB , error amplifier 46 acts to regulate the sum of V OUT and V FB to be substantially equal to the voltage from reference voltage supply 48 . Output 46 c of error amplifier 46 is electrically connected to compensation circuit node 50 . A feedback path between output 46 c and inverting input 46 a of error amplifier 46 includes compensation resistor 52 and compensation capacitor 54 . More particularly, connected to compensation circuit node 50 is one end of compensation capacitor 54 which, in turn, is connected at its other end to compensation resistor 52 . Compensation resistor 52 , at the end thereof opposite to compensation capacitor 54 , is connected to summing node 56 . Compensation resistor 52 and capacitor 54 in the voltage feedback path provide system stability and control system response.

Sample signal 38 a, which is issued by sample and hold circuit 38 and is representative of load current I L , is also connected to the inverting input of error amplifier 46 . There is no other path for direct current at inverting input 46 a of error amplifier 46 except through voltage feedback resistor 44 . Thus, the voltage across voltage feedback resistor 44 , i.e., V FB , is modified by signal 38 a. As stated above, error amplifier 46 regulates the voltage at its inverting input 46 a, which is equal to the sum of V OUT and V FB , to be substantially equal to the reference voltage supply 48 . Thus, for example, as sampling signal 38 a increases, V FB increases proportionally and error amplifier 46 reduces V OUT to maintain the voltage at inverting input 46 a to be equal to reference voltage supply 48 . Since sampling signal 38 a is representative of load current I L , V OUT is in effect modulated in an inversely proportional manner relative to load current I L . Thus, as shown in FIG. 4 , V OUT is varied or droops dependent at least in part upon load current I L . By selecting the value of feedback resistor 44 , the amount of variation or droop in V OUT relative to load current I L is controlled.

Inverting input 58 a of comparator 58 is electrically connected to sawtooth generator 60 , and receives therefrom a sawtooth waveform having predetermined characteristics. Output 58 c of comparator 58 is electrically connected to set-rest (SR) latch 62 . Output 62 a of SR latch 62 is electrically connected to and buffered by driver 64 which, in turn, drives low side FET 12 and high side FET 14 . DC/DC converter 10 is configured, for example, such that a high-level signal at output 62 a of SR latch 62 turns low side FET 12 off and turns on high side FET 14 . Sawtooth generator 60 receives sync pulse 66 from system control circuit 40 . SR latch 62 also receives sync pulse 66 .

Error amplifier 46 produces at output 46 c a signal that is representative of the actual output voltage V OUT relative to, such as, for example, subtracted from or added to, the voltage of reference voltage supply 48 , which represents the desired output voltage of DC/DC converter 10 . For example, output 46 c of error amplifier 46 produces a signal that is more negative, or increases in a negative direction, as V OUT increases above the voltage of reference voltage supply 48 . Conversely, and as a further example, error amplifier 46 produces at output 46 c a signal having a decreasingly negative magnitude (i.e., a more positive magnitude) as V OUT decreases below the voltage of reference voltage supply 48 . Output 46 c of error amplifier 46 is electrically connected to the non-inverting input of comparator 58 . Comparator 58 compares the sawtooth waveform electrically connected to its inverting input 58 a with output 46 c of error amplifier 46 which is electrically connected to its noninverting input 58 b. Output 58 c of comparator 58 is active, such as, for example, high during the time that the sawtooth waveform generated by sawtooth generator 60 is less positive than output 46 c of error amplifier 46 .

Referring to FIG. 3a , the condition of output voltage V OUT being less than the desired output voltage, or less than the voltage of reference voltage supply 48 , is illustrated. Thus, output 46 c of error amplifier 46 is relatively high, thereby placing a relatively high signal at noninverting input 58 b of comparator 58 . At least a substantial portion of the period of the sawtooth waveform will be less positive than the relatively high-level signal present at noninverting input 58 b. Output 58 c of comparator 58 is active, such as, for example, high, during that substantial portion of the period for which the sawtooth waveform has a value that is less positive than the relatively high signal present at noninverting input 58 b. Thus, the pulse width of output 58 c will be relatively wide, or alternatively the active period of output 58 c will be relatively long in duration, when V OUT is less than the voltage of reference voltage supply 48 .

Conversely, and with particular reference to FIG. 3b output 46 c of error amplifier 46 is relatively low when V OUT is greater than the voltage of reference voltage supply 48 . This condition places a relatively low-level signal at noninverting input 58 a of comparator 58 . A relatively small portion of the period of the sawtooth waveform will be less positive than the relatively low-level signal present at noninverting input 58 b. Output 58 c of comparator 58 will be active during only that relatively small portion, if any, of the period of the sawtooth waveform (e.g., the lowest points or bottom peaks) which is less positive than the relatively low signal at noninverting input 58 b. Thus, the pulse width of output 58 c will be relatively narrow, or alternatively the active period of output 58 c will be relatively short in duration, when V OUT is greater than the voltage of reference voltage supply 48 .

When output 58 c is active, output 62 a of SR latch 62 is set, such as, for example, high. Conversely, when output 58 c is not active, output 62 a of SR latch 62 is reset, such as, for example, low. Thus, when the sawtooth waveform is more positive than the voltage level of reference voltage supply 48 , output 62 a of SR latch 62 is reset, i.e., low. Output 62 a of SR latch 62 is set, i.e., high, when the sawtooth waveform drops below the predetermined voltage. Output 62 a of SR latch 62 is electrically connected to and buffered by driver 64 which, in turn, drives low side FET 12 and high side FET 14 . DC/DC converter 10 is configured such that, for example, a high or set condition on output 62 a of SR latch 62 results in driver 64 turning off low side FET 12 and turning on high side FET 14 .

Current tripping or over current protection is provided by overcurrent detection circuit 70 . Overcurrent detection circuit 70 compares the sample signal 38 a to a reference current (not shown) and issues overnight signal 70 a to system control circuit 40 when sample signal 38 a exceeds the reference current. System control 40 responds to overcurrent signal 70 a by shutting down DC/DC converter 10 . System control 40 is configured, for example, to restart the operation of DC/DC converter 10 after a predetermined amount of time.

Negative current source 72 is electrically connected intermediate system control 40 and virtual ground circuit node 30 . Load current I L becomes negative under certain operating conditions, such as, for example, when load current I L has a low average value and the sawtooth waveform created due to the switching of voltage across inductor 24 dips to a negative value. During such operating conditions, i.e., when I L is negative, the voltage at the drain of low side FET 12 is positive. The positive voltage on the drain of low side FET 12 results in the sourcing of current through resistor 26 and into virtual ground circuit node 30 , thereby driving virtual ground circuit node 30 to a positive potential. Negative current source 72 sources I PULL DOWN into virtual ground circuit node 30 in response to signal 40 N , and thereby maintains virtual ground node 30 at ground potential under the conditions when I L is negative. Thus, virtual ground amplifier 46 , variable impedance component 36 and sample and hold circuit 38 are not required to operate in a bi-directional manner (i.e., they source current is one direction only) and the need to include a negative voltage supply in DC/DC converter 10 is eliminated.

As best shown in FIG. 5 , negative current source 72 includes switches 80 , 82 and 84 . Each of switches 80 , 82 and 84 are, for example, MOS transistors. Current source 86 is a pull down current source, such as, for example, an NMOS mirror, and is electrically connected intermediate ground and node 90 . Switch 80 is electrically connected intermediate node 90 and voltage supply 88 , and selectively connects node 90 to voltage supply 88 . Capacitor 92 is electrically interconnected between node 90 and node 94 . Each of switch 82 and 84 have a first side electrically connected to node 94 . The other side of switch 82 is electrically connected to ground, while the other side of switch 84 is electrically connected to virtual ground circuit node 30 . Switches 80 and 82 are closed and switch 84 is open when the reverse current sourced by current source 86 is not required to maintain virtual ground circuit node 30 at ground potential, such as, for example, when low side FET 12 is off. The supply voltage of voltage supply 88 is thus stored across capacitor 92 , with node 90 having a positive potential and node 94 having a negative potential. In order to source pull down current from current source 86 , switches 80 and 82 are each opened and switch 84 is closed. Thus, I PULL DOWN flows into virtual ground node 30 in the same direction as normal forward current induced by the voltage drop on low side FET 12 . The addition of current I PULL DOWN maintains virtual ground circuit node 30 at ground potential, and is optionally subtracted out later so as not to affect subsequent circuit operation, such as, for example, the current limit trip point.

In use, and with continued reference to FIGS. 3a and 3b , the sequence of operation DC/DC converter 10 is as follows. Sawtooth generator 60 receives sync pulse 66 from system control circuit 40 . SR latch 62 also receives sync pulse 66 . Sync pulse resets both the sawtooth waveform and output 62 a of SR latch 62 to low levels. SR latch 62 is configured to reset output 62 a based upon sync pulse 66 , regardless of the condition or state of the output of comparator 58 . Thus, if the output of comparator 58 is, for example, continuously higher than the sawtooth waveform, output 62 a of SR latch 62 will be low during a high level of sync pulse 66 . As shown in FIGS. 3a and 3b at points 200 a and 200 b, respectively, sync pulse 66 resets the sawtooth waveform generated by sawtooth generator 60 to a low level, and resets output 62 a of SR latch 62 . DC/DC converter 10 is configured such that, for example, when output 62 a of SR latch 62 is low, high side FET 14 is off and low side FET 12 is on. Thus, the resetting of output 62 a of SR latch 62 by sync pulse 66 turns on low side FET 12 . During this time period, i.e., when low side FET 12 is on, RDS ON of low side FET 12 is measured. At the training edge of sync pulse 66 , at points 210 a and 210 b, respectively, the sawtooth waveform begins to slope downward (i.e. has a negative slope).

Referring now particularly to FIG. 3a , the condition of DC/DC converter 10 having an output voltage V OUT that is lower than the desired or target level is illustrated. Thus, the voltage across load 32 is lower than desired. This condition results in output 46 c of error a amplifier 46 having a high level relative to the sawtooth waveform. At point 300 a, the leading, or positively sloped, edge of the sawtooth waveform crosses above the output level of output 46 c of error amplifier 46 , thereby sending output 58 c of comparator 58 low. This particular transition in output 58 c does not affect output 62 a of SR latch 62 since sync pulse 66 is still active, and thus output 62 a remains reset or low.

At point 310 a, the trailing, or negatively sloped, edge of the sawtooth waveform crosses below the output level of output 46 c of error amplifier 46 , thereby sending output 58 c of comparator 58 high. This transition in output 58 c to a high level, in turn, sets output 62 a of SR latch 62 high thereby turning high side FET 14 on and turning off low side FET 12 . The high level of output 46 c relative to the sawtooth waveform results in the sawtooth waveform dropping below the level of output 46 c (at point 310 a) relatively early in the period of the sawtooth waveform. Thus, points 300 a and 310 a are relatively close in time, and, therefore, the period of time during which low side FET 12 is off is corresponding brief. Conversely, the period of time during which high side FET 14 is on and sourcing current is relatively long. Thus, high side FET 14 is on for a relatively long period of time and sources a greater amount of current to load 32 when V OUT is less than the desired output voltage.

Referring now FIG. 3b , the condition of DC/DC converter 10 having an output voltage that is higher than the desired or target voltage level is shown. Thus, the voltage across load 32 is greater than desired. The output of error amplifier 46 is therefore low relative to the sawtooth waveform. At point 300 b, the leading, or positively sloped, edge of the sawtooth waveform crosses above the output level of output 46 c of error amplifier 46 , thereby sending output 58 c of comparator 58 low. Output 62 a of SR latch 62 has previously been reset by sync pulse 66 .

At point 310 b, the trailing, or negatively sloped, edge of the sawtooth waveform crosses below the output level of output 46 c of error amplifier 46 , thereby sending output 58 c of comparator 58 high. This transition in output 58 c to a high level, in turn, sets output 62 a of SR latch 62 high thereby turning high side FET 14 on and turning off low side FET 12 . The low level of output 46 c relative to the sawtooth waveform results in the sawtooth waveform dropping below the level of output 46 c (at point 310 b) relatively late in the period of the sawtooth waveform. Thus, points 300 a and 310 a are separated by a substantially greater amount of time relative to the situation illustrated in FIG. 3a (i.e., when output 46 c is high relative to the saw tooth waveform and/or when V OUT is less than the target value). Therefore, the period of time during which low side FET 12 is on is of a correspondingly longer duration. Conversely, the period of time during which high set FET 14 is on and sourcing current is relatively brief. Therefore high side FET 14 sources a lesser amount of current to load 32 when V OUT is greater than the desired output voltage.

In both cases, i.e., whether the voltage across load 32 is higher or lower than desired, output 62 a of SR latch 62 goes low based upon sync pulse 66 rather than dependent upon the relative value of the voltage across load 32 . Output 62 a of SR latch 62 remains low at least during the duration of sync pulse 66 . When output 62 a of SR latch 62 is in the low state, high side FET 14 is in the off condition and low side FET 12 is in the on condition, and the voltage drop across RDS ON of low side FET 12 is sampled and held. However, when low side FET 12 is in the on condition the direction of load current I L is toward load 32 . Thus, load current I L flows from ground through the source to the drain of low side FET 12 when low side FET 12 is in the on condition. This direction of current flow through low side FET 12 develops a negative voltage on the drain of low side FET 12 . The magnitude of this negative voltage is the product of I L and the RDS ON of low side FET 12 .

In the embodiment shown, the source of low side FET 12 is electrically connected to ground. However, it is to be understood that low side FET 12 can be alternately configured such as, for example, having its source tied through a resistor to ground, and electrically connecting sensing resistor 26 to the source of low side FET 12 . The net effect is the same, and the virtual ground amplifier continues to drive virtual ground node 30 to virtual ground. In this alternative configuration, current from Sample and Hold circuit 38 is still representative of load current I L except the load-current-induced voltage drop across the added sense resistor is measured rather than the voltage drop across RDS ON of low side FET 12 . This alternative embodiment is best shown in FIG. 6 .

In the embodiment shown, reference voltage supply 48 is described as a fixed voltage supply. However, it is to be understood that reference voltage supply 48 can be alternatively configured, such as, for example, as a bandgap or other fixed voltage source, or may be configured as a Digital to Analog converter or other variable voltage source.

In the embodiment shown, FET 36 is configured as an FET. However, it is to be understood that FET 36 can be alternately configured, such as, for example, an NPN transistor, with Base substituted for Gate, Emitter for Source, and Collector of Drain.

In the embodiment shown, virtual ground amplifier 34 is configured for continuous operation. However, it is to be understood that virtual ground amplifier 34 can be alternately configured, such as, for example, an auto-zeroed amplifier or other non-continuously operating amplifier, as it is needed only when low side FET 12 is in the on state.

In the embodiment shown, DC/DC converter 10 is configured such that a high-level signal at output 62 a of SR latch 62 turns low side FET 12 off and turns on high side FET 14 . However, it is to be understood that DC/DC converter 10 can be alternately configured such that the operational polarity of FET 12 and FET 14 is reversed.

In the embodiment shown, system control circuit 40 is configured to restart the operation of DC/DC converter 10 after a predetermined amount of time following the detection of an overcurrent condition. However, it is to be understood that system control circuit 40 may be alternately configured, such as, for example, to issue a visual or audible warning signal or to completely shut down DC/DC converter 10 .

In the embodiment shown, DC/DC converter 10 is configured with indicator 24 , load capacitor 28 and load 32 connected to node 16 . However, it is to be understood that DC/DC converter 10 can be alternately configured, such as, for example, without interface 24 , load capacitor 28 and load 32 such that a user, designer, or manufacturer can choose and customize circuitry attached to node 16 of DC/DC converter 10 .

While this invention has been described as having a preferred design, the present invention can be further modified within the spirit and scope of this disclosure. This application is therefore intended to cover any variations, uses, or adaptations of the present invention using the general principles disclosed herein. Further, this application is intended to cover such departures from the present disclosure as come within the known or customary practice in the art to which this invention pertains and which fall within the limits of the appended claims.